LINEAR TECHNOLOGLINEAR TECHNOLOGY

LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLOGYJUNE 1997 VOLUME VII NUMBER 2

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,LinearView, Micropower SwitcherCAD and SwitcherCAD are trademarks of Linear Technology Corporation. Otherproduct names may be trademarks of the companies that manufacture the products.

The LTC1605: New 16-Bit,100ksps ADC by Sammy Lum

IntroductionLinear Technology continues its pushinto the high resolution, high perfor-mance analog-to-digital convertermarket with the introduction of theLTC1605. Linear Technology’s first16-bit ADC has outstanding DCaccuracy and a wide analog inputrange of ±10V. The LTC1605 providesan effective solution for a wide rangeof industrial control applications. Itssimple I/O, low power and high per-formance make it easy to design intoapplications requiring wide dynamicrange and high resolution.

Product Features

16-bits with no missing codesand ± 2LSB INL

Single 5V supply with typicalpower dissipation of 55mW

Complete ADC contains sample-and-hold and reference

±10V analog input with ±20Vovervoltage protection on a 5Vsupply

28-pin PDIP, SO and SSOPpackages

The device will not be damaged ifthe analog input is taken outside itsnominal operating range of ±10V; itcan withstand an overvoltage of ±20V,which makes it easier to protect fromthe harsh environments often foundin industrial applications. The largeleast-significant-bit size (305µ V)makes the input signal conditioningcircuitry easier to design. The DCaccuracy is guaranteed to be 16 bitswith no missing codes, with an inte-gral nonlinearity specification of

±2LSB over the industrial tempera-ture range (–40˚C to 85˚C). Thespace-saving SSOP package occupiesonly 0.12 square inch.

Circuit DescriptionWe will begin by briefly describinghow the analog input s ignalprogresses through the various ele-ments of the LTC1605 to become adigital word. First, how does theLTC1605 handle a ±10V analog inputsignal while operating off a 5V sup-ply? It uses a resistor network, asshown in the LTC1605 block diagramin Figure 1. The input signal isattenuated by a factor of eight andthen one-half of the reference voltageis added to the attenuated signal.This reduced internal signal now hasa least-significant-bit size of 38µV.Next, this attenuated signal is sampledand held. The output of the sample-and-hold is d ig i t i zed wi th aswitched-capacitor differential 16-bitsuccessive approximation registerADC. This differential architectureprovides greater immunity to powersupply noise and to other externalnoise sources that can corrupt theresult. Finally the digitized data isoutput to the user at a rate of up to100ksps. The digital output word canbe read as a parallel 16-bit word or itcan be read as two 8-bit bytes. The2-byte output requires using the BYTEpin. With the BYTE pin low the firsteight MSBs are output on the D15–D8pins. When the BYTE pin is takenhigh the eight LSBs replace the eightMSBs. continued on page 3

IN THIS ISSUE...COVER ARTICLEThe LTC®1605:New 16-Bit 100ksps ADC ...........1Sammy Lum

Issue Highlights ........................2

LTC in the News .........................2

DESIGN FEATURESNew 14-Bit 800ksps ADCUpgrades 12-Bit Systems with81.5dB SINAD, 95dB SFDR ........6Dave Thomas and William C. Rempfer

The LT®1495/LT1496: 1.5µARail-to-Rail Op Amps..................8William Jett

The LTC1624: a Versatile, HighEfficiency, SO-8 N-ChannelSwitching Regulator Controller................................................11Randy G. Flatness

The LTC1514/LTC1515 ProvideLow Power Step-Up/Step-DownDC/DC Conversion withoutInductors .................................15Sam Nork

RS485 Transceivers Operate at10Mbps Over Four Hundred Feetof Unshielded Twisted Pair................................................17Victor Fleury

Hot Swapping the PCI Bus................................................21James Herr, Paul Marshikand Robert Reay

DESIGN IDEAS.......................................... 25–31(Complete list on page 25)

DESIGN INFORMATIONUnderstanding and ApplyingVoltage References (Part One)................................................32Mitchell Lee

New Device Cameos ..................37

Design Tools ............................39

Sales Offices ............................40

Linear Technology Magazine • June 19972

EDITOR’S PAGE

Issue HighlightsOur cover article this month intro-duces Linear Technology’s first 16-bitADC, the LTC1605. This product hasoutstanding DC accuracy and a wideanalog input range of ±10V. TheLTC1605 provides an effective solu-tion for a wide range of industrialcontrol applications. Its simple I/O,low power and high performancemakes it easy to design into applica-tions requiring wide dynamic rangeand high resolution.

Also in the data conversion area,we debut a new 14-bit 800ksps ADC,the LTC1419. The LTC1419 satisfiesthe needs of new communications,spectral-analysis, instrumentationand data acquisition applications byproviding an upgrade path to users of12-bit converters. It provides out-standing 81.5dB SINAD (signal-to-noise and distortion ratio) and 95dBSFDR (spurious free dynamic range)for frequency-domain applications,and excellent ±1LSB DNL and nomissing codes performance for time-domain applications.

On the power control front, thisissue introduces two new products:the LTC1624 SO-8 N-channel switch-ing regulator controller and theLTC1514/LTC1515 switched capaci-tor step-down converters. The newestmember of Linear Technology’s nextgeneration of DC/DC controllers, theLTC1624 uses the same constant fre-quency, current mode architectureand Burst Mode™ operation as theLTC1435–LTC1439 controllers, butwithout the synchronous switch. TheLTC1624 can operate in all standardswitching configurations, includingboost, step-down, inverting andSEPIC, without a limitation on theoutput voltage. A wide input voltagerange of 3.5V to 36V allows operationfrom a variety of power sources, fromas few as four NiCd cells up thoughhigh voltage wall adapters.

A unique architecture allows theLTC1514/LTC1515 to accommodatea wide input voltage range (2.0V to10V) and adjust the operating modeas needed to maintain regulation. As

a result, the parts can be used with awide variety of battery configurationsand/or adapter voltages. Low powerconsumption and low external partscount make the parts well suited forspace-conscious low power applica-tions, such as cellular phones, PDAsand portable instruments.

In the interface area, we presentthe LTC1685–87 family of RS485transceivers. These transceivers canoperate at data rates of >40Mbps overone hundred feet of category 5 un-shielded twisted pair. They employ aunique architecture that guaranteesexcellent performance over processand temperature variations, with com-bined propagation delays for both thereceiver and driver of 18.5ns ±3.5ns.A novel short-circuit protection tech-nique permits indefinite shorts (toeither driver or receiver output) topower or ground while sourcing/sink-ing a maximum of 50mA.

Also in this issue, we have a newapplication for the LTC1421 HotSwap™ controller: hot swapping thePCI bus. The PCI bus is widely used inhigh volume personal computers andsingle-board computer designs. Withthe migration of the PCI bus intoservers, industrial computers andcomputer-telephony systems, theability to plug a peripheral into a livePCI slot becomes mandatory. Usingthe LTC1421 to control the powersupplies, a peripheral can be insertedinto a PCI slot without turning off thesystem power.

The Design Ideas section of thisissue includes a –48V to 5V DC/DCconverter that operates from a tele-phone line, a water tank pressuresensor interface, a chopped amplifierthat requires only 5µA of supply cur-rent and a pair of circuits forgenerating a low noise –5V supply foruse in data acquisition applications.The remainder of this section is occu-pied by part one of an epic disquisitionon IC voltage references, to be con-cluded in the August issue.

The issue concludes with a quintetof new device cameos.

LTC in the News…

LTC Resumes Sequential Growthin Sales and Profits“We resumed our sequential growth insales and profits after three flat quar-ters,” says Robert H. Swanson,president and CEO, concerning LinearTechnology Corporation’s latest salesand earnings report. “Customers’demand continued to acceleratethroughout the quarter and showedstrength across all major end ap-plications markets, particularlycommunications. This improving mar-ket should enable us to have furthersequential growth this next quarter.”

Swanson continued, “In order tomeet this anticipated demand, we com-menced production operations in ournew Camas, Washington wafer fabri-cation facility. This will be ramping upover the next few quarters. We will alsobe ramping up our Milpitas fab, Penangassembly and S ingapore testoperations.”

These comments are based on LTC’snet sales for its third quarter, end-ing March 30, 1997, which were$95,033,000. They represented adecrease of 9% over record net sales ayear ago of $104,710,000 for the thirdquarter of 1996. The company alsoreported net income for third quarterof 1997 of $33,980,000 or $0.43 pershare, a decrease of 10% from the$37,764,000, reported for the samequarter of last year.

Sequentially, the results for the thirdquarter were up 5% and 7%, respec-tively, as compared to net sales andnet income reported for the previousquarter, which ended December 29,1996, of $90,080,000 and $31,631,000or $0.40 per share. A cash dividend of$0.05 will be paid on May 14, 1997 toshareholders of record on April 25,1997.

It’s not surprising that the financialcommunity has taken note of theseproceedings. The San Jose MercuryNews presented in a special report,“Silicon Valley’s Top 150” that LTCranks number one in return on salesbased on FY’96 results. The reportappeared in the April 14 “BusinessMonday” edition and showed thatalthough Linear Technology ranked62nd in sales, it was ninth in return onequity, another common measure ofprofitability.

Linear Technology Magazine • June 1997 3

DESIGN FEATURES

The LTC1605 is easily connected toFIFOs, DSPs and microprocessors viathe convert-start input (R/C) and dataready signal (BUSY). With CS low, thefalling edge of the R/C signal will putthe LTC1605 into the hold mode andstart a conversion. BUSY goes lowduring the conversion and the outputdata can be latched after the conver-sion when BUSY goes back high.

1605_01.eps

–

+

VIN (±10V)

20k

REF (2.5V)

CAP

4k VOLTAGE REFERENCE

SAMPLE- AND- HOLD

COMPARATOR

PRECISION 16-BIT DAC

10k 4k

CLOCK

SAR 16 OUTPUT BUFFER

CONTROL LOGIC

CS R/C BYTE

BUSY

16

D15–D0

–

+

1605_02.eps

INTERNAL CAPACITOR

DAC

BANDGAP REFERENCE

VCC

4k

2.2µF

CAP

2.2µF

REF (2.5V)

4

3

FREQUENCY (Hz)

–130–120

–100–110

–80–90

–20

–40

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0

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MAG

NITU

DE (d

B)

1605_04.eps

0 5k 10k 15k 20k 25k 30k 35k 40k 45k 50k

f SAMPLE =100kHz f IN =1kHz SINAD =87.5dB THD =–101.7dB

The LTC1605 has a bandgap refer-ence trimmed to a nominal voltage of2.500V. As shown in Figure 2, it canbe overdriven with an external refer-ence if greater accuracy is needed.The REF pin is buffered by a unity-gain amplifier that drives the internalDAC, along with the level shiftinginput resistor. The output of the bufferis the CAP pin.

Figure 3 shows the fast Fouriertransform (FFT) of a sine wave signalthat has been digitized by theLTC1605. We see a very good ACresponse from the device. Themeasurement was made with the sam-pling frequency set at 100kHz andwith a 1kHz sine wave applied to theanalog input. The key results ob-tained were a signal-to-noise and

Figure 1. The block diagram shows that the LTC1605 has an onboard reference, sample-and-hold amplifier, clock and a 16-bit differentialswitched-capacitor ADC. The analog input accepts a ±10V signal and can withstand an overvoltage of ±20V on a 5V supply.

Figure 2. The LTC1605 has a 2.500V bandgapreference. The internal reference can beeasily overdriven if greater accuracy isneeded. The output of the internal orexternal reference is buffered by a unity-gainamplifier. The buffer drives the internal DACand the input level-shift resistor.

LTC1605, continued from page 1

Figure 3. The FFT plots shows that the THD of the LTC1605 is better than 100dB with a signal-to-noise and distortion of 87.5dB.


DESIGN FEATURES

distortion (SINAD) of 87.5dB and totalharmonic distortion (THD) of–101.7dB. The ±10V input signal wasgenerated with an Audio PrecisionSystem One audio analyzer.

One of the benefits of using a dif-ferential architecture for an ADC isgood power supply rejection. Figure 5shows the power supply rejection ofthe LTC1605 as a function offrequency.

DC and AC PerformanceFigure 4 shows an INL error plot forthe LTC1605. Guaranteed specifica-tions include ±2.0LSB INL (max) andno missing codes at 16 bits over theindustrial temperature range. Theaccuracy of the ADC is trimmed at thefactory and does not carry the over-head for the user associated withautocalibration-type ADCs.

Histogram NoiseMeasurementOne way of measuring the transitionnoise associated with a high resolu-tion ADC is to use a technique wherea fixed DC signal is applied to theinput of the ADC and the resultingoutput codes are collected over a largenumber of conversions. The shape ofthe distribution of codes will give anindication of the magnitude of thetransition noise. For example, in Fig-ure 6 the distribution of output codesis shown for a DC input that has beendigitized 10,000 times. The distribu-tion is Gaussian and the RMS codetransition noise is about 1LSB.

Printed Circuit Board LayoutThe suggested layout for an LTC1605evaluation circuit included herein isan example of a properly designedprinted circuit board that will help

CODE

–2.0

0

–0.5

–1.0

–1.5

1.5

1.0

0.5

2.0IN

L (L

SBs)

65535

1605_03.eps

0 3276816384 49152RIPPLE FREQUENCY (Hz)

–70

–60

–50

–40

–30

–20

POW

ER S

UPPL

Y FE

EDTH

ROUG

H (d

B)

1M

1605_05.eps

1 10010 1k 100k10kCODE

0

500

1500

1000

2500

2000

4000

3500

3000

4500

COUN

T

1605_06.eps

–5 –4 –3 –2 –1 0 1 2 3 4 5

obtain the best performance from this16-bit ADC. The details of the layoutalong with the circuit schematic areshown in Figures 7a–7d. Pay particu-lar attention to the design of the analogand digital ground planes. The DGNDpin of the LTC1605 can be tied to theanalog ground plane. Placing thebypass capacitors as close as pos-sible to the power supply pin and thereference and reference buffer outputpins is very important. A simple RCfilter can be added to the externalinput resistor network, as shown inFigure 8. This will prevent high fre-quency noise from coupling into theanalog input. An NPO-type capacitorgives the lowest distortion. The digitaloutput latches and the onboard oscil-lator have been placed on the digitalground plane. The two ground planes

Figure 5. Power supply feedthrough isextremely low over a wide frequency range.

Figure 6. The histogram shows the LTC1605has a RMS code transition noise of 1LSB.

Figure 7a. Component side silkscreen for thesuggested LTC1605 evaluation circuit

Figure 7b. The top side of the board has thecomponents and shows the analog groundplane.

Figure 7c. The bottom side of the boardshows how the analog and digital groundplanes are isolated.

Figure 4. The INL error plot shows that theLTC1605 is very accurate. This is achievedwithout autocalibration and its associatedoverhead. The accuracy relies on capacitormatching, which is very stable overtemperature and time.

DIGITAL GROUND PLANEANALOG GROUND PLANEANALOG GROUND PLANE


DESIGN FEATURES

are tied together at the power supplyground connection. In this evalua-tion circuit, after the start convertsignal (R/C) has gone low to start aconversion, it is brought back high50ns later. This signal should bebrought back high within 3µs after

D15

+

3

U6A 74HC221

A

B

Q

Q

CEXT

R21, 2kRCEXT15

1

2

4

13

CLK

1605_07d.eps

D15

D14

U1 LTC1605

D13

D12

D11C5 0.1µF

R19 33.2k 1%

C3 0.1µF

C4 2.2µF

C2 2.2µF

EXT

INT

VREF

JP1

C17 10µF

D10

D9

D8

D7

D6

D5

D4

D3

D2

D1

6

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9

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D13

D12

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D8

D7

D6

D5

D4

D3

D2

D1

D0

4R20 1K3

U4B 74HC04

65

U4C 74HC04

C1 15pF

22

1

2

3

4

5

14

23

24

25

26

27

28

VIN Q0

U2 74HC574

19D0

Q1 18D1

Q2 17D2

Q3 16D3

Q4 15D4

Q5 14D5

Q6 13D6

Q7

1 2

12

U4A 74HC04

D71 OC11 CLK

2

7

6

5

4

3

U7 74HC160

CLR

LOAD

RCO 15

2

3 JP3

1

EXT

CLK

INT

2

3 JP5

1

VCC

CS

GND

ENP

10 ENT

QD 11D

QC 12C

QB 13B

QA 14A

2

U8 1MHz OSC

OUT 3GND

1 NA

E2GND

VIN 7V TO 15V

E1U5

LT1121 D16 MBR0520

C6 22µF 10V

GND

1 3

2

VIN VIN

4

U9 LT1019

TRIM 5GND

1 NC12 INPUT3

8

7

6TEMP

NC2

HEATER

OUT

1

9

CLK

D0

D1

D2

D3

D4

D5

D6

D7

2

3

4

5

6

7

8

9

Q0

U3 74HC574

19D0

Q1 18D1

Q2 17D2

Q3 16D3

Q4 15D4

Q5 14D5

Q6 13D6

Q7 12D71 OC11 CLK

AGND1

REF

CAP

AGND2

DGND

BYTE

R/C

CS

BUSY

VANA

VDIG

C8 0.1µF

C7 10µF

VKK

VCC

VKK

VKK VDDVCC

R16 20 C9

0.1µFC10 0.1µF

DIGITAL I.C. BYPASSING

C11 0.1µF

C12 0.1µF

VCC

C13 0.1µF

C14 0.1µF

C15 10µF

2

3 JP4

1

REVERSE

BYTE

NORNAL

VCC

VCC

VCC

20

19

18

17

16

15

14

13

12

11

10

9

8

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6

5

4

3

2

1

GND

GND

CLK

D15

D0

D1

D2

D3

D4

D5

D6

D7

D8

D9

D10

D11

D12

D13

D14

D15

R8, 1.2k D8

R9, 1.2k D9

R10, 1.2k D10

R11, 1.2k D11

R12, 1.2k D12

R13, 1.2k D13

R14, 1.2k D14

R15, 1.2k

R0, 1.2k D0

R1, 1.2k D1

R2, 1.2k D2

R3, 1.2k D3

R4, 1.2k D4

R5, 1.2k D5

R6, 1.2k D6

R7, 1.2k D7

JP2 LED ENABLE

1011

U4E 74HC04

81

2

9EXT_CLK

J1

1

2

AIN

J2

R17 51

R18 200Ω, 1%

U4D 74HC04

C16 1000pF

the start of a conversion to ensurethat no errors occur in the digitizedresult.

ApplicationsWith its overvoltage protected ±10Vanalog input, the LTC1605 fits easilyinto industrial process control, powermanagement and data acquisitionboard applications. In designs wherewide dynamic range is required, onetraditional way to implement this wasto use a PGA with a lower resolutionADC. Now, with a 16-bit ADC, thePGA can be eliminated. For example,with a 12-bit ADC a PGA with a rangeof 1 to 16 would be required to coverthe same range as a 16-bit ADC.

The LTC1605 has sufficient speedto be used in multiplexed applica-

tions. In such a system, there willtypically be an analog multiplexerfollowed by a signal conditioning cir-cuit, which may include filtering,programmable gain, and the like, andthen the ADC. The LTC1605 needs tobe driven from a low source imped-ance to prevent gain errors due to its20kΩ input resistance.

The offset and full-scale error canbe adjusted to zero using three exter-nal resistors along with two trim pots,as shown in Figure 9a. The full-scaleerror and offset for the LTC1605 havebeen factory trimmed with the twoexternal resistors, RA and RB, in place.Figure 9b shows how the device canbe connected if additional trimmingis not needed.

Figure 7d. LTC1605 suggested evaluation circuit schematic; this circuit includes output latches, conversion clock and an optional externalreference.

1605_08.eps

1000pF 33.2k

VIN

CAP

AIN200Ω

Figure 8. A capacitor can be added to theexternal resistor network to form a simplelowpass filter. This will help prevent highfrequency noise from coupling into the analoginput. continued on page 23


DESIGN FEATURES

New 14-Bit 800ksps ADC Upgrades12-Bit Systems with81.5dB SINAD, 95dB SFDR by Dave Thomas and

William C. Rempfer

Higher Dynamic Range ADCsA new 14-bit 800ksps ADC, theLTC1419, satisfies the needs of newcommunications, spectral-analysis,instrumentation and data acquisitionapplications by providing an upgradepath to users of 12-bit converters. Itprovides outstanding 81.5dB SINAD(signal to noise and distortion ratio)and 95dB SFDR (spurious freedynamic range) for frequency-domainapplications, and excellent ±1LSBDNL and no missing codes perfor-mance for time-domain applications.

LTC1419 Features

Complete 14-bit, 800ksps ADC ±1LSB DNL and ±1.25LSB INL

(max) 81.5dB SINAD and 95dB SFDR Low power—150mW on ±5V

supplies Nap/Sleep power-down modes Small Footprint—28-pin SO or

SSOP

The Big Brotherof the LTC1410The new LTC1419 is a 14-bit deriva-tive of the LTC1410 ADC from LTC. Ithas a similar pinout and function, asshown in the block diagram in Figure1. Inputs are received by the wide-band differential sample-and-hold(S/H). This S/H is capable of sam-pling to Nyquist and beyond andoperates with either differential orsingle-ended signals. In contrast tosome converters, which must bedriven differentially to perform well,this ADC operates equally well withsingle ended or differential signals.(To digitize a single-ended signal, sim-ply ground the negative input.)

1419_1.eps

REF COMP

4.1V

–AIN

SAMPLE/ HOLD

COMPARATOR

LTC1419

14 14

2k

BUSY

SHDN RD CONVST CS

14-BIT CAPACITIVE

DAC

+AIN

VREF 2.5V

LOW DRIFT VOLTAGE

REFERENCE

CLOCK

SAR OUTPUT BUFFER

CONTROL LOGIC

Figure 1. This complete 800ksps, 14-bit ADC has a wideband S/H that cleanlysamples wideband input signals

FREQUENCY (Hz)

–140

0

–10

–20

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ITUD

E (d

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1419_02.eps

0 50k 100k 150k 200k 250k 300k 350k 400k

85dB SFDR (GOOD 12-BIT ADC)

10dB EXTRA SFDR OF 14-BIT LTC1419

95dB SFDR

fSAMPLE = 800ksps fINPUT = 100kHz SINAD = 81.5dB

Figure 2. The LTC1419 gives a 10dB improvement in spectral purity over even the best 12-bitdevices. This FFT shows the LTC1419’s outstanding 81.5dB SINAD and 95dB SFDR.


DESIGN FEATURES

ANALOG INPUT FREQUENCY (Hz)

EFFE

CTIV

E BI

TS SINAD (dB)

10k 100k 1M 2M

1419 TA02

1k

14

13

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7

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5

4

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2

86

80

74

68

62

Figure 3. As the input signal frequency isincreased, many ADCs start to loose spectralpurity due to distortion or noise. TheLTC1419 has essentially flat SINAD andeffective bits out to Nyquist. Even whenundersampling a 2MHz input, it maintains12-bit performance.

+

–

+

–

+ –

1419_05.eps

SINGLE ENDED ADC

DATA OUTPUT

14

VSIGNAL

AIN

VIN

VNOISE

GROUND NOISE

A

(a)

+

–

+

–

+

–

+ –

1419_06.eps

LTC1419 DATA OUTPUT

14

VSIGNALAINVIN

VNOISE

B

(b)

The ADC uses a switched-capaci-tor SAR technique, similar to that ofits predecessor, that yields excellentDC specifications and stability. It is aclean, simple to use design that deliv-ers 800ksps conversion rate at lowpower levels.

The ADC has a flexible parallelI/O, which can interface to a DSP, amicroprocessor, an ASIC or to dedi-cated logic. Conversions can be startedeither under command of a DSP ormicroprocessor or from an external

sample clock signal. An output dis-able allows the outputs to bethree-stated.

The LTC1419, like the 12-bitLTC1410, operates from ±5V sup-plies and draws 150mW of power.

10dB Extra Dynamic Rangefor Signal ApplicationsThe LTC1410 is probably the cleanest12-bit ADC on the market. The partachieves 72dB SINAD and has anSFDR of better than 85dB. Thesenumbers approach the theoreticallimit for 12 bits. Figure 2 shows theimprovements possible with theLTC1419. The 14-bit device achieves81.5dB SINAD (an increase of roughly10dB over the 12-bit device). TheSFDR increases to 95dB. This givesthe converter 10dB more resolvingpower to pick out small signals incommunication and spectrum-analy-sis applications. This clean samplingcapability is maintained even for wide-band inputs. Figure 3 shows higheffective bits and SINAD for inputsbeyond Nyquist.

Four Times Improvementin DC ResolutionThe 12-bit LTC1410 guarantees±1LSB of integral and differentialnonlinearity (INL and DNL). The 4096steps over a 5V input range yield anLSB of 1.22mV. The new 14-bit partalso maintains excellent linearity(±1LSB DNL, ±1.25LSB INL); resolu-tion is increased and the LSB isreduced to 305µV.

Noise-RejectingDifferential InputsWith its higher dynamic range andresolution, the LTC1419 can digitizesignals more cleanly than previousdevices. However, as the resolutionincreases and the noise floor drops,other system noises may show upunless precautions are taken. Thedifferential input of the new ADCprovides a way to keep noise out.Noise can be introduced in a numberof ways including ground bounce,digital noise and magnetic and ca-pacitive coupling (see Figure 4a). Allof these sources can be reduced dra-

COMMON MODE INPUT FREQUENCY (Hz)

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COM

MON

MOD

E RE

JECT

ION

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1419_07.eps

1k 10k 100k 1M

Figure 4. a) In high resolution ADC systems,noise sources such as ground noise andmagnetic coupling can contaminate theADC’s input signal. b) The LTC1419’sdifferential inputs can be used to reject thisnoise, even if it is at high frequencies.

matically by measuring differentiallyfrom the signal source, as in Figure4b. The high CMRR of the differentialinput (Figure 5) allows the LTC1419to reject resulting common mode noiseby over 60dB and maintain a cleansignal.

Other Nice FeaturesSeveral other features make theLTC1419 flexible and easy to use:

Both analog inputs have infiniteDC input resistance, whichmakes them easy to multiplex orAC couple.

The separate convert-start inputpin allows precise control overthe sampling instant. The S/Haperture delay is less than 1nsand the aperture jitter is below1ps RMS.

Conversion results are availableimmediately after a conversionand there is no latency in thedata (no pipeline delay). This isideal for both single shot andrepetitive measurements.

The low 150mW powerdissipation can be reducedfurther using the ADC’s Nap andSleep power-down modes. Wakeup from Nap mode isinstantaneous. Sleep mode wakeup time is several milliseconds.

The LTC1419 is the industry’ssmallest high speed 14-bitconverter: it is available in a 28-pin SSOP package.

Figure 5. The common mode rejection of theanalog inputs rejects common mode inputnoise frequencies to beyond 10MHz.

continued on page 23


DESIGN FEATURES

The LT1495/LT1496:1.5µA Rail-to-Rail Op AmpsIntroductionMicropower rail-to-rail amplifierspresent an attractive solution for bat-tery-powered and other low voltagecircuitry. Low current is alwaysdesirable in battery-powered applica-tions, and a rail-to-rail amplifier allowsthe entire supply range to be used byboth the inputs and the output, maxi-mizing the system’s dynamic range.Circuits that require signal sensingnear either supply rail are easier toimplement using rail-to-rail amplifi-ers. However, until now, no amplifiercombined precision offset and driftspecifications with a maximum qui-escent current of 1.5µA.

Operating on a minuscule 1.5µAper amplifier, the LT1495 dual andLT1496 quad rail-to-rail amplifiersconsume almost no power whiledelivering precision performance as-sociated with much higher currentamplifiers. Input offset voltage is only375µV maximum, with a maximumdrift of 2µV/˚C, and input offset cur-rent is 100pA maximum. The low biascurrents (1nA maximum) and low off-set currents of these amplifiers permitthe use of megohm-level source resis-tors without introducing significanterrors. A minimum open-loop gain of

100V/mV guarantees that gain errorsare small. The device characteristicschange little over the supply range of2.2V to ±15V: worst-case supplyrejection is 90dB and the commonmode rejection ratio is greater than90dB. The LT1495 dual amplifier isavailable in the 8-pin SO and the 8-pin mini-DIP package. The LT1496quad amplifier is available in 14-pinSO and 14-pin DIP.

The LT1495/LT1496 feature “over-the-top” operation: the ability tooperate normally with the inputsabove the positive supply. The de-vices also feature reverse-batteryprotection.

Start-Up CharacteristicsMicropower op amps are sometimesnot micropower during start-up,wreaking havoc on low current sup-plies. In the worst case, there may notbe enough supply current availableto take the system up to nominalvoltages. Figure 1 shows a graph ofLT1495 supply current versus sup-ply voltage for the three limit cases ofinput offset that could occur duringstart-up. The circuits are shown inFigure 2. One circuit creates a posi-tive offset, forcing the output to comeup saturated high, another circuitcreates a negative offset, forcing theoutput to come up saturated low and

by William Jett

SUPPLY VOLTAGE (V)

0

5

4

3

2

1

SUPP

LY C

URRE

NT P

ER A

MPL

IFIE

R (µ

A)

5

1495_01.eps

0 1 2 3 4

OUTPUT HIGH

OUTPUT LOW

OUTPUT VS/2

1495_02.eps

OUTPUT HIGH

VS

OUTPUT AT VS/2

VS

VS 2

–

+

OUTPUT LOW

VS

–

+LT1495 LT1495

+

–LT1495

+

+

V–

V+

1495_03.eps

Q6

Q4 Q16 Q17 Q19

Q18Q22

C1

D1 D2 D3

Q3

Q11

Q2 Q7INN

INP

Q1

Q10 Q13Q20

Q21

OUT

Q12

Q14 Q15

Q5

Q8R1 R2 I2

+I1

D6

D5

D4

D7

Q9

VBIAS

Figure 1. LT1495 supply current vs supplyvoltage for the three limit cases of inputoffset that could occur during start-up

Figure 2. Circuits for start-up characteristics

Figure 3. LT1495 simplified schematic


DESIGN FEATURES

the last brings up the output at halfsupply. In all cases, the supply cur-rent is well behaved. Supply currentis highest with the output forced high,so if one amplifier is unused, it is bestto force the output low or to halfsupply.

A Low CurrentRail-to-Rail ArchitectureThe simplified schematic, Figure 3,details the circuit design approach ofthe LT1495/LT1496. The amplifiertopology is a 3-stage design, consist-ing of a rail-to-rail input stage thatcontinues to operate with the inputsabove the positive rail, a folded-

cascode second stage that developsmost of the voltage gain, and a rail-to-rail common-emitter stage thatprovides the current gain.

The input stage is formed by twodifference amps, Q1–Q2 and Q3–Q6.For signals with a common modevoltage between VEE and (VCC – 0.8V),Q1 and Q2 are active. When the inputcommon mode exceeds (VCC – 0.8V),Q7 turns on, diverting the currentfrom difference amp Q1–Q2 to cur-rent mirror Q8–Q9. The current fromQ9 biases on the other differenceamp, consisting of PNPs Q5–Q6 andNPNs Q3–Q4. Though Q5–Q6 aredriven from the emitters rather than

the base, the basic difference ampaction is the same. When the com-mon mode voltage is between (VCC –0.8V) and VCC, devices Q3 and Q4 actas followers, forming a buffer betweenthe amplifier inputs and the emittersof the Q5–Q6. If the common modevoltage is taken above VCC, Schottkydiodes D1 and D2 reverse bias anddevices Q3 and Q4 then act as diodes.The difference amp formed by Q5–Q6operates normally, but the input biascurrent increases to the emitter cur-rent of Q5–Q6, which is typically180nA.

The collector currents of the twoinput pairs are combined in the sec-ond stage consisting of Q11–Q16,which furnishes most of the voltagegain. Capacitor C1 sets the amplifierbandwidth. The output stage is con-figured for maximum swing by theuse of common-emitter output devicesQ21 and Q22. Diodes D4–D6 andcurrent source Q15 set the outputquiescent current.

TC VOS (µV/°C)

UNIT

S30

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5

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1495_04.eps

2.0–1.6–2.0 –0.8 0 0.8 1.6

100 AMPLIFIERS VS = ±2.5V –40°C TO 85°C

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1495_05.eps

1/2 LT1495

1/2 LT1495

100pF

R1 10M

R2 9k

1.5V

1.5V

R3 2k

0µA-200µA

R4 10k

INPUT CURRENT

2 × 1N914

FREQUENCY (Hz)

–60

–40

–20

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–30

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–10

GAIN

(dB)

1000

1495_07.eps

0 10 100

–

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10k

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200k

100k

10nF

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215k

215k215k

15nF15nF

30nF

eIN

1/2 LT1495

200k

80.6k

10nF

100k

100nF169k169k169k

15nF15nF

30nF

LT1495/96 •TA03

VS = 5V, 0V IS = 2µA + eIN/150k ZEROS AT 50Hz AND 60Hz

OUTPUT

10k

Figure 4. Input offset voltage driftdistribution plot

Figure 5. 0nA–200nA current meter

Figure 6. 6th order 10Hz elliptic lowpass filterFigure 7. Frequency response of Figure 6’s6th order elliptic lowpass filter


DESIGN FEATURES

PerformanceTable 1 summarizes the performanceof the LT1495/LT1496. As can beseen, operation is fully specified at3V, 5V and ±15V. Input offset voltagedrift is very low, guaranteed less than2µV/˚C; a distribution plot is shownin Figure 4.

ApplicationsThe ability to accommodate any inputor output signal that falls within theamplifier supply range makes theLT1495/LT1496 very easy to use.The following applications highlightsignal processing at low currents.

Nanoampere MeterA simple 0nA–200nA meter operatingfrom two flashlight cells or one lithiumbattery is shown in Figure 5. Thereadout is taken from a 0µA–200µA,500Ω analog meter; the LT1495 sup-plies a current gain of 1000 in thisapplication. The op amp is configuredas a floating I-to-I converter. It con-sumes only 3µA when not in use, sothere is no need for an on/off switch.Resistors R1, R2 and R3 set the cur-rent gain. R3 provides a ±10% full-scale adjust for the meter movement.With a 3V supply, maximum currentin the meter is limited by R2 + R3 toless than 300µA, protecting the move-ment. Diodes D1 and D2 and resistorR4 protect the inputs from faults upto 200V. Diode currents are below1nA in normal operation, since themaximum voltage across the diodesis 375µV, the VOS of the LT1495. C1acts to stabilize the amplifier, com-pensating for capacitance betweenthe inverting input and ground. Theunused amplifier should be connectedas shown for minimum supply cur-rent. Error terms from the amplifier(base currents, offset voltage) sum toless than 0.5% over the operatingrange, so the accuracy is limited bythe analog meter movement.

6th Order, 10HzElliptic Lowpass FilterFigure 6 shows a 6th order, 10Hzelliptic lowpass filter with zeros at50Hz and 60Hz. Supply current isprimarily determined by the DC load

on the amplifiers and is approximately2µA + VO/150k (9µA for VO = 1V). Theoverall frequency response is shownin Figure 7. The notch depth of thezeros at 50Hz and 60Hz is nearly60dB and the stopband attenuationis greater than 40dB out to 1kHz. As

with all RC filters, the filter charac-teristics are determined by theabsolute values of the resistors andcapacitors, so resistors should have a1% tolerance or better and capacitorsa 5% tolerance or better.


VS V0,V5= VS V51±=

VS V0,V3=

reifilpmAreptnerruCylppuS Aµ5.1 Aµ0.2 xaM

egatloVtesffOtupnI V(Aµ573 S )V5= Vµ575 xaM

V(Vµ574 S )V3= xaM

tfirDtesffOtupnI C˚/Vµ2 xaM

egatloVesioNtupnI)zH01otzH1.0(

Vµ4 P-P Vµ4 P-P pyT

tnerruCsaiBtupnI Ap0001 Ap0001 xaM

tnerruCtesffOtupnI Ap001 Ap001 xaM

R(niaGpooL-nepO L )k001= V(Vm/V001 S )V5=V(Vm/V05 S )V3=

Vm/V001 niMniM

oitaRnoitcejeRedoMnommoC

V MC V,V4otV0= S V5= Bd09 niM

V MC V,V01otV0= S V5= Bd47 niM

V MC V41otV51–= Bd001 niM

oitaRnoitcejeRylppuSrewoP

VS VotV2.2= S V21= Bd09 niM

VS VotV5±= S V51±= Bd49 niM

woL:egatloVnoitarutaStuptuO

daoLoN Vm001 xaM

I KNIS Aµ001= Vm014 xaM

RL k001= Vm005 xaM

hgiH:egatloVnoitarutaStuptuO

daoLoN Vm07 xaM

I ECRUOS Aµ001= Vm023 xaM

RL k001= Vm083 xaM

tnerruCtiucriCtrohS Aµ007 Aµ007 niM

etaRwelS sm/V4.0 sm/V4.0 niM

tcudorPhtdiwdnaB-niaG zHk3 zHk3 pyT

Table 1. LT1495/LT1496 key specifications: 25˚C


DESIGN FEATURES

The LTC1624: a Versatile, HighEfficiency, SO-8 N-Channel SwitchingRegulator Controller by Randy G. Flatness

IntroductionThe LTC1624 is the newest memberof Linear Technology’s next genera-tion of DC/DC controllers. This 8-pincontroller uses the same constantfrequency current mode architectureand Burst Mode operation as theLTC1435–LTC1439 controllers, butwithout the synchronous switch. TheLTC1624, like the other members ofthe family, drives a cost-effective,external N-channel MOSFET for thetopside switch and maintains lowdropout operation previously avail-able only with P-channel MOSFETs.

The LTC1624 can be configured tooperate in all standard switchingconfigurations, including boost, step-down, inverting and SEPIC, without alimitation on the output voltage. Awide input voltage range of 3.5V to36V allows operation from a variety ofpower sources, from as few as fourNiCd cells up though high voltagewall adapters. Tight load regulation,coupled with a reference voltagetrimmed to 1%, provides very accu-rate output voltage control.

The 8-pin SO package, the need forfew external components and N-chan-nel drive make high efficiency DC/DCconversions possible in the extremelysmall PC board space available intoday’s portable electronics.

High PerformanceArchitectureThe LTC1624 is a current modeswitching regulator controller oper-ating at an internally set frequency of200kHz. A user selectable senseresistor (RSENSE) sets the maximum

–

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Q

DROP OUT DET

ONE SHOT 400ns

INT VCC

5.6V INT VCC

REG

SWITCH LOGIC

OSC SLOPE COMP.

I1

I2

VIN

VIN

CINVIN

RSENSE

SENSE–

INT VCC

DB

ST

B

3µA 3µA

RUN

1.5V

SLOPE COMP.

8k 8k

18

2

3

3µA

180k

8k

1.19V

1.19Vgm = 1m

EA

0.8V

30k

0V1.28V

RC

R1 R2

ITH/RUN

VFB

4GND

7BOOST

CB

6TG

5SW

D1

N-CHANNEL MOSFET

VFB

CC

COSC

Ω

Figure 1. LTC1624 block diagram


DESIGN FEATURES

current. Referencing the sense resis-tor to VIN instead of VOUT removes thelimitation on maximum output volt-age. (The LTC1435–LTC1439 have amaximum output voltage of 10V.) Ablock diagram of the LTC1624 config-ured as a step-down regulator isshown in Figure 1.

During normal operation, the topMOSFET is turned on during eachcycle when the oscillator sets a latch,and turned off when the main currentcomparator resets the latch. The peakinductor current at which the currentcomparator resets the latch is con-trolled by the voltage on the ITH/RUNpin, which is the output of the erroramplifier. The error amplifier receivesan output feedback voltage from anexternal resistive divider though theVFB pin. When the load currentincreases, it causes a slight decreasein VFB relative to the 1.19V reference,which in turn causes the ITH/RUNvoltage to increase until the averageinductor current matches the newload current. After the top MOSFET isturned off, the internal bottomMOSFET is turned on during each

cycle for approximately 300ns–400nsto ensure that the bootstrap capaci-tor CB is always recharged.

The value of RSENSE is chosen basedon the required output current. TheLTC1624 current comparator has amaximum threshold of 160mV/RSENSE. The current comparatorthreshold sets the peak of the inductorcurrent, yielding a maximum averageoutput current (IMAX) equal to thepeak value less half the peak-to-peakripple current, DIL. For step-downapplications, the value of the senseresistor is set to 100mV/IOUT(MAX). Toprevent overcurrent during outputshort-circuit conditions, the operat-ing frequency is dropped to around30kHz to ensure the inductor’s cur-rent safely decays in each cycle.

The LTC1624 includes protectionagainst output overvoltage conditionsor transients. An overvoltage com-parator monitors the output voltageand forces the topside MOSFET offand keeps it off when the outputvoltage is greater than 7.5% of itsregulated value.

Combined RUN/Compensation/Soft-Start PinThe ITH/RUN pin is a multifunctionpin, providing shutdown, control-loopcompensation and optional soft-start.Internal slope compensation (requiredwith constant frequency designs)coupled with external compensation(RC, CC in Figure 1) provides optimumload-step response. The peak induc-tor current is controlled by the voltageat the ITH/RUN pin. The nominal rangefor the ITH/RUN pin is from 1.2V to2.4V with the load dependent charac-teristics shown in Figure 2a.

Pulling the ITH/RUN pin below its1.2V soft clamp voltage puts theLTC1624 into shutdown with a typi-cal quiescent current of 15µ A.Releasing the ITH/RUN pin allows aninternal 3µA current source to pullup the voltage on the ITH/RUN pin,charging the compensation capacitorCC. When the voltage on the ITH/RUNpin reaches 0.8V, the main controlloop is enabled with the ITH/RUN volt-age pulled up by the error amplifier,as shown in Figure 2b.

Soft-start can be implemented byincreasing the voltage on the ITH/RUN pin from 1.2V to its 2.4V maxi-mum, because the internal currentlimit is also ramped at a proportionalrate (See Figure 2). Soft-start reducesinrush surge currents from VIN bygradually increasing the internal cur-rent limit. This pin can also be used tocontrol power supply sequencing.Current limit begins at approximately10mV/RSENSE and ends at 100mv/RSENSE. The circuit in Figure 3c showshow to implement soft-start. Thecapacitor C1 starts at 0V when VIN isapplied and diode D1 pulls the ITH/RUN pin low. As C1 charges, thevoltage on ITH/RUN also increases ata proportional rate together with thecurrent limit. If soft-start is notneeded, the circuits in Figures 3a or3b can be used. An open-drainMOSFET in Figure 3b directly pullsthe ITH/RUN to ground, forcingshutdown.

Loop compensation is accom-plished with RC and CC. For step-downapplications, the typical time con-stant created by RC and CC should bearound 50kHz (1/4 the oscillator fre-quency) as a good starting point. The

(b)

VITH (V)

50

03

150

200

I I TH (µ

A)

2.41.2

1624_02.eps

0 0.8

SHUTDOWN

ACTIVE MODE

IOUT

(a)

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2.4V I

TH/R

UN (V

)

IOUT (MAX)

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0.81624_03.eps

3.3V OR 5V ITH/RUN

D1 D1

CC

RC

ITH/RUN

(b)(a) (c)

CC

RC

ITH/RUN

CC

RC

C1

R1

Figure 2. ITH/RUN pin characteristics

Figure 3. Driving the ITH/RUN pin


DESIGN FEATURES

value of RC should generally trackRSENSE. For example, for a 2A maxi-mum output current, set RSENSE =0.05Ω, with RC = 5.1k and CC = 620pF.With a 4A output current, set RSENSE= 0.025Ω, with RC = 3k and CC =1000pF. Using these guidelines as astarting point, the final values of com-pensation components can be foundusing a load-transient step andobserving the output voltage tran-sient response.

To boost low current efficiency, theLTC1624 behaves like the LTC1435/LTC1438 during low current opera-tion by using Burst Mode operation.When the load current falls to thepoint where the peak inductor cur-rent is approximately 20mV/RSENSE,the topside MOSFET is held off andthe output capacitor supports theload, initiating Burst Mode opera-tion. During this phase the outputvoltage is decaying and the output ofthe error amp (ITH/RUN pin) isincreasing. The topside MOSFET is

not switched, saving power and boost-ing efficiency. When the ITH/RUN pinvoltage exceeds 1.5V the drive is re-turned to the topside MOSFET andthe output voltage ramps up. Figure 4shows the output voltage ripple forcontinuous mode at higher outputcurrents (Trace A) and for Burst Modeoperation at lower output currents(Trace B).

Floating MOSFET DriverAn internal 5.6V supply derived fromVIN provides power to drive the topsideMOSFET (refer to Figure 1). The gatedrive for the topside MOSFET origi-nates from a floating driver operatingfrom the BOOST pin to the SW pin. Anexternal bootstrap capacitor (CB) con-nected from BOOST to SW suppliesthe gate-drive voltage. Capacitor CB ischarged through an internal highvoltage diode from the 5.6V supplywhen the SW pin is low. This elimi-nates the need for an externalSchottky diode in most applications.

When the topside MOSFET isturned on, the driver places the volt-age on CB across the gate-source ofthe MOSFET. This enhances theMOSFET and turns on the top sideswitch. The switch node SW rises toVIN and the BOOST pin rises to VIN +5V. A small internal N-channelMOSFET pulls the switch node (SW)to ground during each cycle after thetopside MOSFET turns off ensuringthe bootstrap capacitor is kept fullycharged.

Significant efficiency gains can berealized by supplying the topsidedriver operating voltage from the out-put, since the VIN current resultingfrom the driver and control currentswill be scaled by a factor of (DutyCycle)/(Efficiency). For 5V regulatorsthis simply means connecting theBOOST pin though a small Schottkydiode (like a CMDH-3) to VOUT.

For operation with VIN < 5V, highergate-drive voltage and higher effi-ciency can be obtained by connectinga Schottky diode from VIN to BOOST.This technique parallels the internalboost diode and increases the en-hancement of the MOSFET. This limitsthe maximum input voltage to 8V soas not to exceed the maximum volt-age from boost to switch of 8V.

Low DropoutAn important feature for extractingmaximum energy from low voltagebattery packs is low dropout. TheLTC1147 (another 8-pin controller)achieves this by using a P-channelMOSFET switch that can operate at100% duty cycle. The LTC1624 usesan N-channel MOSFET to accrue thebenefits of lower RDS(ON) and lowercost than corresponding P-channelMOSFETs.

Driving N-channel MOSFETsrequires periodic recharging of thebootstrap capacitor, CB. This can onlyoccur when the top MOSFET is turnedoff and the switch node is low (duringthe off-time). The ratio of maximumon-time to the clock period is definedas the duty cycle. The LTC1624 detects

LOAD CURRENT

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90

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EFFI

CIEN

CY (%

)

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1624_05.eps

1mA 1A100mA10mA

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LTC1435

LTC1624

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1624_06.eps

LTC1624

SENSE–

ITH/RUN

VFB

CC, 570pFRC 5.1k

100pF

GND

VIN

BOOST

TGCB 0.1µF

1000pF

M1 Si4412DY

L1 10µH

RSENSE 0.05Ω

R2 35.7k

VOUT 3.3V/2A

R1 20k

CIN 22µF 35V x2

COUT 100µF 10V x2

VIN 4.5V–25V

D1 MBRS340T3

SW4 5

6

7

8

3

2

1

Figure 5. Efficiency comparison ofsynchronous and nonsynchronousstep-down converters

Figure 6. High performance 3.3V/2A step-down DC/DC converter

Figure 4. Output ripple: a) continuous mode;b) Burst Mode

10µs/DIV

(a)

(b)

50mV/DIV


DESIGN FEATURES

the number of clock cycles the topMOSFET is allowed to remain on.After two clock cycles, the topside isturned off and a minimum off-time isforced. In this mode the duty cycle is95% and the topside is switching atFOSC/2. This extends the maximumduty cycle from 90% to 95% and stillguarantees that the bootstrap capaci-tor remains charged.

Giving Up theSynchronous SwitchThe LTC1624 nonsynchronousN-channel controller saves switchinglosses (gate-charge current) of thesynchronous MOSFET at the expenseof increased loss due to the Schottkydiode during some operating condi-tions. Printed circuit board area isminimized by fewer required externalcomponents and an 8-pin SO pack-age footprint.

The LTC1624 controller shares thesame loss-reducing techniques asother members of the LTC143X family.Figure 5 shows efficiency plots of two3.3V converters, a nonsynchronousLTC1624 and a 16-pin synchronous

LTC1435 operating at VIN = 10V. Thesame common external componentsand operating frequency are main-tained for both circuits.

At low currents (IOUT < 100mA),while in Burst Mode operation, theefficiency of the LTC1624 exceedsthat of the LTC1435. This is due tosaving gate-charge current by notswitching the bottom synchronousMOSFET. At higher output currents,as expected, the Schottky diode lossdominates and the efficiency of theLTC1435 circuit is greater than thatof the LTC1624 circuit.

At lower input voltages, when theduty cycle forces the topside MOSFETon longer, the loss due to the Schot-tky diode decreases and theefficiencies of the synchronous andnonsynchronous designs converge.At higher input voltages the efficiencydifference in the low current regionincreas ingly favors the non-synchronous LTC1624, but at highcurrents the synchronous LTC1435continues to win.

LOAD CURRENT

50

60

70

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100EF

FICI

ENCY

(%)

10A

1624_07.eps

1mA 1A100mA10mA

VIN = 5V

V IN = 10V

VIN = 20V

+

+

1624_08.eps

LTC1624

SENSE–

ITH/RUN

VFB

CC, 330pFRC 5k

100pF

GND

1

2

3

4

8

7

6

5

VIN

BOOST

TGCB 0.1µF

1000pF

D2 CMDSH-3

M1 Si4412DY

L1 20µH

D1 MBRS130LT3

RSENSE 0.04Ω

R2 35.7k 1%

VOUT 12V/1A

R1 3.92k 1%

CIN 22µF

35V x2

COUT 100µF 16V x2

VIN 5V

SW

Minimum Externals,Maximum VersatilityThe LTC1624 can be used in a widevariety of switching regulator appli-cations, the most common being thestep-down converter. Other switch-ing regulator architectures includestep-up, SEPIC and positive-to-nega-tive converters.

The basic step-down converter isshown in Figure 6. This applicationshows a 3.3V/2A converter operatingfrom an input voltage range of 4.5V to25V. The efficiency for this circuit isshown in Figure 7.

Step-up and SEPIC applicationsrequire a low-side switch pulling theinductor to ground (see Figures 8 and10). Since the source of the MOSFETmust be grounded, the switch pin(SW) on the LTC1624 is also groundedin order for the driver to supply agate-to-source signal to control theMOSFET. In these applications, thevoltage on the boost pin is a constant5V, resulting in a 0V–5V gate-drive

LOAD CURRENT

50

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100

EFFI

CIEN

CY (%

)

1A

1624_09.eps

1mA 100mA10mA

++

+

1624_10.eps

LTC1624

SENSE–

ITH/RUN

VFB

CC, 330pFRC 10k

100pF

GND

1

2

3

4

8

7

6

5

VIN

BOOST

TGCB 0.1µF

L1a, L1b:CTX50-4

1000pF

M1 Si4412DY

L1a

L1b

D1 MBRS130LT3RSENSE

0.082Ω

R2 35.7k 1%

VOUT 12V/0.5A

R1 3.92k 1%

CIN 22µF

35V x2

22µF 35V

COUT 100µF 16V x2

VIN 5V-15V

SW

Figure 7. Efficiency plot for Figure 6’s circuitFigure 8. 12V/1A step-up converter

Figure 9. Efficiency plot for Figure 8’s circuit Figure 10. 12V/0.5A DC/DC converter operates from 5V–15V inputs



DESIGN FEATURES

The LTC1514/LTC1515 Provide LowPower Step-Up/Step-Down DC/DCConversion without Inductors by Sam Nork

IntroductionMany applications must generate aregulated supply from an input sourcethat may be above or below the desiredregulated output voltage. Such appli-cations place unique constraints onthe DC/DC converter and, as a gen-eral rule, add complexity (and cost) tothe power supply. A typical exampleis generating 5V from a 4-cell NiCdbattery. When the batteries are fullycharged, the input voltage is around6V; when the batteries are near end oflife, the input voltage may be as low as3.6V. Maintaining a regulated 5Voutput for the life of the batteriestypically requires an inductor-basedDC/DC converter (for example, aSEPIC converter) or a complex hybridstep-up/step-down solution. TheLTC1514/LTC1515 family of switchedcapacitor DC/DC converters handlesthis task with only three externalcapacitors (Figure 1).

A unique architecture allows theparts to accommodate a wide inputvoltage range (2.0V to 10V) and ad-just the operating mode as needed tomaintain regulation. As a result, theparts can be used with a wide varietyof battery configurations and/oradapter voltages. Low power con-sumption (IQ = 60µA typical) and lowexternal parts count make the partswell suited for space-conscious low

power applications, such as cellularphones, PDAs and portable instru-ments. The parts come in adjustableand fixed output-voltage versions andinclude additional features such aspower-on reset capability (LTC1515family) and an uncommitted com-parator that is kept alive in shutdown(LTC1514 family).

Regulator OperationThe parts combine the relativelysimple architecture of a step-up volt-age doubler with a gated-switchstep-down regulator to create asimple-to-use step-up/step-downregulator. The trick, of course, is know-ing when to step up and when to stepdown. The block diagram shown inFigure 3 illustrates how these partsfunction.

The regulator sections of both theLTC1514 and the LTC1515 consist ofan oscillator, switch network (S1–S4),reference, comparator and controllogic. Regulation is achieved by com-paring the divided-down outputvoltage to the internal reference volt-age. When the divided output dropsbelow the reference voltage, the switchnetwork is enabled to boost the out-put back into regulation. Hysteresisin the comparator forces the regula-tor to burst on and off, and causes

approximately 100mV of peak-to-peakripple to appear at the output. Byenabling the regulator only whenneeded, the LTC1514 and LTC1515are able to achieve high efficiencieswith low output load currents.

The action of the switch network iscontrolled by internal circuitry thatsenses the voltage differential betweenVIN and VOUT. When the input voltageis lower than the output voltage, theswitch network operates as a step-upvoltage doubler with a free-runningfrequency of 650kHz (typical). Whenthe input voltage is greater than theoutput, the switch network operatesas a step-down gated switch. The netresult is a stable, tightly regulatedoutput supply that can tolerate widelyvarying input voltages and loadtransients.

Inrush CurrentsNo Longer a ProblemSwitched capacitor DC/DC convert-ers are touted for their micropoweroperation and are generally used inlight-load applications. However,despite their low power design andenvironment, they have two undesir-able tendencies: 1) to pull very highinrush currents from the input supplyduring power-up; and 2) to generatehigh input and output current spikes

1514_01.eps

LTC1515-3.5

100k

SHDN1ON OFF

5V

RESET

3.3V

POR2

5/33

GND

VOUT

VIN

VIN = 4 CELLS

VOUT = 5V ± 4% OR 3.3V ± 4% IOUT = 0 TO 50mA

C1+

C1–4

8

7

6

5 0.22µF 10µF 10µF+ +

VIN (VOLTS)

4.8

4.9

5.0

5.1

5.2

V OUT

(VOL

TS)

1514_XX.eps

3 4 65

Figure 1. Programmable 5V/3.3V power supply with power-on reset Figure 2. VOUT vs VIN for Figure 1’s circuit


DESIGN FEATURES

when large VIN to VOUT differentialsare present. These traits can causemany bad things to happen. If theswitched cap converter is being pow-ered by another low power DC/DCconverter, the sudden inrush currentduring power-up, which can easilyreach several hundred milliamps, maydisrupt regulation of the main powersupply. High switching currents dueto large VIN to VOUT differentials cancause excessive output ripple and/orpoor regulation. As a result, mostswitched cap voltage converters haverather limited allowable VIN to VOUTdifferentials. These problems areaddressed by the LTC1514/LTC1515.

Internal soft-start circuitry con-trols the rate at which VOUT can becharged from 0V to its final regulated

value (see scope photo, Figure 4).VOUT typically changes from 0V to itsfinal regulated value in a little under5ms. This corresponds to an effectiveVOUT charging current of only 12.5mAfor a 10µF output capacitor (27.5mAfor 22µF, and so forth). This methodof controlling the average start-upcurrent prevents any nasty disrup-tions on the input supply both duringinitial power-up and when comingout of shutdown.

Current spikes due to normaloperation are mitigated by control-ling the effective output impedance ofthe regulator. As the VIN (or boostedVIN) to VOUT voltage differential grows,the effective output impedance (ROUT)of the charge pump is automaticallyincreased by internal voltage sensing

circuitry. This feature minimizes thecurrent spike pulled from VIN eachtime the switch network is enabledand helps to reduce output rippleover a wider VIN range.

Additional FeaturesThe LTC1515 family contains a power-on reset (POR) function. The POR pinis an open-drain output that pullslow when the output voltage is out ofregulation. This feature can be usedto prevent external circuitry fromoperating under invalid supply con-ditions. When VOUT rises to within6.5% of regulation, an internal timeris started, which releases POR (allowsthe pin to be pulled high) after 200ms(typical). In shutdown, the POR out-put is pulled low. In normal operation,an external pull-up resistor is usedbetween the POR pin and VOUT, asshown in Figure 1.

The LTC1514 contains an internallow-battery comparator and a refer-ence that are kept active in shutdown.The comparator-trip voltage is easilyprogrammed via an external resistordivider and has about 1% hysteresisfor stability. Since the low-batterycomparator is kept alive in shutdown,it may be used to protect batteriesagainst deep discharge by shuttingdown the power supply when the bat-tery voltage gets too low. It may alsobe used to implement a battery backupsupply if the main supply fails. Theopen-drain comparator output allowsfor flexible interfacing between theLBO output and external logic.

The LTC1514/LTC1515 family alsocomes equipped with thermal shut-down and can survive an indefiniteshort circuit to ground. The short-

1514_02.eps

MODE/ROUT CONTROL

SWITCH CONTROL

4

S1-S4

OSC

R1

R2

REF

C1 COUT

VOUT

S3

S4

S2

S1

ROUT

VIN

LTC1514

+

–

1514_04.eps

LTC1514-5

SHDN1

ON OFF

LBO2

LBI3

GND

VOUT

VIN

VOUT = 3.3V

VOUT = 5V

VIN = 2.7V TO 10V

C1+

C1–4

8

7

R1 47k

R4 10Ω

R3 750k, 1%

R2 402k, 1%

Q2

Q1

6

5C1 0.22µF

C4 10µF

C5 2.2nF

C3 22µF

C2 10µF

R5 220k

Q1 = TP0610T Q2 = MMBT3906LT1

+ ++

Figure 3. Simplified LTC1514/LTC1515 regulator block diagram

Figure 4. VOUT during power-up

Figure 5. Using the low-battery comparator as a feedback comparator to produce an auxiliary3.3V regulated output from the VOUT of the LTC1514-5 continued on page 24

COUT = 10µF

VOUT1V/DIV

1ms/DIVISION


DESIGN FEATURES

RS485 Transceivers Operate at10Mbps over Four Hundred Feet ofUnshielded Twisted Pair by Victor Fleury

IntroductionThe LTC1685/LTC1686/LTC1687family of RS485 transceivers canoperate at data rates of >40Mbps overone hundred feet of category 5unshielded twisted pair. These RS485transceivers employ a unique archi-tecture that guarantees excellentperformance over process and tem-perature variations, with propagationdelays for both the receiver and thedriver of 18.5ns ±3.5ns. The receiveremploys a fail-safe feature, over theentire 12V to –7V common moderange, whereby the receiver outputremains in a HIGH state when theinputs are left open or shorted to-gether. A novel short-circuit protectiontechnique permits indefinite shorts(to either driver or receiver output) topower or ground while sourcing/sink-ing a maximum of 50mA.

Circuit DescriptionThe timing performance of short chan-nel CMOS circuitry can typicallychange significantly over fabricationand temperature variations. This isdue in part to the large percentagevariation in MOS channel length andto second-order transistor gain andthreshold effects. For example, the

propagation delay of other transceiv-ers can vary by as much as 600% overprocess and temperature. In applica-tions where high speed clock anddata waveforms are sent over longdistances, propagation delay and skewuncertainties can pose system designconstraints and limit the maximumdata rate. The LTC1685/LTC1686/LTC1687 line of high speed RS485transceivers addresses this problemby guaranteeing the propagation tobe 18.5ns ±3.5ns. The propagationdelays change by ±20%, a better thantenfold improvement over other CMOStransceivers/receivers. Figure 1shows a block diagram of the receiverused in the LTC1685/LTC1686/LTC1687 transceivers. Figure 2 showsa block diagram of the driver used inthe transceivers. Note that the receiverand driver are both trimmed in orderto guarantee the tight timing require-ments. This is important because itminimizes the rise/fall skew of thereceiver and the skew between thetwo driver outputs. The input resistornetwork is set up to allow the com-mon mode to go as high as 12V and aslow as –7V with a 5V power supply.

Predictable PropagationDelay and Low SkewThe inherent temperature and pro-cess tolerance make it possible toguarantee a ±3.5ns propagation-delaywindow. Temperature stability is ac-complished by distributing the delayalong the signal chain so that half ofthe delay increases with temperatureand the other half decreases withtemperature, independent of theamount of delay trimming. These cir-cuits employ a novel current sourcewhose current increases with tem-perature. Delay trimming takes outsome of the effect of process varia-tions. Note that both the receiver anddriver also keep the input signal indifferential form as far down the sig-nal chain as possible. The differentialarchitecture allows for very tightreceiver and driver output skew.

High Data Rates overUnshielded Twisted PairThe LTC1685/LTC1686/LTC1687transceivers can have throughputssurpassing 40Mbps over one hun-dred feet of unshielded twisted pair(UTP).The tight propagation delayalong with the low skew make thesedevices well suited for high speed

–

++

–

4

–

+OUT

3

1685_01.eps

VCC

DIFF

BIAS BIASTRIM

GM OUTPUT

IN+

VCC

IN–

PROPTRIM

Figure 1. Receiver-section block diagram


DESIGN FEATURES

transmission over twisted-pair lines.Category 5 unshielded twisted paircable can be used to transmit highdata rates over long distances. TheEIA/TIA568A standard specifies aminimum performance for category 5

cable. The cable used in the followingexperiments was Belden 1588A 2-paircategory 5 UTP. Table 1 shows someof the performance characteristics ofthe cable.

The DC resistance of the cable willdivide down the signal at all frequen-cies. The longer the cable, the higherthe resistance and the larger the volt-age division. The AC attenuation ofthe cable will further divide down thesignal, with the highest frequencysignal components, of course, beingattenuated the most. Note that thecable impedance can vary by ±15%.Tweaking the termination of eachindividual cable with its actual im-pedance will yield best results;however, this might not be practical.

ExperimentsWe set up the LTC1685 transceiver tooperate at different speeds at differ-ent cable distances. Note that thecable had two distinct twisted-pairsets. Only one of the two pairs wasdriven; the other pair was kept in highimpedance or “listen” mode, when allstations connected to that particularpair are in receive mode. Even underthese circumstances (one pair beingdriven, while the adjacent pair is inhigh impedance mode), the receiversconnected to the high impedance cablemaintain a HIGH output state with-out glitching.

The timing of the receivers worksbest if they are being driven by a 50%duty cycle square wave. This tends tokeep a constant average voltage biason the cable and on the internal nodesof the devices. A more stringent test,however, is to try to pass a singlepulse at the highest data rate, thusnot allowing the system to reachsteady state. Figure 3 shows the testsetup, with four LTC1685 transceiv-ers: the LTC1685 on the top left is theonly transceiver with the driverenabled; the other three transceivers

–

+

3 4

1685_02.eps

GMDIFFIN

OUT

BUF

BUF

OUT

PROPTRIM BIASTRIM

1685_03.eps

100Ω

DE

RE

LTC1685

100Ω

RE

DE

2 PAIR CATEGORY 5 UTP

100Ω

DE

5V

RE 5V

LTC1685

100Ω

DE

RE

LTC1685 LTC1685

Figure 2. Driver-section block diagram

Figure 3. Test configuration

Table 1. Performance characteristics ofBelden 1588A 2-pair category 5 UTP

ecnadepmIegarevA 001 Ω ± %51

ecnatsiseRCDmumixaMC˚02ta 68.2 Ω '001/

mumixaMnoitaunettA zHM1 N '001/Bd16.0

zHM01 N '001/Bd79.1

zHM001 N '001/Bd76.6


DESIGN FEATURES

are set to receive mode only. All of thefollowing traces are actual scopephotographs.

One Hundred Feet, 40MbpsFigure 4 shows a 25ns pulse trans-mitted over one hundred feet oftwo-pair, category 5 twisted-paircable. The top trace is the input to thedriver at the left end of the cable. Thesecond trace is the driver output andthe third trace is the receiver input,which shows the attenuation of thepulse at the end of one hundred feetof cable. Figure 5 shows the sameconfiguration, but with a 40Mbpssquare wave as the input to the driver.

Four Hundred Feet, 10MbpsFigure 6 shows a 100ns pulse(10Mbps) propagated over four hun-dred feet of category 5 UTP. The pulsewidth at the far end of the cable isslightly narrower than the pulse widthat the driver output. Note the sharpedges on the receiver output, in spiteof the heavily filtered inputs due tocable losses.

Four Thousand Feet, 1MbpsFigure 7 shows a 1µs pulse propa-gated over four thousand feet ofcategory 5 UTP. The top trace is thedriver input. The 2nd trace is theoutput of an LTC1685 receiver, placedonly one hundred feet away from thedriver (not shown in diagram in Fig-ure 3). The third trace is the differentialinput to the transceiver at the end ofthe four thousand feet of UTP.

Notice the effect of the parasitic DCresistance of the cable. The third tracewaveform in this oscilloscope photo-graph was drawn at 1V/Div. Thismeans that the four thousand feetUTP parasitic resistance has dividedour signal by a factor of two (comparewith the third trace of Figure 6, whichis drawn at 2V/Div). Figure 8 shows a1Mbps square wave propagated downthe same four thousand feet of UTP.

50ns/DIV

2V/DIV

2V/DIV

2V/DIV

5V/DIV

Figure 4. 25ns pulse, 100 feet category 5 UTP

Figure 5. 40Mbps, 100 feet category 5 UTP

Figure 6. 100ns pulse, 400 feet category 5 UTP

Authors can be contacted at (408) 432-1900

50ns/DIV

2V/DIV

5V/DIV

DELAYFROM

100FT UTP

100ns/DIV

2V/DIV

2V/DIV

2V/DIV

5V/DIV

DELAYFROM

400FT UTP


DESIGN FEATURES

Undriven Cable PairThe cable used had two twisted-pairsets. One pair was driven, while theother pair was terminated at bothends but remained in high imped-ance. The undriven pair was tied tothe inputs of an LTC1685 receiver.The two inputs of the LTC1685 thusappeared “shorted” together through

1µs/DIV

Figure 7. 1µs pulse, 4000 feet category 5 UTP

Figure 8. 1Mbps, 4000 feet category 5 UTP

the terminated cable. This in turnactivated the fail-safe feature of thereceiver and the receiver outputremained high during all tests, de-spite the fact that the adjacent cablewas switching at high frequencies(short distance) and low frequencies(long distance).

Other Featuresof the LTC1685 Family Novel short circuit protection:

max 50mA without oscillating inand out of short-circuit mode,and automatically resetting whenshort is removed

Receiver output will go highwhen receiver inputs are eitherfloating or shorted

Three-state outputs High input resistance (>22K)

allows many devices on one line

ApplicationsThese devices can be used for highspeed transmission over twisted-paircables. The RS485 common moderange allows flexibility in connectingsystems with a ground potentialdifference or with power supply dif-ferences. They can be used in hubs,routers, bridges, repeaters, factory-floor controls and other applications.

ConclusionThe LTC1685/LTC1686/LTC1687transceivers can work over a widerange of speed and over a wide rangeof cable distances. The novel archi-tecture maintains a very tightpropagation delay window for boththe receiver and the driver. The pre-cise timing, ruggedness and fail-safefeatures make it easy to use in widevariety of applications.

for the latest information

on LTC products, visit

www.linear-tech.com

1µs/DIV

2V/DIV

5V/DIV

DRIVER INPUT

RECEIVER OUTPUT

2V/DIV

5V/DIV

1V/DIV

5V/DIV

DELAYFROM

4000FT UTP


DESIGN FEATURES

Hot Swapping the PCI Bus

The Peripheral Component Intercon-nect (PCI) bus has become widelyused in high volume personal com-puters and single-board computerdesigns. With a 32-bit data path anda bandwidth of up to 133MB/s, PCIoffers the throughput demanded bythe latest I/O and storage peripher-als. Unfortunately, the original PCIspecification does not require the busto be hot swappable, so the systempower must be turned off when aperipheral is inserted into or removedfrom a PCI slot.

With the migration of the PCI businto servers, industrial computers andcomputer-telephony systems, theability to plug a peripheral into a livePCI slot becomes mandatory. By usingthe LTC1421 to control the powersupplies, and QuickSwitch® QS3384sto buffer the data bus, a peripheralcan be inserted into a PCI slot withoutturning off the system power.

Inrush Currentand Data Bus ProblemsThe problems with plugging a stan-dard peripheral into a fully poweredPCI slot are shown in Figure 1. Whenthe peripheral is inserted, the supplybypass capacitors on the peripheralcan draw huge transient currents fromthe PCI power bus as they charge. The

transient currents can cause perma-nent damage to the connector pinsand board traces, and can causeglitches on the system supply thatforce other peripherals in the systemto reset.

The second problem involves thediodes to VCC at the inputs or outputsof most logic families. With theperipheral initially unpowered, theVCC input to the logic gate is at groundpotential. When the data bus pinsmake contact, the bus lines areclamped to ground through the diodesto VCC and the data is corrupted. Withcurrent flowing into the VCC diode,the logic gate may latch-up and de-stroy itself when power is applied.

Hot-Swappable PCI SlotUsing the LTC1421The circuitry for a hot-swappable PCIslot on the motherboard or backplaneis shown in Figure 2. The power sup-plies for each PCI slot are controlledby an LTC1421 and four externalFETs and the data bus is buffered byseveral QS3384 QuickSwitches orequivalent. A PCI power control ASIC,FPGA, microprocessor or the like con-trols all of the slots within the system.

The 12V, 5V, 3.3V and –12V sup-plies are controlled by placing externalN-channel pass transistors, Q1–Q4,

in the power path. By ramping thegate of the pass transistors at a con-trolled rate, the transient surgecurrent (I = C × dV/dt) drawn from thePCI supplies can be limited to a safevalue. The ramp rate for the positivesupplies is set by dV/dt = 20µA/C2.The –12V supply ramp rate is set byR7 and C3; resistor R5 and transistorQ5 help transistor Q2 turn off quickly.Resistors R9, R10 and R11 preventpotential high frequency FET oscilla-tions. Resistors R13 and R14 pull upPWRGD and FAULT to the properlogic level.

Sense resistors R1, R2 and R3 pro-vide current-fault protection. Whenthe voltage across R1 or R2 is greaterthan 50mV for more than 10ms, theLTC1421 circuit breaker is tripped.All of the FETs are immediately turnedoff and the FAULT pin is pulled low.The circuit breaker is reset by cyclingthe POR pin. The current-fault pro-tection for the 3.3V supply is providedby resistive divider R6 and R8 and theuncommitted comparator in theLTC1421. Because the current levelson the –12V supply are so low, over-current protection is not necessary.

The QuickSwitch contains a lowresistance N-channel FET placed inseries with the data bus. The switch isturned off when the board is insertedand then enabled after the power isstable. The switch inputs and out-puts do not have a parasitic diodeback to VCC and have very low capaci-tance.

System TimingThe system timing is shown in Figure3. The PCI power controller senseswhen a board has been inserted intothe PCI via the power-select bits.Alternatively, the user can inform thecontroller that a board has been

by James Herr, Paul Marshikand Robert Reay

I

V12V

PCI CONNECTOR

I

V5V

I

V3.3V

I

V–12V

GND

DN153 F1A

PCI CONNECTOR

DN152 F1B

DATA BUS

VCC

VCC

Figure 1. Problems with plugging a standard peripheral into a fully powered PCI slot

QuickSwitch is a registered trademark of Quality Semi-conductor Corp.


DESIGN FEATURES

+

–

+

*

*

*CONNECT PULLUP RESISTORS TO LOGIC SUPPLY

DI_HOTSWAP_02.eps

SETLO22

VCCLO23

CON11

AUXVCC24

FAULT4

DISABLE5

POR3

CON22

GND12

GATELO21

VOUTLO20

VCCHI19

SETHI18

GATEHI17

VOUTHI16

RAMP10

FB11

PWRGD6

RESET7

REF8

COMP– 14

COMP+ 13

COMPOUT15

CPON9C1

1µF 16V

R1, 0.005 5%, 1W

Q4 IRF7413

Q1 IRF7413

5V WITH 10A CIRCUIT BREAKER

12V WITH 3.3A CIRCUIT BREAKER

–12V WITH NO CIRCUIT BREAKER

GND

3.3V WITH 11.5A CIRCUIT BREAKER

Q3 1/2 IRF7101

R12, 10 5%, 1/16W

R2, 0.015 5%, 1/2W

R3 0.005

5%, 1W

R10 100k

5% 1/16W

R14 5.1k 5% 1/16W

FAULT

5V AT 5A

12V AT 500mA

3.3V AT 7.5A

R6 100 1% 1/16W

R8 5.62k 1% 1/16W

R5 20k

5%, 1/16W

R9 10 5% 1/16W Q2

1/2 IRF7101

C3 1µF 24V

R7 130k 5%, 1/16W

Q5 TP0610T

R4 30, 5% 1/16W

C2 0.22µF 24V

58mV

R11, 10

5%, 1/16W

R13, 5.1k 5%, 1/16W

ON/OFF

POWER GOOD

PCI POWER CONTROLLER

RST#

SELECT BITS

BUS ENABLE

DATA BUS

–12V AT 100mA

QuickSwitch®

LOGIC

RST#

PCI CONNECTOR

LTC1421

inserted via the front panel or key-board. The PCI controller holds theRST# pin low and disables theQuickSwitches, then turns on theLTC1421 via the POR pin. The powersupplies turn on at a controlled rateand when the 12V supply is within

10% of its final value, the PWRGDsignal pulls high. The PCI power con-troller waits one reset time-out period,then pulls RST# high and enables theQuickSwitches.

When the board is turned off, RST#is pulled low, the QuickSwitches are

disabled and the LTC1421 turned offby pulling the POR pin low. After a20ms delay, the external FETs areturned off and the supply voltagescollapse.

Figure 2. Hot-swappable PCI slot


DESIGN FEATURES

ConclusionUsing the LTC1421 and a Quick-Switch, a PCI slot can be made hotswappable so the system power canremain on when a peripheral is in-serted or removed. Up to now, thedesign of the Hot Swap circuitry hasrequired the talents of an analog guru,but with the LTC1421, safe hot-swapping becomes as easy as hookingup an IC, a couple of power FETsand a handful of resistors andcapacitors.

12V SUPPLY

3V SUPPLY5V SUPPLY

–12V SUPPLY

PWRGDPOR

12V SUPPLY

3V SUPPLY5V SUPPLY

–12V SUPPLY

PWRGDPOR

ConclusionThe LTC1605 is a complete 16-bitADC with a built-in sample-and-holdand reference. Its wide analog inputrange and DC accuracy make it agood candidate for industrial process-

+

+

1605_09b.eps

2.2µF

2.2µF

VIN

AGND 1

REF

CAP

AGND 2

1

2

3

4

5

INPUT ±10V

RA 200Ω

RB 33.2k

+

+

1605_09a.eps

2.2µF

2.2µFOFFSET 50k

GAIN 50k

5VVIN

AGND 1

REF

CAP

AGND 2

1

2

3

4

5

INPUT ±10V

RA 200Ω

576kRB

33.2k

Figure 9a. Gain and offset errors can be reduced to zero by adding trimming resistors. Figure 9b. If the specified gain and offseterrors are adequate, connect the externalresistors as shown.

control applications. The LTC1605 isthe first of many new 16-bit ADCsthat will be introduced as Linear Tech-nology continues to broaden its dataacquisition product line. Having a

selection of ADCs with 8, 10, 12, 14and now 16-bits of resolution willmake it easier for users to find theright ADC from Linear Technology fortheir applications.


Time to Upgrade?The new, low cost LTC1419 is theideal converter to upgrade new 12-bit, high performance designs to 14bits. Its exceptional dynamic perfor-mance gives a 10dB improvement in

dynamic range compared to a thebest 12-bit devices. Its low power andflexibility make it useful in a variety oftime- and frequency-domain applica-tions. This and the LTC1419’s low

cost and ultrasmall size make it theideal candidate for designerswho need the next step in ADCperformance.



Figure 3a. System timing: power up

Figure 3b. System timing: power down


DESIGN FEATURES

circuit protection not only preventsthe part from blowing up, but alsolimits the current pulled from theinput supply during a fault condition.When VOUT is held below 100mV by ashort on the output, a 15mA currentlimit in the regulator output kicks inuntil the short goes away.

Dual Output Supplyfrom a 2.7V to 10V InputThe circuit shown in Figure 5 usesthe low-battery comparator as a feed-back comparator to produce anauxiliary 3.3V regulated output fromthe VOUT of the LTC1514-5. A feed-back voltage divider formed by R2and R3 connected to the comparator

input (LBI) establishes the outputvoltage. The output of the comparator(LBO) enables the current sourceformed by Q1, Q2, R1 and R4. Whenthe LBO pin is low, Q1 is turned on,allowing current to charge outputcapacitor C4. Local feedback formedby R4, Q1 and Q2 creates a constantcurrent source from the 5V output toC4. Peak charging current is set byR4 and the VBE of Q2, which alsoprovides current limiting in the caseof an output short to ground. R5 pullsthe gate of Q1 high when the auxiliaryoutput is in regulation. C5 is used toreduce output ripple. The combinedoutput current from the 5V and 3.3Vsupplies is limited to 50mA. Since theregulator implements a hysteretic

feedback loop in place of the tradi-tional linear feedback loop, nocompensation is needed for loop sta-bility. Furthermore, the high gain ofthe comparator provides excellent loadregulation and transient response.

ConclusionWith low operating current, lowexternal parts count and robust pro-tection features, the LTC1514 andLTC1515 are well-suited to low powerstep-up/step-down DC/DC conver-sion. The shutdown, POR andlow-battery detect features provideadditional value and functionality.The simplicity and versatility of theseparts make them ideal for low powerDC/DC conversion applications.

level. A capacitor from boost to switchis still required, because this capaci-tor supplies the gate-charge currents.

The basic step-up converter isshown in Figure 8. The LTC1624 isused to create 12V/1A from a 5Vsource with the efficiency shown inFigure 9. Efficiency is above 90%from 20mA up to close to full load,dropping only to 89% at 1A.

In order to allow input voltagesboth above and below the output volt-

age, a SEPIC converter can be used.An example of the LTC1624 used as a12V/0.5A SEPIC converter operatingfrom an input range of 5V to 20V isshown in Figure 10.

ConclusionThe LTC1624 is the latest member ofLinear Technology’s family of con-stant-frequency, N-channel, highefficiency controllers. With only 8 pins,an internal boost diode and the abil-

ity to operate in multiple topologies, itcan be used to implement a widevariety of different applications in avery small amount of space. The highperformance of this controller, withits wide input range, 1% referenceand tight load regulation, makes itideal for next generation designs.


LTC1514/LTC1515, continued from page 16

–

+

–

+

1495_08.eps

RSENSE0.1Ω

ILCHARGE

RA

Q1 2N3904

CHARGEOUT

DISCHARGEOUT

DISCHARGE

Q2 2N3904

RA

RA RA

RBRB

A11/2 LT1495

5V

A21/2 LT1495

V IRR

R

FOR R kR kVI

VA

O LB

ASENSE

A

B

O

L

=

==

=

110

1

12V

Battery-Current Monitor with“Over-the-Top” OperationThe bidirectional current sensorshown in Figure 8 takes advantage ofthe extended common mode range ofthe LT1495 to sense currents intoand out of a 12V battery while oper-ating from a 5V supply. During thecharge cycle, op amp A1 controls thecurrent in Q1 so that the voltage dropacross RA is equal to IL × RSENSE. Thisvoltage is then amplified at the chargeoutput by the ratio of RA to RB. Duringthis cycle, amplifier A2 sees a nega-tive offset, which keeps Q2 off and thedischarge output low. During the dis-charge cycle, A2 and Q2 are activeand operation is similar to that dur-ing the charge cycle.

Figure 8. Battery-current monitor

ConclusionsThe LT1495/LT1496 extends LinearTechnology’s range of rail-to-rail am-plifier solutions to a truly micropowerlevel. The combination of extremely

low current and precision specifica-tions provides designers with aversatile solution for battery-oper-ated devices and other low powersystems.

LT1495, continued from page 10


DESIGN IDEAS

–48V to 5V DC/DC ConverterOperates from the Telephone Line

by Gary ShockeyDC/DC converters for use inside

the telephone handset require opera-tion from the high source-impedancephone line. Additionally, the CCITTspecifications call for on-hook powerconsumption of 25mW maximum. TheDC/DC converter circuit presentedhere is 70% efficient at an inputpower of 25mW, providing 5V at3.4mA. Controlled, low peak switchcurrent ensures that the –48V inputline does not experience excessivevoltage drops during switching.

The circuit shown in Figure 1operates as a flyback regulator withan auxiliary winding to provide powerfor the LT1316. To understand theoperation of this circuit, examine Fig-ure 1. When power is first applied,the LBI pin is low, causing the SHDNpin to be grounded through LBO.This places the part in shutdownmode and only the low-battery com-parator remains active. During thisstate, VIN rises at a rate determinedby R1 and C1. The LT1316 draws

+

+

DI_48-5_01.eps

LT1316

LBI

LB0 FB

SHDN7

1

2

R3 604k 1%

Q3 2N3904

– 48V

R2 1.30M 1%

R5 69.8k 1%

R6 121k 1%

R7 432k, 1%

Q2 MPSA928

3 4

6R4 2M

R1 1.3M

C1 0.1µF

C2 0.022µF

C4 47µF

D2 1N4148

C3 47µF

D1 1N5817T1

10:1:1

L3L1

VOUT 5V

5

Q1D3

1N4148

VIN SW

RSET GND

T1 =DALE LPE-4841-A313, L PRI = 2mH Q1 =ZETEX ZVN 4424A R6, Q2 AND R7 MUST BE PLACED NEXT TO THE FB PIN

L2VA

only 6µA in shutdown mode; R1 needsto supply only this current, the cur-rent through R2 and R4, and C1’scharging current. When LBI reaches1.17V (VIN ≈ 3.7V) the LBO pin lets goof SHDN and the part enters the activemode. Once this state is reached,switching action begins and the out-put voltage begins to increase. As thedevice switches, the LT1316 VIN pindraws current out of C1; VIN then

50µS/DIV

Figure 1. –48V to 5V flyback converter

Figure 2. Switch voltage and current waveforms Figure 3. Output ripple voltage and current waveforms


DESIGN IDEAS–48V to 5V DC/DC ConverterOperates from the Telephone Line................................................25Gary Shockey

Water Tank Pressure Sensing,a Fluid Solution ..................... 26Richard Markell

0.05µV/˚C Chopped AmplifierRequires Only 5µA Supply Current................................................28Jim Williams

Making –5V 14-Bit Quiet ..........29Kevin R. Hoskins

VOUTAC COUPLED

100mV/DIV

SECONDARYCURRENT

200mA/DIV

PRIMARYCURRENT50mA/DIV

SWITCH PINVOLTAGE10V/DIV

SECONDARYCURRENT

200mA/DIV

PRIMARYCURRENT50mA/DIV

1µS/DIV


DESIGN IDEAS

Water Tank Pressure Sensing,a Fluid Solution by Richard Markell

IntroductionLiquid sensors require a media com-patible, solid state pressure sensor.The pressure range of the sensor isdependent on the height of the col-umn or tank of fluid that must besensed. This article describes the useof the E G & G IC Sensors Model 90stainless steel diaphragm, 0 to 15psigsensor used to sense water height ina tank or column.

Because large chemical or watertanks are typically located outside in“tank farms,” it is insufficient to pro-vide only an analog interface to adigitization system for level sensing.This is because the very long wiresrequired to interconnect the systemcause IR drops, noise and other cor-ruption of the analog signal. Thesolution to this problem is to imple-ment a system that converts theanalog to digital signals at the sensor.In this application, we implement a“liquid height to frequency converter.”

Circuit DescriptionFigure 1 shows the analog front-endof the system, which includes theLT1121 linear regulator for powering

the system. The LT1121 is a micro-power, low dropout linear regulatorwith shutdown. For micropowerapplications of this or other circuits,the ability to shut down the entiresystem via a single power supply pinallows the system to operate onlywhen taking data (perhaps everyhour), conserving power and improv-ing battery life.

In Figure 1, U3, the LT1121, con-verts 12V to 9V to power the system.The 12V may be obtained from a wallcube or batteries.

The LT1034, a 1.2V reference, isused with U1D, 1/4 of an LT1079quad low power op amp, to provide a1.5mA current source to the pressuresensor. The reference voltage is alsodivided down by R5, R8, R4 and the10k potentiometer and used to offsetthe output amplifier, U2A, so that thesignals are not too close to the supplyrails.

Op amps U1A and U1B (each 1/4of an LT1079) amplify the bridge pres-sure sensor’s output and provide adifferential signal to U2A (an LT1490).Note that U2A must be a rail-to-rail

op amp. The system’s analog outputis taken from U2A’s output.

Figure 3 plots the output voltagefor the sensor system’s analog frontend versus the height of the watercolumn that impinges on the pres-sure transducer. Note that thepressure change is independent ofdiameter of the water column, so thata tank of liquid would produce thesame resulting output voltage. Figure4 is a photograph of our test setup.

The remainder of the circuitry,shown in Figure 2, allows transmis-sion of analog data over long distances.The circuit was designed by Jim Wil-liams. The circuit takes a DC inputfrom 0V to 5V and converts it to afrequency. For the pressure circuit inFigure 1, this translates to approxi-mately 0Hz to 5kHz.

The voltage-to-frequency convertershown in Figure 2 has very low powerconsumption (26µA), 0.02 % linear-ity, 60ppm/˚C drift and 40ppm/Vpower supply rejection.

In operation, C1 switches a chargepump, comprising Q5, Q6 and the100pF capacitor, to maintain its nega-tive input at 0V. The LT1004s and

–

+–

+

–

+

–

+

DI_WT_01.eps

OUT

OUT

IN

SHDN

8+12

C1 0.1µF C2

1µF

C3 0.1µF

12 4

11

GND

U1D LT1079

14

13

5

U1B LT1079

7

6

2

U1A LT1079

1

3

5

1 9V

TO FIGURE 2 (9V)

2

R1 13k

5k INSIDE SENSOR IN MODEL 93 REPLACES R13 AND 10k POT MODEL 90/MODEL 93 E G & G IC SENSORS (408) 432-1800

R2 18k

R3 35.7k

LT1034

-1.2R5 4.99k

R4 4.99k

10k POT

R14 100k

R13 3.32k

R21 100k

R16 100k

R20 100k

10k POT

R15 100k

R18 249k

R17 100k

R19 249k

R8 3.01k

R6 823Ω

U3 LT1121

GND3

PRESSURE SENSOR

MODEL 902 7

3 6

51 2

U2A LT1490

1

VO3

Figure 1. Pressure-sensor amplifier


DESIGN IDEAS

+

+

–

10kHz TRIM 200k

C1 1/2 LTC1441

+

–C2

1/2 LTC1441

1.2M*

15k

10M

100Hz TRIM 3M TYP

100pF†

INPUT FROM

PRESSURE SENSOR

AMPLIFER (FIG 1) 0.01

100k

= HP5082-2810 OR 1N5711

= 1N4148

= 2N5089 = 2N2222 = POLYSTYRENE = 1% METAL FILM

Q1, Q2, Q8 ALL OTHER

† *

6.04k*

LM334

2.7M

100k

0.1

50pF

FROM LT1121 (FIGURE 1) +9V

74C14

2.2µF

+0.47

0.1

LT1004 1.2V x 3

Q7Q5

Q1

Q6

Q4

Q8

Q3

Q2

GROUND ALL UNUSED 74C14 AND 74C74 INPUTS PINS

DI_WT_02.eps

CLK

D

QVCC

74C74

CLR

7

Q6

5 390k

2

3

14 14

PRE

associated components form a tem-perature-compensated reference forthe charge pump. The 100pF capaci-tor charges to a fixed voltage; hence,the repetition rate is the circuit’s onlydegree of freedom to maintain feed-back. Comparator C1 pumps uniformpackets of charge to its negative inputat a repetition rate precisely propor-tional to the input-voltage-derivedcurrent. This action ensures that cir-cuit output frequency is determinedstrictly and solely by the input voltage.

Figure 5 shows the output fre-quency versus column height for two

FEET

0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

VOLT

S

16

DI_WT_03.eps

0 2 4 6 8 10 12 14FEET

0

1000

2000

3000

4000

5000

6000

FREQ

UENC

Y (H

z)

16

DI_WT_05.eps

0 2 4 6 8 10 12 14

SENSOR #2

SENSOR #1

different Model 90 transducers. Notethe straight lines, which are repre-sentative of excellent linearity.

ConclusionA cost effective system is shown hereconsisting of a fluid pressure sensor,IC Sensors Model 90. This sensor’soutput is fed to signal processingelectronics that convert the low levelDC output of the bridge-based pres-sure sensor to a frequency in theaudio range depending on the heightof the fluid column impinging on thepressure transducer.

Figure 2. This 0.02% V/F converter requires only 26µA supply current.

Figure 5. Output frequency vs column heightfor two Model 90 sensorsFigure 3. Output voltage vs column height Figure 4. Test setup for water-column sensor


DESIGN IDEAS

–

+–

+

–

+–

+LTC1440

LTC1440

10M

10M

10M

0.047µF

Ø2

Ø1

5V

–5V

LT1495 LT1495

1/2 CD4016 1/2 CD4016

1µF

1µF

CCOMP 0.1µF

1M

1M

1M

10M

10k

10M

10k

1

2

3

4

5

13

Ø2

Ø1

Ø1

Ø2

1112

10

9

6

8

A1A A1B

–5V

5V

OUT

IN

C1A

C1B

0.05µV/˚C Chopped AmplifierRequires Only 5µA Supply Current

by Jim Williams

Figure 1 shows a chopped ampli-fier that requires only 5.5µA supplycurrent. Offset Voltage is 5µV, with0.05µV/˚C drift. A gain exceeding 108

affords high accuracy, even at largeclosed-loop gains.

The micropower comparators (C1Aand C1B) form a biphase 5Hz clock.The clock drives the input-relatedswitches, causing an amplitude-modulated version of the DC input toappear at A1A’s input. AC-coupledA1A takes a gain of 1000, presentingits output to a switched demodulatorsimilar to the aforementionedmodulator.

The demodulator output, a recon-structed, DC-amplified version of thecircuit’s input, is fed to A1B, a DCgain stage. A1B’s output is fed back,via gain setting resistors, to the inputmodulator, closing a feedback looparound the entire amplifier. Theconfiguration’s DC gain is set by thefeedback resistor’s ratio, in this case1000.

The circuit’s internal AC couplingprevents A1’s DC characteristics frominfluencing overall DC performance,

accounting for the extremely low off-set uncertainty noted. The highopen-loop gain permits 10ppm gainaccuracy at a closed-loop gain of 1000.

The desired micropower operationand A1’s bandwidth dictate the 5Hzclock rate. As such, the resultantoverall bandwidth is low. Full-powerbandwidth is 0.05Hz with a slew rateof about 1V/s. Clock-related noise,about 5µ V, can be reduced byincreasing CCOMP, with commensu-rate bandwidth reduction.

Figure 1. 0.05µV/˚C chopped amplifier requires only 5µA supply current



www.linear-tech.com


DESIGN IDEAS

Making –5V 14-Bit Quiet by Kevin R. Hoskins

Many high performance dataacquisition systems reap multiplebenefits when using ±5V suppliesrather than a single 5V supply. Thesebenefits include the ability to handlelarger signal magnitudes than is pos-sible with a single 5V supply. Thisincreases a system’s dynamic rangeand helps improve the signal-to-noiseratio. Operating on ±5V also increasesheadroom, which is important for sig-nal conditioning. Compared tooperating on 5V, conditioning cir-cuitry operating on ±5V has twice theheadroom, allowing it to easily handle±2.5V signals without clipping. Addi-tionally, the greater headroom avoidsthe limitations of rail-to-rail opera-tion and widens the selection of highperformance operational amplifiersand analog-to-digital converters, suchas the LTC1419.

Although a switching or charge-pump power supply is an efficientway to create a –5V supply from asingle 5V supply, they are not gener-ally recommended for use with ADCs.Typical ADCs have inadequate PSRR,which decreases with increasing fre-quency. This poor PSRR performance

cannot sufficiently attenuate thenoise created by switching or charge-pump supplies. However, LTC’s newfamily of ADCs, here represented bythe LTC1419, has excellent PSRR.This family make it easy to achievehigh performance data conversion,even at 14 bits, using a switch-basedregulator for a –5V supply.

The LTC1419’s high PSRR is shownin Figure 1. It shows that when oper-ating on ±5V, the negative and positivePSRR are typically 80dB and 90dB,respectively, up to 200kHz for a 100mVripple voltage. Combined with properlayout, the LTC1419’s high PSRRallows it to convert signals withoutsignal degradation while using switch-ing regulators and charge pumps togenerate its –5V supply. Applicationsincluding high speed communi-cations, high resolution signalprocessing and wideband multiplex-ing benefit from the LTC1419’sadvantages—its 20MHz S/H band-width, 800ksps conversion rate and14-bit resolution. This Design Ideashows two supply designs that arequiet enough to use with the LTC1419.

Low Noise Cuk ConverterThe switching regulator shown in Fig-ure 1 is configured as a Cuk converter,creating –5V from 5V. This configura-tion has the advantage of a smalltriangular switching-current wave-form through the secondary inductor.This current waveform is continu-ous, producing much less harmoniccontent than is created by a typicalpositive-to-negative voltage converter,with its rectangular switching cur-

RIPPLE FREQUENCY (Hz)

AMPL

ITUD

E OF

POW

ER S

UPPL

Y FE

EDTH

ROUG

H (d

B)

0

–20

–40

–60

–80

–100

–1201k 100k 1M 10M

1410 G08

10k

VRIPPLE = 0.1V

VSS

VDD

DGND

+

+

+

+

DI_1419_01.eps

AVDD28

+AIN1

DVDD27

–AIN2

VSS26

VREF3

BUSY25

COMP4

CS24

C71µF CER

C5

C8 22µF 10V

TANT

C6

5V

ANALOG INPUT

U1 LTC1419

AGND5

CONVST23

D13 (MSB)6

RD 22

MICROPROCESSOR/ MICROCONTROLLER INTERFACED12

7

SHDN21

D118

D020

D109

D119

D910

D218

D811

D317

D712

D416

D613

D515

DGND14

VSW

1

2

4

3

L1

VIN5

NFB3

8

S/S4

GND7

VC

C9 0.01µF

D1

R3 4.99k

R5 4.99k 1%

R6 499Ω 1%

R4 4.99k 1%

C11 100µF

10V TANT

C12 0.1µF

1GND S

6

U2 LT1373

C10 10µF CER

CUK* CONVERTER

C5, C6, C7 =10 µF CERAMIC L1 =OCTAPAC CTX-100-1 D1 =1N5818 *PATENTS MAY APPLY

Figure 1. The LTC1419’s positive supplyPSRR of 80dB and negative supply PSRR of90dB to 200kHz is a significant contributionto this ADC’s wideband conversionperformance and 80dB SINAD.

Figure 2. The LTC1419’s 80dB PSRR allows the LTC1373 to generate the –5V and power the ADC without signal-conversion degradation.


DESIGN IDEAS

+

+

DI_1419_02.eps

281

272

263

BUSY25

COMP4

CS24

C71µF CER

C5C1

10µF TANT

C6

5V

ANALOG INPUT

U2 LTC1419

AGND5

CONVST23

D13 (MSB)6

RD 22

MICROPROCESSOR/ MICROCONTROLLER INTERFACED12

7

SHDN21

D118

D020

D109

D119

D910

D218

D811

D317

D712

D416

D613

D515

DGND14

V+FB/SHDN1

VREF6

OSC7

8

CAP+2

GND3

VOUT5

CAP–4

U1 LT1054

R2, 120k

R1, 30.1k C3 0.002µF

C4 100µF TANT

C2 2µF

C5, C6, C7 =10 µF CERAMIC

AVDD+AIN

DVDD–AIN

VSSVREF

rent waveform. With the componentsshown, the LT1373 operates continu-ously with load currents above 10mA.Because the LTC1419s typically draw18mA of negative supply current, theLT1373 will always operate in thequiet continuous mode.

RegulatedCharge Pump ConverterThe LTC1419’s negative PSRR alsoallows the use of charge pumps tocreate –5V. The circuit shown in Fig-ure 3 uses the LT1054 regulatedcharge pump. This circuit has theadvantage of reduced board space,since it lacks an inductor and requiresfewer passive components.

Performance ResultsWhat is the effect of using either ofthese switch-based supplies on theLTC1419’s conversion performance?The FFTs in Figures 4–6 show theexcellent results. Figure 4 is an FFT ofa typical LTC1419 operating on ±5Vfrom a lab supply and converting afull-scale 91kHz sine wave at 800ksps.The noise floor is approximately114dB below full scale, the secondharmonic’s amplitude is approxi-mately 90dB below full scale and the

FREQUENCY (kHz)

–160

–80

–100

–120

–140

0

–20

–40

–60

AMPL

ITUD

E (d

B)

400

DI_1419_03.eps

0 50 100 150 200 250 300 350

LTC1419 ±5V LAB SUPPLIES f SAMPLE =800kHz f IN =91kHz S/N =80.5dB

FREQUENCY (kHz)

–160

–80

–100

–120

–140

0

–20

–40

–60

AMPL

ITUD

E (d

B)

400

DI_1419_04.eps

0 50 100 150 200 250 300 350

LTC1419 5V LAB SUPPLY –5V LT1373fSAMPLE =800kHz f IN =91kHz S/N =80.5dB

Figure 3. The LTC1419’s high negative supply PSRR also allows the use of the LT1054 regulated charge pump to generate –5V without loss ofperformance.

Figure 4. This FFT of an LTC1419 powered by a ±5V lab supply shows a SINAD of 80.5dB for a91kHz input at a 800ksps sampling rate.

Figure 5. When the LTC1419’s –5V supply is generated by an LT1373 switching regulator, theSINAD (80.5dB), the noise floor and the 91kHz fundamental’s harmonic components remainessentially the same as those shown in Figure 4.


DESIGN IDEAS

SINAD is 80.5dB. Figure 5 shows theFFT of the same LTC1419 operatingon a 5V lab supply and –5V from theLT1373 circuit. The noise floor andthe second harmonic’s amplituderemain the same relative to full scaleand the SINAD remains the same at80.5dB. Figure 6 shows the LTC1419’sresponse when its –5V is generatedby the LT1054 circuit. As with theLT1373 circuit, the noise floor andthe amplitude of harmonics remainthe same and the SINAD is 80.8dB.

FREQUENCY (kHz)

–160

–80

–100

–120

–140

0

–20

–40

–60

AMPL

ITUD

E (d

B)

400

DI_1419_05.eps

0 50 100 150 200 250 300 350

LTC1419 5V LAB SUPPLY –5V LT1054 f SAMPLE =800kHz f IN =91kHz S/N =80.8dB

Figure 6. When the LTC1419’s –5V supply is generated by an LT1054 inverter, the SINAD(80.8dB), the noise floor and the 91kHz fundamental’s harmonic components again remainunchanged from those shown in Figure 4.

the LT1316 from VA rather than fromthe –48V rail, increasing efficiency.VOUT must not be loaded until itreaches 5V or the circuit will notstart.

During each switch cycle, currentin the transformer primary ramps upuntil current limit is reached (SeeFigures 2 and 3). This peak switchcurrent can be set by adjusting R5.The circuit shown uses a 69.8kΩresistor to give a peak switch currentof 50mA. Increasing R5 decreases thecurrent limit. Secondary peak cur-rent will be approximately equal tothe primary peak current multipliedby the transformer turns ratio. TheFB pin has a sense voltage of 1.23Vand VOUT can be set by the followingformula:

LOAD CURRENT (mA)

40

70

60

50

80

90

EFFI

CIEN

CY (%

)

100

DI_48-5_04.eps

1 10

VIN = 48V VIN = 36V

VIN = 72V

POWER OUT (mW)

0

0.1

0.2

0.3

INPU

T CU

RREN

T (m

A)5

DI_48-5_05.eps

0 1 2 3 4

VIN = 48V

VIN = 72V

VIN = 36V

Figure 4. Efficiency vs load current Figure 5. Input current vs power out

–48V, continued from page 25

decreases sufficiently to trip the low-battery detector, stopping theswitching. Start-up proceeds in thisirregular fashion until, eventually,the voltage at VA increases to 5V. (VAis the same as VOUT, because L2 andL3 have the same number of turns.)After start-up, current is supplied to

Efficiency versus load current isdetailed in Figure 4. Note that for therange of 4mA to 80mA, 70% efficiencyor greater is achieved. Figure 5 showsinput current versus output power.Less than 80µA quiescent currentflows when the converter supplies0.5mW over the 36V–72V range.

VOUT = 1.23(R7/R6) + 0.6V.



www.linear-tech.com


DESIGN INFORMATION

Understanding and ApplyingVoltage References: Part One

by Mitchell Lee

Specifying the right reference andapplying it correctly is a more difficulttask than one might first surmise,considering that references are only2- or 3-terminal devices. Althoughthe word “accuracy” is most oftenspoken in reference to references, it isdangerous to use this word too freelybecause it can mean different thingsto different people. Even more per-plexing is the fact that a referenceclassified as a dog in one applicationis a panacea in another. This articlewill familiarize the reader with thevarious aspects of reference “accu-racy” and present some tips onextracting maximum performancefrom any reference.

As with other specialized electronicfields, the field of monolithic refer-ences has its own vocabulary. We’vealready learned the first word in ourreference vocabulary, “accuracy.” Thisis the yardstick with which refer-ences are graded and compared.Unfortunately, there are at least fiveor six good units for gauging accu-racy. To keep you from reaching a fullunderstanding of the topic, industrypundits use a special technique called“unit-hopping” to confuse and con-found everyone from newcomer toseasoned veteran. You mention anaccuracy figure and the pundit quicklyhops to a new unit so that you cannotfollow his line of reasoning. Figure 1neutralizes the pundits’ callous in-tentions and allows its possessor tounit-hop with equal ease and fullcomprehension. Refer to Figure 1 asyou read this article.

Today’s IC reference technology isdivided along two lines: bandgapreferences, which balance thetemperature coefficient of a forward-biased diode junction against that ofa ∆VBE (see sidebar on page 33); andburied Zeners, which use subsurfacebreakdown to achieve outstandinglong-term stability and low noise. Withfew exceptions, both reference types

use additional on-chip circuitry tofurther minimize temperature driftand trim output voltage to an exactvalue. Bandgap references are gener-ally used in systems of up to 12 bits;buried Zeners take over from there inhigher accuracy systems.

In circuits and systems, monolithicreferences face competition from dis-crete Zener diodes and 3-terminalvoltage regulators only where accu-racy is not a concern. 5% Zeners and3% voltage regulators are common-place; these represent 4- or 5-bitaccuracy. At the other end of thespectrum—laboratory standards—theperformance of the best monolithicreferences is exceeded only by satu-rated Weston cells and Josephsonarrays, leaving monolithic referencesin command of every conceivable cir-cuit and system application.

Reference accuracy comprisesmultiple electrical specifications.These are summarized in Table 1.Most commonly specified by circuitdesigners is initial accuracy. This is ameasure of the output voltage errorexpressed in percent or in volts. Ini-tial accuracy is specified at roomtemperature (25°C), with a fixed in-put voltage and zero load current, orfor shunt references, a fixed biascurrent.

Tight initial accuracy is a concernin systems where calibration is eitherinconvenient or impossible. More com-monly, absolute accuracy is only asecondary concern, as a final trim isperformed on the finished product toreconcile the summation of all sys-tem inaccuracies. A final trim affectsconsiderable cost savings by elimi-nating the need for tight initialaccuracy in every reference, DAC,ADC, amplifier and transducer in thesystem.

Monolithic reference initial accu-racy ranges from 0.02% to 1%,representing 1LSB error in 6-bit to12-bit systems. Weston cells and

1

2

3

4

5

6

7

8

9

10

11

12

13

14

15

16

17

18

19

20

21

22

23

24

2

4

8

16

32

64

128

256

512

1024

2048

4096

8192

16,384

32,768

65,536

131,072

262,144

524,288

1,048,576

2,097,152

4,194,304

8,388,608

16,777,216

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

–110

–120

–130

–140

50

30

20

10

5

3

2

1

0.5

0.3

0.2

0.1

0.05

0.03

0.02

0.01

0.005

0.003

0.002

0.001

0.0005

0.0003

0.0002

0.0001

–1

–2

–3

–4

–5

–6

–7

10,000

5,000

3,000

2,000

1,000

500

300

200

10

5

3

2

1

0.5

0.3

100

50

30

20

0.2

0.1

6

5

4

3

2*

COUNTS dB

PERCENT

POWERS OF TEN

PPM

DVM DIGITS

BITS

1 – 2

1 – 2

1 – 2

1 – 2

*HP MIRRORED SCALE

Figure 1. Accuracy translator


DESIGN INFORMATION

∆VBE: Integrated CircuitWorkhorseIt is, perhaps, a cruel fate for ICdesigners that no single IC device orstructure is invariant with changesin temperature. Various combina-tions of devices have been devised tostabilize circuits against changes intemperature. As explained in thetext, Zener-based references use aZener and a forward-biased diodeconnected in series to achieve near-zero temperature coefficient, and abandgap relies on a ∆VBE in serieswith a forward-biased diode.

An indispensable technique in in-tegrated circuit design, the ∆VBE isnot widely known in other fields.Before explaining the theory of ∆VBE,let’s skip ahead to the two mostimportant results: two identicaldiode (or base-emitter) junctionsrunning different currents producedifferent voltage drops. The ratio ofthe currents controls the absolutevalue of the offset voltage. Further,this offset has a predictable, posi-tive temperature coefficient ofapproximately 3.4µV/°C for eachroom-temperature millivolt of off-set. By combining the positive TC ofa ∆VBE with the negative TC of adiode drop, a zero TC bandgap refer-ence is formed. As we shall soon see,it takes a ∆VBE offset of 650mV tocancel the –2.18mV/°C TC of ahypothetical diode*.

Two transistors (or diodes) pro-duce an offset given by the followingequation:

∆VBE = VBE1 – VBE2= (kT/q) ln(Je1/Je2) (1)

where ∆VBE = of fset voltage,k = Boltzmann’s constant (1.381 ×10–23 Joules/K), T = absolute tem-

perature (298K at room), q = charge ofan electron (1.6 × 10–19 Coulombs),and Je = emitter current density. Theactual units of area used to calculateJe1 and Je2 cancel each other, sothat only the area ratio is important.Similarly, only the current ratio isimportant. If we restrict ourselves tousing two identical transistors, Equa-tion (1) reduces to

∆VBE = VBE1 – VBE2= (kT/q) ln(IC1/IC2) (2)

where IC = collector current (see Fig-ure A). The temperature coefficient isgiven by

TC = d∆VBE/dT= (k/q) ln(IC1/IC2) (3)

where k/q = 86.3µV/°C.Calculating the current ratio

required to produce +2.18mV/°C (cor-responding to 650mV offset) we

find that it is unmanageably large,about 9.44 × 1010:1. In practice, amuch smaller offset is generated bya ∆VBE cell and then amplified to650mV. As an example, see FigureB. Using a 10:1 current ratio**, wefind a room temperature offset fromEquation (2) of 59.2mV, and a tem-perature coefficient of 199µV/°C.Applying a gain of slightly less thaneleven brings us to 650mV and+2.18mV/°C.

Adding a PNP emitter follower tothe output of this circuit forms acrude “bandgap” reference, with anoutput voltage equal to the sum of650mV and the PNP’s VBE. Assum-ing VBE = 600mV, the output wouldbe 1.25V. The reference could befurther improved by trimming thegain of eleven so that the ∆VBE exactlycanceled the PNP’s base-emitter tem-perature coefficient. IC bandgapreferences are constructed in a simi-lar way.

–

+

∆VBE

I 10I

59.2mV

AV = 11650mV

+ 2.18mV/˚C

VBE 600mV

–2.18mV/˚C

1.25V 0mV/˚C

Figure B. A bandgap reference is formed by stacking a ∆VBE generator and a VBE.

VBE2VBE1

IC1 IC2

Figure A. The current ratio required toproduce a certain VBE offset is definedby equations (1) and (2).

Josephson arrays clock in at 1ppm–10ppm and 0.02ppm initial accuracy,respectively (0.02ppm is less than1LSB error in a 25-bit system).

Temperature-induced changes inreference output voltage can quicklyovershadow a tight initial accuracyspecification. Considerable effort istherefore expended to minimize the

temperature coefficient (tempco) of areference. Most references are guar-anteed in the range of 2ppm/°C to40ppm/°C, with a few devices fallingoutside this range. A properly appliedLTZ1000 temperature stabilized ref-erence can demonstrate 0.05ppm/°C.

Tempco is specified as an averageover the operating temperature range

in units of ppm/°C or mV/°C. Thisaverage is calculated in what is calledthe “box” method. Figure 2 showshow box method tempco figures aredefined and calculated. The referencein question (LT1019 bandgap) is testedover the specified operating tempera-ture range. The minimum andmaximum recorded output voltages

*The numbers have been massaged for those thatwant to reproduce the calculations.

**or a combination of current and area scaling toachieve a 10:1 current density ratio in Equation(1).


DESIGN INFORMATION

are applied to the equation shown,resulting in an average temperaturecoefficient expressed in V/°C. This isfurther manipulated to find ppm/°C,as used in the data sheet. The tempcois an average over the operating range,rather than an incremental slopemeasured at any specific point. In thecase of the LT1021 and LT1236, theincremental slope at 25°C is alsoguaranteed.

A data sheet figure for tempco canbe used to directly calculate the out-put voltage tolerance over the entireoperating temperature range. A devicewith a tempco of 10ppm/°C, specifiedfor 0°C to 70°C, could drift up to700ppm from the initial value (about3 counts in a 12-bit system). A 0.1%reference with 700ppm tempco erroris guaranteed 0.17% accurate over itsentire operating temperature range.

Two exceptions to this rule are theLT1004 and LT1034, which simplyguarantee absolute output voltageaccuracy over the entire operatingtemperature range. The LT1009 andLT1029 use a combination of the two,called the “bow tie” or “butterfly”method (see the LT1009 data sheetfor a detailed explanation).

Neither the bandgap nor buriedZener, in their basic form, are inher-ently low drift. Special on-chipcircuitry is used to improve the tempcoof the reference core. A buried Zeneris first-order compensated againsttemperature changes by adding a P-Njunction diode. The Zener itself mea-sures +2mV/°C and the diode–2mV/°C. The combination of the twoin series cancel to about 0.2mV/°C

(≈30ppm/°C) out of a total of 7V.Interestingly, this is very close to thetempco of a saturated Weston cell,which measures –40µ V/°C, or–39ppm/°C. Weston cells are held ina temperature-controlled bath; mono-lithic buried Zener references arefurther compensated against tem-perature changes by carefully addingfractional VBE and/or ∆VBE terms tothe output. Post-manufacturing trimsare used on both bandgap and buriedZener products to further minimizetempco of the finished reference.

Another detractor from accuracy islong-term stability. The output of areference changes, usually in onedirection, as it ages. The effect islogarithmic; that is, the outputchanges less and less as timeprogresses. The units of long-termstability, ppm/√kh (kh = 1000 hours),reflect the logarithmic decline of theoutput change vs time. Because long-term changes in the output are smalland occur over the course of monthsor years, it is impossible to devise anaffordable manufacturing test to guar-antee the true stability of allreferences. Instead, this parameter ischaracterized by aging dozens of unitsin a temperature-controlled chamberat 25°C to 30°C for 1000 hours ormore. Note that the absolute tem-perature is unimportant, but it mustremain invariant during the course ofthe test. Mathematically extrapolat-ing long-term stability data from hightemperature, accelerated life testsleads to erroneously optimistic roomtemperature results.

When long-term stability is guar-anteed, it is done by means of a4-week burn-in, during which mul-tiple output voltage measurementsare made. Even with this elaborate,costly procedure, the guaranteed limitis about three to four times the typicaldrift.

Unless the product is designed forfrequent calibration or is relativelylow performance, long-term stabilitymay be an important aspect of refer-ence performance. Products designedfor a long calibration cycle must holdtheir accuracy for extended periods oftime without intervention. These prod-ucts demand references with goodlong-term stability. You can expectburied Zeners to perform better than20ppm/√kh, and bandgaps between20ppm and 50ppm/√kh. Some of thisdrift is attributed to the trim andcompensation circuitry wrappedaround the reference core. TheLTZ1000 dispenses with trim andcompensation overhead in favor of anon-chip heater. The remaining Zener/diode core drifts 0.5ppm/√kh in thefirst year of operation, approachingthe stability of a Weston cell.

Most of the long-term stability fig-ures shown in LTC reference datasheets are for devices in metal canpackages, where assembly and pack-age stresses are minimized. You canexpect somewhat less performancefor the same reference in a plasticpackage.

One last factor that affects accu-racy is short-term variation of outputvoltage, otherwise known as noise.

TEMPERATURE (˚C)–50

NORM

ALIZ

ED O

UTPU

T VO

LTAG

E (V

)

1.001

1.002

1.003

25 75

REFACC_02.eps

1.000

0.999

–25 0 50 100 125

0.998

0.997

10ppm/°C FULL TEMP RANGE “BOX”

5ppm/°C 0°C TO 70°C “BOX”

LT1019 CURVE

AVERAGETEMPERATURE

V VT T

VCCOEFFICIENT

MAX MIN

MAX MIN=

−− °

Figure 2. The box method expresses absoluteoutput accuracy over temperature as a driftterm.

retemaraP noitpircseD )s(tinUderreferP

ycaruccAlaitinI C˚52taegatlovtuptuolaitinI %,V

tneiciffeoCerutarepmeTV XAM V– NIM

EGNARERUTAREPMETLATOTC˚/mpp

ytilibatSmreT-gnoL der usaememit svtuptuoniegnahCeromrosruoh0001revo /mpp hk

esioNzH01otzH1.O µV P-P mpp, P-P

zHk1otzH01 µV SMR mpp, SMR

Table 1. Reference accuracy specifications


DESIGN INFORMATION

Reference noise is typically charac-terized over two frequency ranges:0.1Hz to 10Hz for short-term, peak-to-peak drift, and 10Hz to 1kHz fortotal “wideband” RMS noise. Noisevoltage is usually proportional to out-put voltage, so the output noiseexpressed in ppm is constant for allvoltage options of any given refer-ence. Wideband noise ranges from4ppm–16ppm RMS for bandgap ref-erences, to 0.17ppm–0.5ppm RMSfor buried Zeners. Noise improveswith increased reference current,regardless of reference type. But sincethe reference core operating currentis set internally, the noise character-istics cannot be changed except byexternal filtering (the LT1027 featuresa noise filtering pin). The LT1034 andLTZ1000 buried Zeners are externallyaccessible, allowing the user toincrease the bias current and reducenoise.

Adding output bypassing or exter-nal compensation will affect thecharacter of a reference’s noise. In

particular, if the compensation is“peaky,” the spot noise will likely riseto a peak somewhere in the 100Hz to10kHz range. Critical damping willeliminate this noise peak.

Reference noise can affect thedynamic range of a high resolutionsystem, obscuring small signals. Lowfrequency noise also complicates themeasurement of output voltage.Modern, high accuracy digital volt-meters can average many readings tohelp filter low frequency noise effectsand provide a stable reading of areference’s true output voltage.

Essential FeaturesThere are two styles of references:shunt, functionally equivalent to aZener diode; and series, not unlike a3-terminal regulator. Bandgaps andburied Zeners are available in bothconfigurations (see Figure 3). Someseries references are designed to alsooperate in shunt mode by simplybiasing the output pin and leavingthe input pin open circuit. Series-

mode references have the advantagethat they draw only load and quies-cent current from the input supply,whereas shunt references must bebiased with a current that exceedsthe sum of the maximum quiescentand maximum expected load cur-rents. Since they are biased by aresistor, shunt references can oper-ate on a very wide range of inputvoltages.

About half of LTC’s reference offer-ings include a pin for external(customer) trimming. Some aredesigned for precision trimming ofthe reference output, whereas othershave a wide trim range, allowing theoutput voltage to be adjusted severalpercent above or below the intendedoperating point.

Many voltage options are availablefor both bandgaps and buried Zeners.Table 2 shows the voltage options foreach LTC reference, plus a summaryof reference type, operating modesand external trim option.

traP pagdnaBdeiruBreneZ seireS tnuhS V52.1 V5.2 V5.4 V0.5 V0.7 V01 mirT

4001TL

9001TL

9101TL

1201TL )V01,V7( **

7201TL

9201TL

1301TL

4301TL * *

6321TL )ylnoV01(

0641TL

4361TL

0001ZTL 2.7

* .reneZdeirubedom-tnuhsasedulcniecnereferpagdnab4301TLehT*.demmirtebtonnac7-1201TLehT**

Table 2. True to late twentieth century form, LTC references offer many choices.


DESIGN INFORMATION

If load current steps must behandled, transient response isimportant. Transient response varieswidely from reference to referenceand comprises three distinct qualities:turn-on characteristics, small-signaloutput impedance at high frequencyand settling behavior when subjectedto a fast, transient load. Referencesexhibit these qualities because al-most all contain an amplifier to bufferand/or scale the output.

The LT1009 is optimized for faststart-up characteristics, and it settlesin a little over 1µs, as shown in Figure4. For some references, optimum set-tling is obtained with an externalcompensation network. As shown inFigure 5, a 2µF/2Ω damper optimizesthe settling and high frequency outputimpedance of an LT1019 reference.Fastest settling is obtained with anLT1027, which settles to 13 bitsaccuracy in 2µs. This impressive featis illustrated by the oscillograph ofFigure 6, which clearly shows theoutput recovering from a 10mA loadstep.

Reference PitfallsReferences look deceptively simple touse, but like any other precision prod-uct, maximum performance is notnecessarily easy to achieve. Here area few common pitfalls reference usersface, and ways to beat them.

Current-Hungry LoadsMost references are specified for maxi-mum load currents (or shunt currents)of 10mA–20mA. Nevertheless, bestperformance is not obtained by run-ning the reference at maximumcurrent. A number of effects, includ-ing thermal gradients across the dieand thermocouples formed betweenthe leads and external circuit con-nections, may limit the short-termstability of the output voltage. Addingan external pass transistor, as shownin Figure 7, removes the load currentfrom the reference. For loads greaterthan 300µA, the pass transistor car-ries almost all of the current andeliminates short-term thermal drift.This circuit is also useful forapplications requiring more than20mA, and easily supports up to100mA, limited only by transistor betaand dissipation.

“NC” PinsIf references need only two or threeexternal connections, why are theysupplied in 8-pin packages? Thereare several reasons, but the one we’llcover here is post-package trimming.To guarantee tight output tolerances,some factory trimming is necessaryafter the device has been packaged.In packaged form we no longer have

direct access to the die, so the extrapins on an 8-pin package are used toeffect post-package trimming.

For some ICs, “NC” means “this pinis floating, you can hook it up towhatever you want.” In the case of areference, it means “don’t connectanything to this pin.” That includesESD and board leakage, as well asintentional connections. External con-nections will, at best, cause outputvoltage shifts and, at worst, perma-nently shift the output voltage out ofspec.

A similar caution applies to theTRIM pin on references with adjust-able outputs. The TRIM pin is akin toan amplifier’s summing node; do notinject current into a TRIM pin—unlessyou want to trim the output, of course.Here board leakage or capacitive cou-pling to noise sources are pitfalls toavoid.

REFACC_03.eps

A

(a) SHUNTK

VREF OUTIN

GND

(b) SERIES

VREF

TIME (µs)

0

0.5

0

8

4

3.5

3.0

2.5

2.0

1.5

1.0 5k

VOLT

AGE

SWIN

G (V

)

20

REFACC_04.eps

0 1

OUTPUT

OUTPUT

INPUT

INPUT

VIN

2Ω TO 5Ω

LT1019

REFACC_05.eps

+2µF TANTALUM

LT1460-10

OUT

V+ ≥ (VOUT + 1.4V)

GND

IN

REFACC_07.eps

+

2N2905

10V AT 100mA

2µF SOLID TANT

R1 1.8kΩ

Figure 3. References are supplied in either 2-terminal Zener style (a) or 3-terminal voltageregulator style (b).

Figure 4. The LT1009 is optimized for rapidsettling at power-up.

Figure 5. Optimum settling realized withRC compensation at output

Figure 6. The LT1027 is optimized forfast settling in response to load steps.

Figure 7. An external transistor is useful forboosting output current as well as forremoving load current from the reference.This trick works on all 3-terminal references.

This article will conclude in the August issue ofLinear Technology; if you can’t wait for the thrill-packed conclusion, you can order the second halfby checking the appropriate box on the responsecard.

VOUT400µV/DIV

AC COUPLED

10mALOAD STEP

2µs/DIV


NEW DEVICE CAMEOS

LT1635 MicropowerOp Amp and ReferenceThe LT1635 is a new analog buildingblock that includes a high quality opamp, precision reference and refer-ence buffer. The LT1635 combinesprecision specifications with single-supply micropower operation. Animportant feature of the device isoperation on an unusually low 1.2Vsingle supply, or dual supplies of upto ±5V; the LT1635 consumes a mere130µA of supply current.

The input common mode range ofthe op amp includes ground andincorporates phase-reversal protec-tion to prevent false outputs fromoccurring when the input is below thenegative supply. The rail-to-rail out-put stage can swing to within 15mV ofeach rail with no load and can deliver20mA output current while driving towithin 400mV of either supply. Thegain bandwidth of the op amp is200kHz; it is unity gain stable with upto 1000pF of load capacitance.

The 0.2V precision bandgap refer-ence is referred to V– and includes abuffer amplifier to enhance the flex-ibility of the LT1635. The referenceand buffer combine to achieve a driftof only 30ppm/˚C, a load regulationof 150ppm/mA and a line regulationof 20ppm/V.

The LT1635 is offered in SO-8 and8-pin DIP packages, in both commer-cial and industrial temperaturegrades, and has been optimized forboth single 5V and ± 5V operation.

LT1492/LT1493:550µA, 5MHz, 3V/µsSingle-Supply, PrecisionDual and Quad Op AmpsThe LT1492 and LT1493 dual andquad precision operational amplifi-ers are ideal for low power andsingle-supply applications thatrequire DC accuracy, high speed andhigh output current.

The LT1492/LT1493 operate overa wide supply range of 2.5V to 36Vtotal and draw a maximum supply

current of only 550µA. The devicesfeature 5MHz gain bandwidth, a slewrate of 3V/µs and can deliver a mini-mum of 20mA output-drive current.

In addition to the aforementionedAC specifications, the LT1492/LT1493 have excellent DC specs. Withless than 180µV of input offset volt-age, 100nA input bias current and20nA offset current, the LT1492/LT1493 eliminate trims in most sys-tems. A minimum open-loop voltagegain (AVOL) of 1500V/mV (VS = ±15V,RL = 5k) ensures a very small gainerror. Furthermore, the inputs can bedriven beyond the supplies withoutdamage or phase reversal of theoutput.

The LT1492 is available in plastic8-pin DIP and SO-8 packages withthe standard dual op amp pinout.The LT1493 is available in 16-pin SOpackage.

LTC1540 Ultralow PowerComparator and ReferenceThe LTC1540 is an ultralow powercomparator with a built-in reference.The comparator draws only 0.35µAsupply current with a 5V power sup-ply and features an internal, 1.182V(±2%) reference. It also has program-mable hysteresis and a TTL/CMOSoutput that can sink or source cur-rent. The reference output can drive abypass capacitor of up to 0.01µF with-out oscillation and can source up to1mA and sink up to 20µA.

The comparator operates from asingle 2V–11V supply or dual ±1V to±5.5V supplies. Comparator hyster-esis is easily programmed using tworesistors and the HYST pin. Thecomparator’s input range extendsfrom the negative supply to within1.3V of the positive supply.

The LTC1540 is pin compatiblewith the LTC1440. It is available in8-pin SO and MSOP packages.

New Device CameosUltralow IQ LTC1474/LTC1475 High EfficiencyStep-Down DC/DCConverters Now Availablewith Fixed Output VoltagesThe LTC1474/LTC1475, featuring3V–18V operation, 10µA typical qui-escent current and a tiny 8-pin MSOPpackage, are now available in fixed3.3V and 5V output versions. TheLTC1474-3.3 and LTC1474-5 con-tain internal feedback resistorstrimmed for output voltages of 3.3Vand 5V, respectively. As with theadjustable version, they are controlledby a RUN pin and feature a low-battery comparator that remainsactive in shutdown. The LTC1475-3.3 and LTC1474-5 have all of theabove features, plus an ON/OFF latchfor push-button control of power. Theadjustable versions of the LTC1474and LTC1475 are also available.

All six members of the LTC1474/LTC1475 family feature operatingefficiencies exceeding 90% and a com-bination of cycle-by-cycle inductorcurrent control and ultralow quies-cent current previously unavailablein switching regulators. Strapping twopins together defines a 400mA peakinductor current with no externalcurrent sense resistor, allowing out-put currents of up to 300mA . Byadding an inexpensive external resis-tor, the user can program the peakinductor current to be as low as 10mA,for efficient low current operation withsmall inductors.

The LTC1474/LTC1475 are idealfor many quiescent-current sensitiveapplications, such as battery-pow-ered, handheld devices, keep-alivepower supplies and industrial 4–20mAloops. In addition to the small-foot-print MS8 package, all device typesare also available in the standard8-lead SO package.



NEW DEVICE CAMEOS



www.linear-tech.com

For further information on anyof the devices mentioned in thisissue of Linear Technology, usethe reader service card or callthe LTC literature servicenumber:

1-800-4-LINEAR

Ask for the pertinent data sheetsand Application Notes.

LTC1439: a 40% SmallerPackage for LTC’s Full-Function, Low Noise,Multiple Output ControllerThe LTC1439 is now offered in anarrower and shorter “G” package,measuring 0.2″ × 0.5″, down from the0.3″ × 0.6″. “GW” package. The totalpackage “footprint” including the pinshas been reduced from 0.4″ × 0.6″ to0.3″ × 0.5″, a 40% PC board savings.

The LTC1439 offers the most com-pact power supply system solutionfor applications requiring a constant-frequency, dual controller with a 1%guaranteed reference and 1% loadand line regulation over its entireoperating temperature range. TheAdaptive Power™ output stage maxi-mizes efficiency while maintainingconstant frequency operation by dy-namically switching between twooptimally sized N-channel output

power MOSFETs, depending uponloading conditions. This techniquedelivers true constant frequencyoperation over two decades of outputcurrent—down to typically 1% of thedesigned maximum output load. Thistechnique eliminates the possibilityof audible artifacts that can be pro-duced by the switching power supply’sinductor or transformer undernoncontinuous inductor operation.The controller switches over to BurstMode operation at very low outputcurrents, maximizing efficiency whena system is in standby mode. Exter-nal frequency compensation ensuresoptimal transient response and over-all loop stability in a variety ofapplications and topologies. A power-on reset output holds its output lowfor system reset for 65,536 clock cycles(typically 300ms) after the firstcontroller’s output has risen to 95%

of its final output voltage. An auxil-iary linear regulator with an externalpass device is capable of supplyingany required voltage/current combi-nation that might be required for thepower supply system. An extra com-parator whose negative input is tiedto the internal reference is availableto be used for a low-battery compara-tor or other system function. The firstcontroller can be pin selected toprovide a 5V or a 3.3V output and thesecond controller can be programmedto be a 5V, 3.3V or an adjustableoutput having a range of from 1.2V to9V. The controllers have logic-con-trolled independent shutdown andprogrammable soft-start. A truephase-locked loop can lock the “con-stant” frequency over a 2:1 range orcan be used for frequency shifting orspread-spectrum operation.


DESIGN TOOLS

Applications on DiskNoise Disk — This IBM-PC (or compatible) programallows the user to calculate circuit noise using LTC opamps, determine the best LTC op amp for a low noiseapplication, display the noise data for LTC op amps,calculate resistor noise and calculate noise using specsfor any op amp. Available at no charge

SPICE Macromodel Disk — This IBM-PC (or compat-ible) high density diskette contains the library of LTCop amp SPICE macromodels. The models can be usedwith any version of SPICE for general analog circuitsimulations. The diskette also contains working circuitexamples using the models and a demonstration copyof PSPICE™ by MicroSim. Available at no charge

SwitcherCAD™ — The SwitcherCAD program is a pow-erful PC software tool that aids in the design andoptimization of switching regulators. The program cancut days off the design cycle by selecting topologies,calculating operating points and specifying compo-nent values and manufacturer’s part numbers. 144page manual included. $20.00

SwitcherCAD supports the following parts: LT1070series: LT1070, LT1071, LT1072, LT1074 and LT1076.LT1082. LT1170 series: LT1170, LT1171, LT1172 andLT1176. It also supports: LT1268, LT1269 and LT1507.LT1270 series: LT1270 and LT1271. LT1371 series:LT1371, LT1372, LT1373, LT1375, LT1376 andLT1377.

Micropower SwitcherCAD™ — The MicropowerSCADprogram is a powerful tool for designing DC/DC con-verters based on Linear Technology’s micropowerswitching regulator ICs. Given basic design param-eters, MicropowerSCAD selects a circuit topology andoffers you a selection of appropriate Linear Technologyswitching regulator ICs. MicropowerSCAD also per-forms circuit simulations to select the other componentswhich surround the DC/DC converter. In the case of abattery supply, MicropowerSCAD can perform a bat-tery life simulation. 44 page manual included.

$20.00

MicropowerSCAD supports the following LTC micro-power DC/DC converters: LT1073, LT1107, LT1108,LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,LT1300, LT1301 and LT1303.

Technical Books1990 Linear Databook, Vol I —This 1440 page collec-tion of data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conver-sion and interface products (bipolar and CMOS), inboth commercial and military grades. The catalogfeatures well over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 pagesupplement to the 1990 Linear Databook is a collectionof all products introduced in 1991 and 1992. Thecatalog contains full data sheets for over 140 devices.The 1992 Linear Databook, Vol II is a companion to the1990 Linear Databook, which should not be discarded.

$10.00

1994 Linear Databook, Vol III —This 1826 pagesupplement to the 1990 and 1992 Linear Databooks isa collection of all products introduced since 1992. Atotal of 152 product data sheets are included withupdated selection guides. The 1994 Linear DatabookVol III is a companion to the 1990 and 1992 LinearDatabooks, which should not be discarded. $10.00

1995 Linear Databook, Vol IV —This 1152 pagesupplement to the 1990, 1992 and 1994 Linear Da-tabooks is a collection of all products introduced since1994. A total of 80 product data sheets are includedwith updated selection guides. The 1995 Linear Data-book Vol IV is a companion to the 1990, 1992 and 1994Linear Databooks, which should not be discarded.

$10.00

1996 Linear Databook, Vol V —This 1152 page supple-ment to the 1990, 1992, 1994 and 1995 LinearDatabooks is a collection of all products introducedsince 1995. A total of 65 product data sheets areincluded with updated selection guides. The 1996Linear Databook Vol V is a companion to the 1990,1992, 1994 and 1995 Linear Databooks, which shouldnot be discarded. $10.00

1990 Linear Applications Handbook, Volume I —928 pages full of application ideas covered in depth by40 Application Notes and 33 Design Notes. This cata-log covers a broad range of “real world” linear circuitry.In addition to detailed, systems-oriented circuits, thishandbook contains broad tutorial content togetherwith liberal use of schematics and scope photography.A special feature in this edition includes a 22-pagesection on SPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —Continues the stream of “real world” linear circuitryinitiated by the 1990 Handbook. Similar in scope to the1990 edition, the new book covers Application Notes40 through 54 and Design Notes 33 through 69.References and articles from non-LTC publicationsthat we have found useful are also included. $20.00

1997 Linear Applications Handbook, Volume III —This 976 page handbook maintains the practical outlookand tutorial nature of previous efforts, while broaden-ing topic selection. This new book includes ApplicationNotes 55 through 69 and Design Notes 70 through144. Subjects include switching regulators, measure-ment and control circuits, filters, video designs,interface, data converters, power products, batterychargers and CCFL inverters. An extensive subjectindex references circuits in LTC data sheets, designnotes, application notes and Linear Technology maga-zines. $20.00

Interface Product Handbook — This 424 page hand-book features LTC’s complete line of line driver andreceiver products for RS232, RS485, RS423, RS422,V.35 and AppleTalk® applications. Linear’s particularexpertise in this area involves low power consumption,high numbers of drivers and receivers in one package,mixed RS232 and RS485 devices, 10kV ESD protec-tion of RS232 devices and surface mount packages.

Available at no charge

Power Solutions Brochure — This 84 page collectionof circuits contains real-life solutions for commonpower supply design problems. There are over 88circuits, including descriptions, graphs and perfor-mance specifications. Topics covered include batterychargers, PCMCIA power management, microproces-sor power supplies, portable equipment power supplies,micropower DC/DC, step-up and step-down switchingregulators, off-line switching regulators, linear regula-tors and switched capacitor conversion.


High Speed Amplifier Solutions Brochure —This 72 page collection of circuits contains real-lifesolutions for problems that require high speedamplifiers. There are 82 circuits including descrip-tions, graphs and performance specifications. Topicscovered include basic amplifiers, video-related appli-cations circuits, instrumentation, DAC and photodiodeamplifiers, filters, variable gain, oscillators and currentsources and other unusual application circuits.


Data Conversion Solutions Brochure — This 52 pagecollection of data conversion circuits, products andselection guides serves as excellent reference for thedata acquisition system designer. Over 60 productsare showcased, solving problems in low power, smallsize and high performance data conversion applica-tions—with performance graphs and specifications.Topics covered include ADCs, DACs, voltage refer-ences and analog multiplexers. A complete glossarydefines data conversion specifications; a list of se-lected application and design notes is also included.


Telecommunications Solutions Brochure — This 72page collection of circuits, new products and selectionguides covers a wide variety of products targeted forthe telecommunications industry. Circuits solving reallife problems are shown for central office switching,cellular phone, base station and other telecom applica-tions. New products introduced include high speedamplifiers, A/D converters, power products, interfacetransceivers and filters. Reference material includes atelecommunications glossary, serial interface stan-dards, protocol information and a complete list of keyapplication notes and design notes.



DESIGN TOOLS

Acrobat is a trademark of Adobe Systems, Inc. AppleTalkis a registered trademark of Apple Computer, Inc. PSPICE™is a trademark of MicroSim Corp.

Information furnished by Linear Technology Corporationis believed to be accurate and reliable. However, LinearTechnology makes no representation that the circuitsdescribed herein will not infringe on existing patent rights.

Linear Technology Magazine • June 1997© 1997 Linear Technology Corporation/Printed in U.S.A./

LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7417(408) 432-1900 FAX (408) 434-0507www.linear-tech.comFor Literature Only: 1-800-4-LINEAR

InternationalSales OfficesFRANCELinear Technology S.A.R.L.Immeuble “Le Quartz”58 Chemin de la Justice92290 Chatenay MalabryFrancePhone: 33-1-41079555FAX: 33-1-46314613

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UNITED KINGDOMLinear Technology (UK) Ltd.The Coliseum, Riverside WayCamberley, Surrey GU15 3YLUnited KingdomPhone: 44-1276-677676FAX: 44-1276-64851

World HeadquartersLinear Technology Corporation1630 McCarthy BoulevardMilpitas, CA 95035-7417Phone: (408) 432-1900FAX: (408) 434-0507

U.S. AreaSales OfficesNORTHEAST REGIONLinear Technology Corporation3220 Tillman Drive, Suite 120Bensalem, PA 19020Phone: (215) 638-9667FAX: (215) 638-9764

Linear Technology Corporation266 Lowell St., Suite B-8Wilmington, MA 01887Phone: (508) 658-3881FAX: (508) 658-2701

NORTHWEST REGIONLinear Technology Corporation1900 McCarthy Blvd., Suite 205Milpitas, CA 95035Phone: (408) 428-2050FAX: (408) 432-6331

SOUTHEAST REGIONLinear Technology Corporation17000 Dallas ParkwaySuite 219Dallas, TX 75248Phone: (214) 733-3071FAX: (214) 380-5138

Linear Technology Corporation5510 Six Forks RoadSuite 102Raleigh, NC 27609Phone: (919) 870-5106FAX: (919) 870-8831

CENTRAL REGIONLinear Technology CorporationChesapeake Square229 Mitchell Court, Suite A-25Addison, IL 60101Phone: (708) 620-6910FAX: (708) 620-6977

SOUTHWEST REGIONLinear Technology Corporation21243 Ventura Blvd.Suite 227Woodland Hills, CA 91364Phone: (818) 703-0835FAX: (818) 703-0517

Linear Technology Corporation15375 Barranca ParkwaySuite A-211Irvine, CA 92718Phone: (714) 453-4650FAX: (714) 453-4765

CD-ROMLinearView — LinearView™ CD-ROM version 2.0 isLinear Technology’s latest interactive CD-ROM. It al-lows you to instantly access thousands of pages ofproduct and applications information, covering LinearTechnology’s complete line of high performance ana-log products, with easy-to-use search tools.

The LinearView CD-ROM includes the complete prod-uct specifications from Linear Technology’s Databooklibrary (Volumes I–V) and the complete ApplicationsHandbook collection (Volumes I–III). Our extensivecollection of Design Notes and the complete collectionof Linear Technology magazine are also included.

A powerful search engine built into the LinearView CD-ROM enables you to select parts by various criteria,such as device parameters, keywords or part numbers.All product categories are represented: data conver-sion, references, amplifiers, power products, filtersand interface circuits. Up-to-date versions of LinearTechnology’s software design tools, SwitcherCAD,Micropower SwitcherCAD, FilterCAD, Noise Disk andSpice Macromodel library, are also included. Every-thing you need to know about Linear Technology’sproducts and applications is readily accessible viaLinearView. LinearView 2.0 runs under Windows® 3.1,Windows 95 and Macintosh® System 7.0 or later.

Available at no charge.

World Wide Web SiteLinear Technology Corporation’s customers can nowquickly and conveniently find and retrieve the latesttechnical information covering the Company’s prod-ucts on LTC’s new internet web site. Located atwww.linear-tech.com, this site allows anyone withinternet access and a web browser to search throughall of LTC’s technical publications, including data sheets,application notes, design notes, Linear Technologymagazine issues and other LTC publications, to findinformation on LTC parts and applications circuits.Other areas within the site include help, news andinformation about Linear Technology and its salesoffices.

Other web sites usually require the visitor to downloadlarge document files to see if they contain the desiredinformation. This is cumbersome and inconvenient. Tosave you time and ensure that you receive the correctinformation the first time, the first page of each datasheet, application note and Linear Technology maga-zine is recreated in a fast, download-friendly format.This allows you to determine whether the document iswhat you need, before downloading the entire file.

The site is searchable by criteria such as part numbers,functions, topics and applications. The search is per-formed on a user-defined combination of data sheets,application notes, design notes and Linear Technologymagazine articles. Any data sheet, application note,design note or magazine article can be downloaded orfaxed back. (Files are downloaded in Adobe Acrobat™PDF format; you will need a copy of Acrobat Reader toview or print them. The site includes a link from whichyou can download this program.)

DESIGN TOOLS, continued from page 39

Acrobat is a trademark of Adobe Systems, Inc.; Windows isa registered trademark of Microsoft Corp.; Macintosh is aregistered trademark of Apple Computer, Inc.

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