LINEAR TECHNOLOGY LINEAR TECHNOLOGY · Product Group at Linear Technology Corp. Bill has been...

JUNE 1992 VOLUME II NUMBER 2

IntroductionThe LT1227 is the best low-cost,

current-feedback amplifier for videoapplications. The amplifier is unsur-passed for use in professional studioequipment, security systems, per-sonal-computer display adapters,and workstations. The LT1227 isLTC’s fastest amplifier; its bandwidthis over 140MHz and its slew rate isover 1100V/µs. The LT1227 operateson all supplies from +5V to ±15V. Itcomes in 8-pin DIPs and 8-pin SOsand uses the industry standard pin-out, with offset adjust and a shutdownpin for multiplexing several amplifiers’outputs.

Types of VideoThere are many different video for-

mats and standards, but they allbelong to one of two groups: RGB orcomposite. RGB stands for red, green,and blue; these are the three primarycolors in a color picture tube. All colorsfrom black (no R, G, or B) to white(maximum R, G, and B) can be pro-duced from various combinationsof these three components. The RGBformat requires three signals that rep-resent the amplitudes of the respectiveprimary colors plus timing signals(sync) that are sometimes combinedwith the green component. Compositevideo refers to a single signal thatcontains all of the color and timinginformation described above. Compos-ite video modulates the RF signal thatyour TV receives.

Video-Amplifier RequirementsIn order to better understand the

design of the LT1227, we will brieflyreview the requirements of video sys-tems. Video signals typically swingabout 1V positive and 400mV nega-tive; they have information spread overa wide frequency range, from near DCto several MHz. To preserve the lowfrequency information, most video am-plifiers are DC coupled, if possible.However, DC offset is not a very impor-tant parameter for video amplifiers.This is because the information neededto restore or clamp the DC zero level isin the video signal. The DC level isusually restored at the inputs of moni-tors and test equipment, and at theinputs of A-to-D converters. Therefore,the DC offset of a video amplifier needsonly to be low enough to prevent dis-tortion of the output signal. Large sup-plies make this easier, and today mostvideo systems operate on ±12V sup-plies, although a growing number ofsystems operate on ±5V.

Introducing the LT1227:A Video Amplifierfor All Seasons

IN THIS ISSUE . . .

COVER ARTICLEIntroducing the LT1227:A Video Amplifierfor All Seasons ............... 1William H. Gross

Editor's Page .................. 2Richard Markell

DESIGN FEATURESIllumination Circuitryfor Liquid Crystal Displays....................................... 3Jim Williams

The LTC1153 ElectronicCircuit Breaker .............. 8Tim Skovmand

The LTC1063 Low-Offset,Clock-Tunable, Fifth-OrderButterworth Lowpass Filter....................................... 10Nello Sevastopoulos

The LTC1047: New DualMicropower Zero-Drift OpAmp................................ 12Dave Dwelley

The LT1121 Low-DropoutRegulator ....................... 14Dennis O'Neill

DESIGN IDEASCascaded 8th-Order Butter-worth Filters Provide SteepRoll-Off Lowpass Filter ... 18Philip Karantzalis and Richard Markell

LT1110 Supplies 6 Voltsat 550mA from2 AA NiCad Cells ............ 19Steve Pietkiewicz

DESIGN SOFTWARESwitcherCAD Release Slatedfor August ...................... 20Brian Huffman

New Device Cameos ........ 22LTC Marketing

continued on page 16

ISO 9000

COMPLIANT

LINEAR TECHNOLOGY LINEAR TECHNOLOGY LINEAR TECHNOLOGY

by William H. Gross

2 Linear Technology Magazine • June 1992

DESIGN FEATURES

Another Lonely Dayat the Word Processor

This issue is one of our most diverseyet. We have articles on the design ofnew integrated circuits, such as theLT1227 Video Amplifier, we have anarticle applying the LTC1047 to a “realworld” instrumentation amplifier prob-lem, and we have application articleson cold-cathode fluorescent lightingcircuits, palm-top pager circuits andthe SwitcherCAD design software. Adiverse group of articles all in all.

Linear Technology Corporation con-tinues to offer highly cost-effectivesolutions to an ever increasing varietyof linear circuit problems. Our inte-grated-circuit design group hasbecome increasingly involved in the

design of power-supply solutions forall types of 5 Volt and 3.3 Volt comput-ers and hand-held devices such aspalm-top computers, pagers, cellularphones, and bar-code readers. Ournew high-speed line of operational am-plifiers provide solutions for a plethoraof RF, analog, and digital video appli-cations. These amplifiers provide cost-effective solutions for the growing worldof Multimedia, DVI, CD-I and CD-Vapplications.

Another area that LTC is involved inis power control. We are developingcircuits for driving power MOSFETs,including Motor Control IC’s, solid-

state circuit breakers, switchingregulators, and related devices.

Linear Technology is also involvedin a variety of products that defy easyclassification. These include CCFLdisplays, portable FAX’s, palm top com-puters, pen computers, LANs andother networking schemes, andgeopositioning devices.

We at LTC continue to offer what-ever design and application assistanceis required to support our integratedcircuits. Do not hesitate to call us todiscuss your circuit, your applicationor how we can help you do the best,most cost effective job on your design.

Issue HighlightsBill Gross leads off this issue

with his article on the LT1227 VideoAmplifier. The LT1227 is LTC’s fastestamplifier, with a bandwidth of over140MHz and a slew rate greater than1100 Volts/microsecond. This is Bill’ssecond article for LT. He is the designmanager for the High Speed AmplifierProduct Group at Linear TechnologyCorp. Bill has been designing inte-grated circuits in the semiconductorindustry for over 20 years. He is mar-ried and is the father of two teenagedsons, whose sports activities keep himglued to the playing field.

Nello Sevastopoulos describes an-other innovation in filter technology,the LTC1063. This is the lowest-offsetswitched capacitor filter currently avail-able anywhere. Nello has worked in thesemiconductor industry for 20 years,eight years as an applications man-ager and twelve years as an IC de-signer. Nello designed the MF10, theMF4, the MF8, the LTC1060, theLTC1064, and the entire line of LinearTechnology’s switched-capacitor filters.His outside interests include his fam-ily, travel, and Harley Davidsons (hisparticular mid-life crisis). He and theeditor, and many other designers, look

forward to the first crop of tomatoes,olives, and feta cheese to celebrate themid-summer tomato season.

Dennis O’Neill has been with LTCfor eight years. Before that he workedat another famous brand name wherehe toiled for Carl Nelson. At LTC hehas designed many of our bread-and-butter voltage regulators, includingthe LT1083, LT1084, LT1085, LT1086,and LT1054. He also designed theLT1241 series of PWM controllers, theLT1123, and the new LT1121. Dennis’interests include bicycling, photo-graphy, cooking, and the infiniteremodeling of his house.

Brian Huffman has been an Appli-cations Engineer at LTC for five years.Before that he designed a ground-faultinterrupter and did device modelingfor another company. Brian has beencompletely immersed in theSwitcherCAD software for the past year.He may be able to recite spreadsheetequations in his sleep. His outsideinterests include country western danc-ing and being a “thrill junkie.” It’s notunusual to find Brian at “The SaddleRack” on Tuesdays or climbing insidecaves (spelunking) on weekends.

Jim Williams needs no introductionto these pages. He has been by farthe most significant author of LTCApplication Notes, Design Notes andarticles. Jim has been with LTC foreight years. His article on backlights is“enlightening” both from a technicalperspective and for the reason that thisis a hot topic these days. Jim’s outsideinterests include his son, whose out-grown shoes continue to accumulatein Williams’ office, and antique scien-tific instruments (what a combo!).

Dave Dwelley, the baby of the Inte-grated Circuit Design Group, workedin sonar for two years before beinghired by Linear Technology as a FieldApplications Engineer. Dave actuallygot a close up view of a relative of thegiant squid that Captain Nemo saw (in20,000 Leagues Under the Sea) whenhe got to ride in a submarine to test hiscircuits. Dave has been designingCMOS zero-offset amplifiers since hemoved from being an FAE to being anIC designer. His outside interests in-clude a 1959 Chevy (not Chebyshev)Impala that he is restoring, famouswriters school, mountain bike riding,and preparing to restore his very ownhouse.

by Richard Markell

EDITOR’S PAGE

DESIGN FEATURES

Linear Technology Magazine • June 1992 3

Illumination Circuitryfor Liquid Crystal Displays

Current-generation portable com-puters and instruments employback-lit liquid crystal displays (LCDs).Cold-cathode fluorescent lamps(CCFLs) provide the highest availableefficiency for backlighting the display.These lamps operate on high-voltageAC and therefore require efficient high-voltage DC-AC converters. In additionto providing high efficiency, convert-ers used with CCFLs should deliverthe lamp drive in the form of a sinewave. This is desirable to minimize RFemissions, which can cause interfer-ence with other devices and degradeoverall operating efficiency. The cir-cuit should also permit lamp-inten-sity control from zero to full brightnesswith no hysteresis or “pop-on.”

The LCD also requires a bias supplyfor contrast control. The supply’s nega-tive output should be regulated andshould be variable over a considerablerange.

Because of their small size andbattery-powered operation, LCD-equipped devices require low compo-nent count and high efficiency. Sizeconstraints place severe limitationson circuit architecture, and long bat-tery life is usually a priority. Laptopand hand-held portable computersare excellent examples. In these appli-cations, the CCFL and its power sup-ply are responsible for almost 50% ofthe battery drain. Additionally, thesecomponents, including the PC boardand all hardware, must usually fitwithin the LCD enclosure, with a heightrestriction of 0.25".

CCFL Power SuppliesAny discussion of CCFL power

supplies must consider lamp charac-teristics. These lamps are difficult loadsto drive, particularly for a switchingregulator. They have a “negative re-sistance” characteristic; the startingvoltage is significantly higher than theoperating voltage. Typically, the start-ing voltage is about 600 volts, althoughhigher and lower voltage bulbs arecommon. Operating voltage is usually300–400 volts, although other bulbsmay require different potentials. Bulbsize or length does not necessarilycorrelate to break-down voltage. Thebulbs will operate from DC, but migra-tion effects within the bulb will quicklydamage it. Hence, the waveform mustbe AC with no DC content.

Bulb operating frequencies aretypically 20 to 100kHz, and asine-like waveform is preferred. Thesine-like drive has low harmoniccontent, which minimizes RF emis-sions that can cause interference andefficiency degradation.

Figure 1’s circuit meets CCFL-driverequirements. Efficiency can be ashigh as 78% with an input voltagerange of 4.5V–20V. 82% efficiency ispossible if the LT1172 is driven from alow voltage (3–5V) source. Addition-ally, lamp intensity is continuouslyand smoothly variable from zero to full

Figure 2. CCFL power supply waveformsFigure 1. Cold-cathode fluorescent lamp power supply

by Jim Williams

A = 20V/DIV

B = 0.4V/DIV

C = 20V/DIV

D = 20V/DIV

E = 1000V/DIV

F = 5V/DIV

A AND B HORIZ = 10µs/DIVC THRU F HORIZ = 20µs/DIV

TRIGGERS FULLY INDEPENDENT

+

+

D2 1N4148

Q2 MPS650

1N5818

D1 1N4148

+

50kΩ INTENSITY

ADJUST

562Ω

10kΩ

1kΩ

LAMPC2 15pF 3kV 9 7

43215

Q1 MPS650

10µF

C1 0.02µF

+VIN

+VIN 4.5V TO +20V

VIN

VSW

VFB

VCGND

E2

E1

LT1172

5

7

3

2µF

1

8

6

2

1µF

L2 300µH

L1

CCFL_1.epsC1 = MUST BE A LOW LOSS CAPACITOR. METALIZED POLYCARB WIMA FKP2 (GERMAN) RECOMMENDED.L1 = SUMIDA-6345-020 OR COILTRONICS-CTX110092-1. PIN NUMBERS SHOWN FOR COILTRONICS UNITL2 = COILTRONICS-CTX300-4

Q1, Q2 = AS SHOWN OR BCP 56 (PHILIPS SO PACKAGE) DO NOT SUBSTITUTE COMPONENTS

SUMIDA (708) 956-0666 COILTRONICS (305) 781-8900


DESIGN FEATURES

intensity. When power is applied, theLT1172 switching regulator’s feedbackpin is below the device’s internal 1.23Vreference, resulting in full duty-cyclemodulation at the VSW pin (trace A,Figure 2). L2 conducts current (traceB), which flows from L1’s center tap,through the transistors, into L2. L2’scurrent is deposited in switched fash-ion to ground by the regulator’s action.

L1 and the transistors comprise acurrent-driven, Royer-class converter,which oscillates at a frequency deter-mined primarily by L1’s characteris-tics (including its load) and the 0.02µFcapacitor. LT1172 driven L2 sets themagnitude of the Q1–Q2 tail current,and hence L1’s drive level. The 1N5818diode maintains L2’s current flow whenthe LT1172 is off. The LT1172’s 100kHzclock rate is asynchronous with re-spect to the push-pull converter’s(60kHz) rate, accounting for trace B’swaveform thickening.

The 0.02µF capacitor combines withL1’s characteristics to produce sinewave voltage drive at the Q1 and Q2collectors (traces C and D, respec-tively). L1 furnishes voltage step-up,and about 1400VP–P appears at itssecondary (trace E). Current flowsthrough the 15pF capacitor into the

lamp. On negative waveform cycles,the lamp’s current is steered to groundvia D1. Positive waveform cycles aredirected, via D2, to the ground referred562Ω–50kΩ potentiometer chain. Thepositive half-sine appearing acrossthese resistors (trace F) represents 1/2the lamp current. This signal is filteredby the 10kΩ-1µF pair and presented tothe LT1172’s feedback pin. This con-nection closes a control loop whichregulates lamp current. The 2µF ca-pacitor at the LT1172’s VC pin providesstable loop compensation. The loopforces the LT1172 to switch-modemodulate L2’s average current to what-ever value is required to maintain aconstant current in the lamp. Theconstant current’s value, and hencelamp intensity, can be varied with thepotentiometer. The constant currentdrive allows full 0–100% intensity con-trol with no lamp dead zones or “pop-on” at low intensities. Additionally,lamp life is enhanced because currentcannot increase as the lamp ages.

Several points should be kept inmind when observing this circuit’soperation. L1’s high-voltage second-ary can only be monitored with a wide-band, high voltage probe fully specifiedfor this type of measurement. The vast

A Related Circuit:Helium Neon Laser Driver

The high-voltage-compliancecurrent-loop approach of the CCFLpower supply is suitable for otherapplications as well. Current sens-ing permits precise, high-efficiencycontrol of a wide variety of difficultloads. A HeNe laser is such a load.Lasers are negative impedances op-erating at very high voltages. Typi-cally, they require from 6 to 10kV tostart, with an operating voltage inthe 1 to 3kV range. Simple high-voltage drive does not provide this.The circuit in Figure A1 adapts theCCFL circuitry to control a laser.Both tube-current stability and elec-trical efficiency are improvedover the more conventional voltage-mode drive.

Q1 and Q2 combine with L2 toform a self-oscillating DC-DC con-verter. No resonating capacitor isused in this design. The DC-DC con-verter operates in square-wave mode,although resonant operation couldbe used to minimize harmonics. Theconverter’s high-voltage output ismultiplied by the D1-D2 voltage dou-bler and applied to the laser via the50kΩ ballast resistor. This resistoris required to isolate the laser fromthe 0.1µF output capacitor. Lasercurrent flow is converted to a voltageby the 340Ω shunt and is fed back tothe LT1074 step-down switchingregulator. D3, L1, and C1 smooththe LT1074’s pulsed output to DCand drive the DC-DC converter. Thiscompletes a loop that controls thelaser current. The components atthe LT1074’s VC pin stabilize theloop. The frequency compensation isarranged so transients (e.g., turn-on) will not cause a brief lowering ofthe tube current. Such an event willcause laser impedance to rise dra-matically, extinguishing the laser.

L2 and the voltage doubler sup-ply a maximum of about 3.5kV, notenough to start laser conduction.A1, L3, and associated componentsprovide a high voltage start-up.

continued on page 7Figure A1. Laser-driver power supply

Q2 D45

0.17 8

32146

Q1 D45

VIN

VSW

FB

VCGND

LT1074

L2

LASER = HUGHES 3121

+ C1 470µF

5

0.1µF+28V

47Ω

–

+

–

+

C2 2µF 300V

Z1 250V

10M

61.9k

1k

10M

LT1004 1.2V

340Ω

50k 5W

L1 = PE92108 – PULSE ENG.L2 = PE6197 – PULSE ENG.L3 = TRIAD PL-11Z1 = MOTOROLA 1.5KE-250

D1,D2 = SEMTECH SH-100

+28V

D3 MUR415

L1 100µH +28V

10k 100k

1N914 1N914

1N914+28V

A1 LT1018

C106

IK = 6.5mA

2k

D2 0.1µF

D1

L3

START RING

CCFL_A1.eps

2k

0.01µF

LASER

Q1,Q2 = MOTOROLA D45H11 OR EQUIVALENT

PULSE ENGINEERING (619) 268-2400

NOTE: THE START RING IS A RING OF WIRE AROUND THE LASER TUBE

DESIGN FEATURES


majority of oscilloscope probes willbreak down and fail if used for thismeasurement.1 Tektronix type P-6009(acceptable) or types P6013A andP6015 (preferred) probes must be usedto read L1’s output.

Another consideration involves ob-serving waveforms. The LT1172’sswitching frequency is completelyasynchronous from the Q1–Q2 Royerconverter’s switching. As such, mostoscilloscopes cannot simultaneouslytrigger and display all the circuit’swaveforms. Figure 2 was obtained us-ing a dual-beam oscilloscope (Tektronix556). LT1172-related traces A and Bare triggered on one beam, while theremaining traces are triggered on theother beam. Single-beam instrumentswith alternate sweep and trigger switch-ing (e.g., Tektronix 547) can also beused, but are less versatile and arerestricted to four traces.

Obtaining and verifying high effi-ciency requires some diligence.2 Theoptimum efficiency values given forC1 and C2 are typical, and will varyfor specific types of lamps. C1 setsthe circuit’s resonance point, whichvaries to some extent with the lamp’scharacteristic. C2 ballasts the lamp,effectively buffering its negative-resis-tance characteristic. Small values ofC2 provide the most load isolation, butrequire relatively large transformer out-put voltage for loop closure. Large C2values minimize transformer outputvoltage, but degrade load buffering.Also, C1’s “best” value depends some-what on the type of lamp used. BothC1 and C2 must be selected for givenlamp types. Some interaction occurs,but general guidelines are possible.

Typical values for C1 are 0.01 to0.047µF. C2 usually ends up in the10pF–47pF range. C1 must be a low-loss capacitor; substitution for theWIMA device is not recommended. Apoor-quality dielectric for C1 can eas-ily degrade efficiency by 10%. C1 andC2 are selected by trying different val-ues for each and iterating towardsminimum supply input current. Dur-ing this procedure, ensure that loopclosure is maintained by monitoringthe LT1172’s feedback pin, whichshould be at 1.23 volts. Several trialsusually produce the optimum C1 andC2 values. Note that the highest effi-ciencies are not necessarily associatedwith the most aesthetically pleasingwaveshapes, particularly at Q1, Q2,and the output.

Other issues influencing efficiencyinclude bulb wire length and energyleakage from the bulb. The high-volt-age side of the bulb should have thesmallest practical lead length. Exces-sive length results in radiative losses,which can easily reach 3% for a three-inch wire. Similarly, no metal shouldcontact or be in close proximity tothe bulb. This prevents energy leak-age, which can exceed 10%. (Theseconsiderations should be made withknowledge of other LCD issues. SeeAppendix B of AN49, “MechanicalDesign Considerations for LiquidCrystal Displays.”)

Special attention should be givento the layout of the circuit board,since high voltage is generated atthe output. The output couplingcapacitor must becarefully located tominimize leakagepaths on the circuitboard. A slot in theboard will furtherminimize leakage.Such leakage can per-mit current flowoutside the feedbackloop, wasting power.In the worst case, longterm contaminationbuild-up can increaseleakage inside theloop, resulting instarved lamp drive or

destructive arcing. To minimize leak-age, it is good practice to break thesilk-screen line which outlines trans-former T1. This prevents leakage fromthe high voltage secondary to theprimary. Another technique for mini-mizing leakage is to evaluate andspecify the silk-screen ink for its abil-ity to withstand high voltages.

Once these procedures have beenfollowed, efficiency can be measured.Efficiency may be measured by de-termining bulb current and voltage.Measuring current involves measur-ing RMS voltage across the 562Ωresistor (short the potentiometer). Thebulb current is

IBULB = (E/R) x 2

Multiplication by two is necessarybecause the diode steering dumps thecurrent to ground on negative cycles.Bulb RMS voltage is measured at thebulb with a properly compensated high-voltage probe. Multiplying these tworesults gives power in watts, whichmay be compared to the DC-input-supply E x I product. In practice, thelamp’s current and voltage contain

+

D2 1N4148

Q2 MPS650

1N5818

D1 1N4148

INTENSITY

ADJUST 1M

3.3k

1M

330Ω

LAMP

15pF 3kV 9 7

43215

Q1 MPS650

10µF

C1 0.01µF

+VIN

+VIN 2V TO 6V

VIN

SW1

FB

GNDSW2

SET

AO

LT1173

NC

0.01µF

L2 82µH

L1

CCFL_4.eps

C1 = MUST BE A LOW LOSS CAPACITOR. METALIZED POLYCARB WIMA FKP2 (GERMAN) RECOMMENDED.L1 = SUMIDA-6345-020 OR COILTRONICS-CTX110092-1. PIN NUMBERS SHOWN FOR COILTRONICS UNIT L2 = TOKO 262LYF-0091K DO NOT SUBSTITUTE COMPONENTS

1N5818

NC

ILIM

47Ω

1N4148

SHUTDOWN

Figure 3. Separate ballast capacitors allowone transformer to drive two tubes

CCFL_3.eps

BALLAST CAPACITORS CCFLs

TO D1, D2 JUNCTION

COILTRONICS CTX110459-1

TO DRIVE CIRCUITRY

Figure 4. Low-current CCFL power supply


DESIGN FEATURES

+

3V 2 AA CELL

R1 100

OPERATE SHUTDOWN

* TOKO 262LYF-0092K

D4 1N4148 C3

22µF

R2 120k

R4 2.21M

C1 0.1µF

C2 4.7µF

D3 1N5818

D2 1N5818

OUTPUT –12V TO –24V

L1* 100 Hµ

CCFL_6.eps

U1 LT1173

ILIM VIN

SW1

FBSW2GND

×

D1 1N5818

R3 100k

+

AN49, “Achieving Meaningful Effi-ciency Measurements.”

Two-Tube DesignsSome displays require two tubes

instead of the more popular single-tube approach. These two-tube designsusually require more power. Accom-modating two tubes involves separateballast capacitors (see Figure 3), butcircuit operation is similar. Higherpower may require a different trans-former rating. Figure 1’s transformercan supply 7.5mA, although morecurrent is possible with appropriatetransformer types. For reference, an11mA-capability transformer appearsin Figure 3.

Low-Power CCFL SupplyFigure 4 represents the other ex-

treme. This design is optimized forsingle-tube operation at very low cur-rents. Figure 1’s circuit typically drives5mA maximum, but this design topsout at 1mA. This circuit maintainscontrol down to tube currents of 1µA,a very dim light. It is intended for ap-plications where the longest possiblebattery life is desired. Maintaininghigh efficiency at low tube currentsrequires modifying the basic design.

Achieving high efficiency at lowoperating currents requires loweringFigure 1’s quiescent power drain. Todo this, the LT1172, a pulse-width-modulator-based device, is replaced

with an LT1173. The LT1173 is aburst-mode-type regulator. When thisdevice’s feedback pin is too low, itdelivers a burst of output currentpulses, putting energy into the trans-former and restor-ing the feedbackpoint. The regulator maintains controlby appropriately modulating the burstduty cycle. The ground-referred diodeat the SW1 pin prevents substrateturn-on due to excessive L2 under-shoot. During the off periods, the regu-lator is essentially shut down. Thistype of operation limits available out-put power, but cuts quiescent currentlosses. In contrast, Figure 1’s LT1172pulse-width-modulator-type regulatormaintains “housekeeping” current be-tween cycles. This results in moreavailable output power but higher qui-escent currents. Figure 5 shows oper-ating waveforms for the circuit in Figure4. When the regulator comes on (traceA) it delivers bursts of output currentto the L1–Q1–Q2 high-voltage con-verter. The converter responds withbursts of ringing at its resonant fre-quency. The circuit’s loop operation issimilar to that of Figure 1.3

LCD Bias SuppliesLCDs also require a bias supply for

contrast control. The supply’s variablenegative output permits adjustmentof display contrast. Relatively littlepower is involved, easing RF radiationand efficiency requirements. The logicsections of display drivers operate fromsingle 5V supplies, but the actual driveroutputs swing between +5V and anegative bias potential. Varying thisbias varies the contrast of the display.

An LCD bias generator, developedby Steve Pietkiewicz of LTC, is shownin Figure 6. In this circuit U1 is anLT1173 micropower DC-DC converter.The 3V input is converted to +24V byU1’s switch, L1, D1, and C1. Theswitch pin (SW1) then drives a chargepump composed of C2, C3, D2, and D3to generate –24V. Line regulation isless than 0.2% from 3.3V to 2.0Vinputs. Although load regulation suf-fers somewhat because the –24V out-put is not directly regulated, it stillmeasures 2% from a 1mA to 7mA load.

Figure 5. Low-current CCFL power-supplywaveforms

Figure 6. LCD bias generator (3V to -12 to -24V)

small out-of-phase components, buttheir error contribution is negligible.

Both the current and voltage mea-surements require a wide-band true-RMS voltmeter. The meter must employa thermal-type RMS converter—themore common logarithmic-computingtype instruments are inappropriatebecause their bandwidths are too low.(See AN49 for a discussion of the op-erational theory and limitations of vari-ous AC voltmeters.)

The previously recommended highvoltage probes are designed to see a1MΩ, 15–22pF oscilloscope input. TheRMS voltmeters have a 10MΩ input.This difference necessitates an im-pedance matching network betweenthe probe and the voltmeter. Detailsof this and other efficiency measure-ment issues appear in Appendix C of

A = 5V/DIV

B = 5V/DIV

50µs/DIV

DESIGN FEATURES


L1 = SUMIDA CD75-470M C1, C2 = TANTALUM CCFL_7.eps

LT1172

VC

VIN

FB

GND

VSW

+D1 1N914

R3 15kΩ

C5 2µF

D2 1N5817

VIN = 5V

VBATT = 3V TO VOUT + 1V

R2 100kΩ

R1 200kΩ

+ C1 2µF

VOUT = –10V TO –30V

C2 4.7µF

D3 1N5817

C3 4.7nFC4

4.7nFQ1 VN2222LL

L1 47µH

OPTIONAL SHUTDOWN

+

The circuit will deliver 7mA from a2.0V input at 75% efficiency.

If greater output power is required,Figure 6’s circuit can be driven from a+5V source. R1 should be changed to47Ω and C3 to 47µF. With a 5V input,40mA is available at 75% efficiency.Shutdown is accomplished by bring-ing the anode of D4 to a logic high,forcing the feedback pin of U1 to goabove the internal 1.25V referencevoltage. Shutdown current is 110µAfrom the input source and 36µA fromthe shutdown signal.

Another interesting modification ofa boost converter that can providenegative bias from a 5V supply is shownin Figure 7. The converter, developedby Jon Dutra of LTC, is half switcherand half charge pump. The chargepump (C1, C2, D2, and D3) is driven bythe flying node at VSW. The output isvariable from –10 to –30V, providingcontrast control for the display.

On low-voltage supplies (6V or less)VIN and VBATT can be tied together.With higher battery voltages, high effi-ciency is obtained by running theLT1172 VIN pin from 5V. Shutting offthe 5V supply will automatically turnoff the LT1172. The maximum valuefor VBATT is equal to the negative out-put plus 1V. Also, the difference be-tween VBATT and VIN must not exceed16V. R1, R2, and R3 are made large tominimize battery drain in shutdown,since they are permanently connectedto the battery via L1 and D1. Efficiencyis about 80% at IOUT = 25mA.

When power is applied, laser currentis zero and A1’s output goes toground. The high-voltage convertersupplies full output ( 3.5kV), but thelaser does not start. Under theseconditions, the C106 SCR is off andC2 charges. A divided-down portionof C2’s voltage biases A1’s positiveinput. When C2 reaches about 200V,A1’s output goes high, triggering theSCR. This dumps C2’s chargethrough L3 to ground. L3’s output, avery-high-voltage spike, ionizes the

laser, driving it into conduction. La-ser impedance drops and the maincurrent control loop begins be-having as described above. Thepresence of voltage across the 340Ωshunt causes A1’s output to go low,preventing further SCR conduction.Zener clamp Z1 limits C2’s voltageunder these conditions. If the laserfails to start, the cycle will repeatuntil start-up occurs. Electrical effi-ciency for this circuit is about 75%for inputs from 22 to 35V.

Laser Driver, continued from page 4

3 The discontinuous energy delivery to the loopcauses substantial jitter in the burst-repetitionrate, although the high-voltage section maintainsresonance. Unfortunately, circuit operation is in the“chop” region of most oscilloscopes, precluding adetailed display. “Alternate” operation causes wave-form-phasing errors, also producing an inaccuratedisplay. Hence, waveform observation requires spe-cial techniques. Figure 5 was taken with a dual-beam instrument (Tektronix 556) with both beamsslaved to one time base. Single-sweep triggeringeliminated jitter artifacts. Most 'scopes, whetheranalog or digital, will have trouble reproducing thisdisplay.

1 Don’t say we didn’t warn you!

2 The term “efficiency” as used here refers to electri-cal efficiency. In fact, the ultimate concern centersaround the efficient conversion of power-supplyenergy into light. Unfortunately, lamp types showconsiderable variation in their current-to-light con-version efficiency. Similarly, the emitted light for agiven current varies over the life of any particularlamp. Hence, this article treats “efficiency” on anelectrical basis: as the ratio of power removed fromthe primary supply to the power delivered to thelamp. When a lamp has been selected, this ratio canbe measured with a photometer.

Figure 7. 5V LCD bias generator (5V to -10 to -30V)

This article was extracted from LTCApplication Note 49 available inAugust. For more details, andadditional information, please requestAN49 on the response card or call the800 number listed on the last page ofthis magazine.


DESIGN FEATURES

The LTC1153 Electronic Circuit Breakerby Tim Skovmand

The LTC1153 electronic circuitbreaker is designed to work with alow-cost, N-channel power MOSFETto interrupt power to a sensitiveelectronic load in the event of an over-current condition. Under normaloperating conditions, the voltage dropacross the switch is extremely low,limited primarily by the RDS(ON) of theMOSFET switch. The breaker is trippedby an over-current condition and re-mains tripped for a period of timeprogrammed by an external timingcapacitor, CT. The switch is thenautomatically reset and the load mo-mentarily retried. If the load current isstill too high, the switch is shutdownagain. This cycle continues until theover-current condition is removed,thereby protecting the sensitive loadand the power MOSFET.

The gate voltage for the high-side MOSFET N-channel switch isgenerated completely on-chip by ahigh-frequency charge pump. No ex-ternal charge-pump components arerequired. Further, the charge-pumpcircuitry is designed for maximum effi-ciency with an operating supplycurrent of only 85µA when poweredfrom a 5V supply. Supply current dropsto 8µA when the circuit is switched tothe standby mode (input low).

Figure 2. Trip-delay time versus circuit-breaker current (1ms RC time constant)for circuit of Figure 1

Programmable TimingThe trip current, trip-delay

time, and auto-reset period are pro-grammable over a wide range toaccommodate a variety of load imped-ances. Figure 1 demonstrates how theLTC1153 is used in a typical circuit-breaker application. The DC tripcurrent is set by a small-valued resis-tor, RSEN, in series with the drain lead,which drops 100mV when the currentlimit is reached. In the circuit ofFigure 1, the DC trip current is set at1A (RSEN = 0.1Ω).

The trip-delay time is set by the twodelay components, RD and CD, whichestablish an RC time constant in se-ries with the drain sense resistor, pro-ducing a trip delay which is shorter forincreasing breaker current (similar tothat of a mechanical circuit breaker).Figure 2 is a graph of the trip-delaytime versus the circuit-breaker cur-rent for a 1ms RC time constant. Notethat the trip time is 0.63ms at 2A, butfalls to 55µs at 20A. This characteris-tic en-sures that the load and theMOSFET switch are protected againsta wide range of overload conditions.

Auto-Reset FunctionThe auto-reset time is typically set

in the range of 10s of milliseconds to afew seconds by selecting the timingcapacitor, CT. The auto-reset periodfor the circuit in Figure 1 is 200ms,i.e., the circuit breaker is automati-cally reset (retried) every 200ms untilthe overload condition is removed. Theswitch then returns to normal opera-tion and continues to power the loaduntil another fault condition is en-countered.

An open-drain fault output is pro-vided to warn the host microprocessorwhenever the circuit breaker has beentripped. The microprocessor can ei-ther wait for the auto-reset function toreset the load, or shut the switch OFFafter a fixed number of retries.

The shutdown input interfaces di-rectly with a PTC thermistor to senseover-temperature conditions and tripthe circuit breaker whenever the loadtemperature or the MOSFET-switchtemperature exceeds a safe level. Thethermistor shown in Figure 1 trips thecircuit breaker when the load tem-perature exceeds approximately 70°C.The breaker is then automatically re-set when the load temperature fallsback to a safe level. A small amount of

Figure 1. LTC1153 5V/1A circuit breaker with thermal shutdown

1153_1.eps

SENSITIVE 5V LOAD

TO µP

ON/OFFCT

0.22µF, Z5U

IN

CT

FLT

GND

VS

DS

G

SD

CD 0.01µF

RD 100k

RSEN* 100mΩ

IRLR024

70°C** PTC

51k

LTC1153

5V

51k

5V

ALL COMPONENTS SHOWN ARE SURFACE MOUNT. IMS026 INTERNATIONAL MANUFACTURING SERVICE, INC. (401) 683-9700 RL2006-100-70-30-PT1 KEYSTONE CARBON COMPANY (814) 781-1591

* **

CIRCUIT BREAKER CURRENT (A)

10.01

TRIP

DEL

AY (m

s)

0.1

1

10

5 10 20 100

1153_2.eps

2 50

RSEN = 0.1Ω RD = 100k CD = 0.01µF

DESIGN FEATURES


hysteresis is provided to ensure cleanswitching. Standby efficiency is main-tained by powering the PTC thermistorfrom the switch output, as shown inFigure 1.

Surface-Mount PackagingAll the components shown in Fig-

ure 1, including the LTC1153 and thedrain sense resistor, are available insur-face-mount packaging and, there-fore, consume a very small amount ofprinted circuit board space. The totalpower dissipation of this circuit is lessthan 100mW when powering a nomi-nal 5V, 0.5A load. This efficiency is duein part to the extremely low drop acrossthe MOSFET switch and the drain-sense resistor (50mV each), and by theextremely low quiescent current re-quired by the LTC1153 charge pumpand protection circuitry.

Additionally, power dissipation doesnot rise appreciably during a currentoverload condition. This is becausethe switch is only engaged for a veryshort period of time (<50µs) every200ms, i.e., an extremely low dutycycle. This ensures that the MOSFETis never required to operate in a high

dissipative mode, even with the outputshorted.

Figure 3 is a timing diagram withsome typical waveforms generated bythe circuit breaker in the normal oper-ating mode, the overload mode, andthe shutdown mode. Note that thetiming capacitor, CT, is held low until afault condition is encountered andthen charged by a small internal cur-rent source until the threshold isreached and the is switch turned backON. This cycle continues until theoverload is removed and the switchreturns to normal operation.

SCSI Termination PowerThe termination power for a SCSI

interface is protected to avoid damag-ing the drivers, the connectors, and theprinted circuit board in the event of theconnector or the interconnecting cablebeing shorted. This protection is pro-vided by the circuit breaker circuitshown in Figure 4. With the componentvalues shown, the DC current is limitedto 1A with a trip-delay time constant of10ms. The breaker will trip if the cableor connector is accidently shorted andwill retry every second until the short

circuit is removed. The terminationpower will then return to normal andthe interface will be reconnected. Themicroprocessor can continuously moni-tor the status of the termination powervia the fault flag output of the LTC1153and can take further action if the faultcondition persists.

DC Motor ProtectionA 5V DC motor can be powered and

protected using the circuit shown inFigure 5. The DC current delivered tothe motor is limited to 5A and a ratherlong trip delay is used to ensure thatthe motor starts properly. The motortemperature is also continuously moni-tored and the breaker is tripped if themotor temperature exceeds 70°C. Thefault output of the LTC1153 informsthe host microprocessor whenever thebreaker is tripped. The microproces-sor can disable the motor if a setnumber of faults occur or it can ini-tiate a retry after a much longer periodof time has elapsed. A rectifier diodeacross the motor returns the motorcurrent to ground and restricts theoutput of the switch to less than 1Vbelow ground.

1153_4.eps

ON/OFF

CT 1µF

IN

CT

FLT

GND

VS

DS

G

SD

CD 0.01µF

RD 100k

RSEN 0.1Ω

IRLR024

LTC1153

5V

51k

5V

µP

FAULT

PROTECTED TERMINATION

POWER

1N5817

Figure 4. SCSI termination power circuitry

1153_5.eps

TO µP

ON/OFFCT

0.47µF, Z5U

IN

CT

FLT

GND

VS

DS

G

SD

CD 0.22µF

RD 100k

RSEN 20mΩ

IRLR024

70°C PTC

51k

LTC1153

5V

51k

5VD.C. 1N5400

Figure 5. DC motor driver with over-current andover-temperature protection

Figure 3. LTC1153 typical timing diagram

1153_3.eps

OFF NORMALOVER-CURRENT (AUTO-RESET) NORMAL SHUT-DOWN OFF

INPUT

OUTPUT (MOSFET SOURCE)

FAULT

TIMING CAP

SHUT-DOWN

OVERLOAD OVERLOAD REMOVED

* TIMES FOR COMPONENTS SHOWN IN FIGURE 1.

50µs*

200ms*


DESIGN FEATURES

8

7

6

54

3

2

1VIN

LTC1063

– 5V

1063_1.eps

CLOCK OUT

VOUT

+5V

R

0.1µF

C = 200pF (TYPICAL)

0.1µF

1 RCfCLOCK

8

7

6

54

3

2

1VIN

LTC1063

–5V

0.1µF

1063_2.eps

VOUT

+5V

R0.1µF

C

8

7

6

54

3

2

1VIN

LTC1063

– 5V

O.1µF

VOUT

+5V

0.1µF

TO OTHER LTC1063s

The LTC1063 is the first monolithiclowpass filter simultaneously offeringoutstanding DC and AC characteris-tics, internal or external clocktunability, cutoff frequencies to 50kHz,1 millivolt typical output DC offset,and a dynamic range in excess of twelvebits over a decade of input-voltagerange. The LTC1063 approximates afive-pole Butterworth polynomial. Afive-pole linear phase version of thisfilter will be offered soon.

AC CapabilitiesThe unique internal architecture of

the filter allows outstanding ampli-tude matching from device to device.Typical matching ranges from 0.01dBat 25% of the filter passband to 0.05dBat 50% of the filter passband.This capability is important for multi-channel data-acquisition systems,where channel-to-channel matchingis critical.

An external or internal clock pro-grams the filter’s cutoff frequency.The clock-to-filter cutoff-frequencyratio is set to 100:1. In the absence ofan external clock, the LTC1063’s in-ternal precision oscillator can be used.An external resistor and capacitor setthe LTC1063’s internal clock frequency(Figure 1). Note that unlike the popu-lar DC-accurate LTC1062 Butterworthfilter, the internal clock frequency tol-erance of the LTC1063 does not affect

the flatness of its passband. The inter-nal oscillator output is brought out atpin 4 so that it can be used as a syn-chronized master clock to drive otherLTC1063s. Ten or more LTC1063smay be locked together to a singleLTC1063 clock output (see Figure 2).

The LTC1063 has both low noiseand very low clock feedthrough. Thewide-band noise and clockfeedthrough are both measured withthe input of the filter grounded, asillustrated in Figure 3. The wide-bandnoise is the integral of the noise spec-tral density; it is usually expressed inµVRMS. The wide-band noise is virtuallyindependent of filter cutoff frequency.The LTC1063 has an excellent noisespecification of 90µVRMS. This num-ber is clock-frequency and power-sup-ply independent.

Some users prefer to describe wide-band noise in dBV/(Hz)1/2. Thesedesigners limit their measurementsto noise in the filter’s passband. Inthis case the filter’s cutoff frequencymust be known. The method for cal-

culating noise in dBV/(Hz)1/2 is illus-trated in Figure 4. Remember that thearea in the rectangle is an approxima-tion of the measured noise contribu-tion and must be treated as such. Alsonote that an output buffer is not nec-essary when measuring noise and clockfeedthrough, but it is good practice touse it when evaluating the filter’s dy-namic range.

Dynamic rangeThe LTC1063’s AC design is based

on optimum dynamic range ratherthat just wide-band noise. Dynamic-range measurements take into theaccount the device’s total harmonicdistortion. Figure 5a shows the typi-cal connection for dynamic range mea-surement. An inverting buffer ispreferred over a unity gain follower.Large-input common-mode signalscan severely degrade the distortionperformance of a non-inverting buffer.It is also important to make sure theundistorted op-amp swing is equal toor better than that of the filter. Figure5b shows the device’s operating dis-tortion plus noise versus input-signalamplitude measured with a standard1kHz pure sine wave input. The per-formance improves with increasedpower supply voltage. This improvedperformance at higher power supply

The LTC1063 Low-Offset, Clock Tunable,Fifth-Order Butterworth Lowpass Filter

The LTC1063’s Unique Features• Trivial to use

• Precision internal clock

• S/N ratio ≥ 90dB at 0.02% THD level

• Better than 80dB noise+THD,0.5VRMS <VIN <2.5VRMS

• VOS ≤ 1 millivolt (typical)

• Clock tunable without VOS change

• Excellent low-frequency amplitudematching ±0.01dB at fIN = 0.2fC

by Nello Sevastopoulos

Figure 1. Setting the LTC1063 internal clockwith an external R,C

Figure 2. Synchronizing multiple LTC1063s

DESIGN FEATURES


8

7

6

54

3

2

1VIN

LTC1063

V –

1063_5a.eps

CLOCK IN

VOUT

V+

0.1µF0.1µF

–

+

50kLT1022

+

–50k

20pF

Power-supply decoupling and PC-board layout are extremely critical toachieve a constant output offset overa wide range of clock frequencies.Ideally the power-supply pins shouldbe decoupled with good quality ce-ramic capacitors exhibiting low ACimpedance over a wide range of fre-quencies. If a single capacitor cannotbe used, two or more capacitors ofdifferent values should be connectedin parallel.

NOIS

E

FREQUENCY fCUTOFF = 10kHz

APPROXIMATION

= APPROXIMATION TO NOISE SPECTRUM = ADDITIONAL CONTRIBUTION FOR MEASURED NOISE SPECTRUM

EXAMPLE: LET ∫ fCUTOFF = 10kHz

= 100µVRMS

1063_4.eps

χµV

(Hz)1/2x √10kHz = 100µVRMS

1µVRMS

(Hz)1/2= –120dBV

(Hz)1/2

THEN

OR

Figure 3. Measuring LTC1063 noise and spurious signals Figure 4. Calculating wideband noise in dBV/(Hz)1/2

Figure 5a. Typical connection for measuring Distortion + Noise and Signal to THD + Noise ratio

1063_5b.eps

VIN (VRMS)

0.001

% T

HD +

NOI

SE

0.01

0.1

1

0.1 1 5

VSUPPLY = ±5.0V

VSUPPLY = ±7.5V

DISTORTION + NOISE (fC = 20kHz, fIN = 1kHz)

8

7

6

54

3

2

1 LTC1063

V–

1063_3.eps

VOUT

V+

0.1µF0.1µF

SPECTRUM ANALYZER

fCLOCK = 1MHz, fC = 10kHz

nV/ √

/Hz

1 10k 100k 1MHz 10M

CLOCK FEEDTHROUGH = FUNDAMENTAL CLOCK FREQUENCY + ODD HARMONICS

WIDEBAND NOISE

FREQUENCY (Hz)

LTC1063: WIDEBAND NOISE = 90µVRMS TO 100µVRMS (VS = ±2.5V TO ±8V) CLOCK FEEDTHROUGH = 50µVRMS (VS = ±5V) = 75µVRMS (VS = ±7.5V)

voltages is possible because theLTC1063’s noise does not increasewith higher supply voltages.

Figure 5b illustrates how the filtercan handle inputs to 4VRMS (11.2VP–P)with less than 0.02% THD. At thisinput level, the dynamic range is onlylimited by distortion and not by wide-band noise. The signal-to-noise ratioat 4VRMS input is 93dB. Optimumsignal-to-noise plus distortion accord-ing to Figure 5b is 83dB, yet a com-

Figure 5b. Plot of Distortion + Noise vs VIN

fortable 80dB (0.01%) is achieved forinput levels between 1VRMS and2.4VRMS.

DC PerformanceThe LTC1063’s output DC offset

voltage is optimized for ±5V supplyapplications. Output offset is lowenough to compete with discrete-typeR,C active filters using low-offsetop amps. These discrete filters canprovide only one cutoff frequency.

continued on page 17


DESIGN FEATURES

It’s 8:00 am at ACME Instrumenta-tion and Handgun, Inc. I’m sittin’here, feet up, suckin’ on a cup of java,tryin’ to chase away the excesses oflast night. Gotta stop hanging outwith Johnny... But enough moaning.Let’s talk about me. I’m Rex Linear,manager of analog design. You cancall me Mr. Linear. Nice digs, eh?These corner offices give you a realpanoramic view of the parking lot. Letme tell you how I got here...

Big Trouble in Silicon ValleyIt was a dark and stormy night. I

was holed up in the lab, trying to tunean 8th-order elliptic filter made fromOP27s and old chewing gum. I’d justgot the first notch to stop oscillatingwhen the boss came in. Apparentlywe’re competing with Very Large Cor-poration for a new contract from Smok-ing Gun Security, Inc. He handed me astrange looking transducer with a dif-ferential output. He tells me it cansense body heat at 100 yards, but itneeds a high-input-impedance instru-mentation amplifier with 16-bit DCperformance. Not only that, but thewhole ball o’ wax has to run on batter-ies. The goons at Very Large Corp. saythey’ve already perpetrated a workingprototype, so I’d better get right on it orhe’ll have me scrubbing toilets until Iretire. Sixteen bits DC? Running onbatteries? I often wonder how he got tobe the boss; maybe it’s his sparklinginterpersonal skills...

LTC Mag Saves the DayNow, normally, I would have started

reading up on porcelain cleaners, butthat night I was up late reading thenew issue of the LTC mag with my pal,Johnny. Johnny Walker Black Label.That’s Mr. Walker to you, pal. I waslooking for the latest Jim Williamsarticle when I stumbled across thisstory: “LTC1047: New Dual Micropower

Zero-Drift Op Amp” by some guy namedDooley. As I glanced through it, thevisions of endless plumbing fixturesstarted to fade from my fogged brain. Iwas just getting to the good part whenJohnny hit me in the head. Still hurts...

Between aspirins the next day, I dida little checking around, and foundout some more about this part theycall LTC1047. It’s a zero-drift (all right,chopper-stabilized) dual op amp, basedon the same architecture as the popu-lar LTC1050/51 parts. It uses internalsample-and-hold capacitors, unlikesome competing chopper amps whichrequire external capacitors. Becausethe 1047 requires no external parts, itfits the standard dual op-amp pinout,and comes in a standard 8-pin plasticDIP or a 16-pin 300mil SO package(those caps make the die way too bigfor the SO8). It’s optimized for lowsupply current by squeezing each in-ternal section until it starts to scream,then backing off a little. The result is10µV VOS with 50nV/°C drift, 105dBminimum CMRR and PSRR, 120dBminimum DC gain, and 200kHz gain-bandwidth product, all at 60µA/side,typical. The only trade-offs are frontend noise and output drive, both pa-rameters that depend heavily on sup-ply current. Noise is 3.5µVP–P, roughlytwice that of the LTC1050. The 1047’soutput will sink 5mA and source 1mA,three times less than the 1050; it willswing to within 5mV of either rail witha 100k load.

LTC1047 ArchitectureInternally, the 1047 uses the same

architecture as the rest of the LTCchopper line, with a wide-band ampli-fier in parallel with a nulling amplifier(see Figure 1, over there next to thecoffee stain). The nulling amp alter-nately zeros its own output, thencorrects the wide-band amplifier. Thewide-band amplifier is always actively

1047_1.eps

–

+

–

+

C2

C1

φ1

φ2

N

N

–IN

+IN

φ2

φ1

OUT

WIDEBAND AMP

NULLING AMP

Figure 1. LTC1047 Block diagram

The LTC1047: New Dual MicropowerZero-Drift Op Amp by Dave “wish I wrote

movie scripts” Dwelley

connected to the output and hassubstantial bandwidth at the internalclock frequency, allowing its feedbackloop to correct most of the clock-feedthrough and aliasing errors causedby the nulling amp. As a result, the1047 has output-clock and feedthroughlevels far better than those of tradi-tional choppers, micropower or not.

Soldering Iron Memories andMicropower Applications

I called up my bookie and put $10on “Starlight Lounge” to show. Then Icalled up the rep and had him bring bya couple of these LTC1047s, scratchedout a circuit on the back of an overduebill (Figure 2a), and gave it to my side-kick Rachel. Rachel... I remember theday when our paths first crossed. I wasin the lab, sweating over a particularlynasty switching-regulator breadboard.I’d just sent another power MOSFET tothe great semiconductor in the sky,and was considering calling my oldpals, Smith & Wesson, to really fix thisbreadboard. The stench of burningrosin was thick in the air when out ofnowhere she appeared, packing a 15WRadio Shack iron. She tacked a 0.22µF

DESIGN FEATURES


cap across the snubber network, andhad that circuit singing like a paroleviolator in no time. I offered her a newWeller soldering station as a token ofgratitude, but she turned me down,mumbling something like “all in aday’s work.” The world needs morewomen like her...

Anyhow, I figured she’d be eitherfighting noise or loading the trans-ducer with those input resistors, butshe came up with this slick IA circuitwhose inputs eat only the pA-level biascurrent of the 1047 (Figure 2b). Notonly that, it eats 50% less current thanthe three-amplifier configuration, andonly uses one 8-pin DIP per trans-ducer. I should pay her more...

Sampling IA—Better Specs,Less Juice

That next week, I was drinking lunchwith the local distributor and my oldnemesis Rocco, the government in-spector. The disti-shark was telling meabout an improvement to the LTC1043switched-capacitor building block; saidhe’s got samples of a new lower powermask set. He slipped me one whenRocco wasn’t looking. I thought aboutit for a minute, and the lightbulb wasjust lighting up over my head when thewaitress showed up. I ordered a screw-driver (hold the OJ) and startedscratching on the napkin. Now, if Icombine this 1043 with a 1047, throwin a resistor or two here... let’s see...Ohm’s law is I = ZV, no, E = I/V, no,that’s not right...

When I got back to the office, I threwout the napkin (couldn’t read it any-how after the Rocco spilled his drinkon it), and sat down at the computer.Let’s see... just hook up pin 3 to node 5

and...what’s this? “Unable to find biaspoint for transient analysis?” Damnthese simulations! Gonna have tobreadboard this thing.

Sampling IA PerformanceI put Rachel on the case. She

whipped up a breadboard that wouldmake a Linear Technology technicianjealous, and came back with the num-bers (Figure 3). Numbers like 120dBCMRR at 60Hz, 10µV total system VOS,only 60µA supply current for two chan-nels with a single 5V supply, andPittsburgh down by six in the thirdquarter. I took these numbers andmade a couple of calls. Then I startedthinking about circuits. This one usesthe 1043 switched-capacitor block tocreate a sampling front end that givesexceptional AC common-mode rejec-tion, as well as rail-to-rail input com-mon mode range. The 1047 buffers the1043’s output, providing gain withminimum DC error. The 1047 also

provides rail-to-rail output swing intothe system’s A/D input, allowing theguys doing the digital design to use allof the A/D’s bits. All this performancewith less current draw than the self-discharge of the battery pack—thisshould earn me a few points with theguys upstairs.

The Happy EndingAnyhow, the sensor was a big suc-

cess; we beat Very Large Corporation tothe market by seven months, endedup selling millions of those things,and Pittsburgh pulled it out to win inthe fourth, 21-20. The boss took allthe glory for that 1047 design, reallyrode it for a while until one of those1047s caught him sneaking in theback door of the Pussycat Club down-town. Now I’m the boss, and Rachel’sdesigning circuits that make thosecollege grads at Very Large Corp. lookbad. So, let me tell you about the timeJohnny and me went fishing...

–

+

–IN

+IN

R1

R1

R2

R2

2

3

+5V

–5V

1/2 LTC1047

8

4

1 OUT

1047_2a. eps

GAIN = R2 R1

CMRR = RESISTOR LIMITED

–

+

–IN

R2 2

3

–5V

1/2 LTC1047

4

1

1047_2b. eps

–

+

R1

R2

6

5

+5V

1/2 LTC1047

8

7 OUT

+IN

R1

GAIN = R2 R1

CMRR = RESISTOR LIMITED

Figure 2a. Rex's IA circuit Figure 2b. Rachael's IA circuit

1047_3. eps

1/2 LTC1047

+5V

VOUT

R1

1/2 LTC1043

–IN

+IN

+5V

0.01µF

VOS = 10µV ∆VOS = 50nV/°C IS = 30µA/SIDE CMRR > 120dB AT 60Hz SINGLE 5V SUPPLY INPUT RANGE INCLUDES BOTH SUPPLIES

11

12

1413

16 17

4

87

CS = 1µF CH 1µF

R2

1µF

+

–

R2 R1

GAIN = 1 +

Figure 3. The final IA circuit


DESIGN FEATURES

OUTPUT CURRENT (mA)

00

DROP

OUT

VOLT

AGE

(V)

0.1

0.2

0.3

0.4

0.5

40 80 120 160

1121_1.eps

20 60 100 140

INPUT VOLTAGE (V)

00

INPU

T CU

RREN

T (µ

A)

50

100

150

200

250

2 5 7 10

1121_2.eps

1 4 6 93 8

ILOAD = 100µA

ILOAD = 0

IntroductionAn ideal low-dropout regulator

should have zero dropout voltage, zeroquiescent current, and a stable, regu-lated output voltage for a wide range ofloads. The previous architecture ofchoice was the PNP-transistor typelow-dropout regulator supplied bymany manufacturers. This architec-ture has many deficiencies when ap-plied to micropower operation, rangingfrom minor inconveniences to majorsystem problems. These problems in-clude large increases in ground-pincurrent when the regulator approachesits dropout voltage, and the require-ment of large output capacitors forfrequency compensation.

A new micropower low-dropoutregulator, the LT1121, provides over150 milliamps of output current, yetoperates with less than 30 microampsof quiescent current (no load). Drop-out at high output currents (150mA) isonly 0.4 volts. (Dropout voltage versusoutput current is shown in Figure 1.)Quiescent current does not rise appre-ciably when the device enters the drop-out region. The LT1121 is available infixed output voltages of 5V and 3.3V.The device is designed to be stable withoutput capacitors as low as 0.33µF.

New FeaturesAn important feature unavailable

until now in low-dropout regulators isthe ability to reverse or decrease theinput voltage below the output with-out reverse current flow. This is im-portant in battery powered circuits,which are becoming increasinglypopular. The new LT1121 regulatoroperates as if an ideal diode wereconnected in series with its input. Thedropout voltage is equal to the collector-

The LT1121 Low-Dropout Regulatorby Dennis O’Neill

device enters the dropout region whichis exhibited by other PNP-type low-dropout regulators. This featureextends the useful life of the battery, amost important consideration intoday’s battery-powered world.

The design of low-dropout regula-tors requires a capacitor on the out-put of the regulator for frequencycom-pensation. Careful design of theAC performance characteristics of theLT1121 allows the use of outputcapacitors as low as 0.33µF in com-parison to tens of microfarads on olderregulators. Figure 3 shows the basicLT1121 circuit. The output capacitordoes not need to have extremelylow effective series resistance (ESR).Transient response will, of course,suffer somewhat with smaller outputcapacitors. This is because the overallfeedback loop is relatively slow andthe output capacitor is required toprovide current into transient loadsfor several microseconds until the loopresponds. The LT1121’s ability to op-erate with small-value output capaci-tors is a big advantage in systemswhere transient output currents aresmall and space is limited, as in manybattery operated circuits.

to-emitter saturation voltage of the PNPoutput transistor, but there is no re-verse current flow from output to in-put. The output voltage may be heldhigh, perhaps by a battery-backuppower system, while the input is pulledto ground or is even reversed. Therewill be no current flow from the outputpin back to the input. The outputstage of the LT1121 draws only 15µAunder these conditions. If used in thistype of application, currently avail-able regulators draw large reverse cur-rents and quickly drain the backupbattery.

The control circuit of the LT1121allows the device to operate with only30µA quiescent current (no load) whilesupplying up to 150mA of outputcurrent. Figure 2 shows the quiescentcurrent versus input voltage charac-teristics of the device. The LT1121incorporates anti-saturation circuitrythat controls the drive current of theoutput transistor. This prevents theincrease in quiescent current as the

The LT1121’s ability tooperate with small-valueoutput capacitors is a big

advantage in systems wheretransient output currents

are small and space islimited, as in many battery

operated circuits.

Figure 1. Dropout voltage versus outputcurrent for the LT1121

Figure 2. Quiescent current versus inputvoltage for the LT1121

DESIGN FEATURES


OUTLT1121-5

GND

1121_3 . eps

IN

SHUTDOWN

6V+

5V OUT 150mA

OPEN COLLECTOR GATE SHUTDOWN = INPUT HIGH

3

18

5

5MΩ

0.33µF

OUTPUT CURRENT (mA)

00

GROU

ND P

IN C

URRE

NT (m

A)

2

4

6

8

12

25 75 100 150

1121_4.eps

50 125

10

VIN - VOUT ≤ DROPOUT

VIN - VOUT = 5V

1121_5.eps

SHUTDOWN

Q1

INPUT

OUTPUT

GROUND

–

+

Q2

START-UP, BIAS CURRENT LIMIT, THERMAL LIMIT

BOX A

OUTLT1121-5

GND

1121_6 . eps

IN

SHUTDOWN

LOADBATTERYVCC

MAIN POWER

IR

N/C

0.33µF

Figure 3. LT1121 basic circuit

The LT1121 exhibits only a smallincrease in the ground-pin currentwhen the device nears its dropoutvoltage. This is a major improvementover other PNP low-dropout regula-tors. This is graphically illustrated inFigure 4.

Architecture and PackagesThe basic architecture of the

LT1121 is shown in Figure 5. Theoutput voltage is sampled by the cir-cuitry shown in box A. The output of

box A drives transistor Q1, whichdrives the base of the output PNPtransistor Q2. This architecture al-lows either the input or the output tobe held high or subjected to reversevoltage without excessive current flowor reverse current flow through theoutput transistor.

The LT1121 is available in three-lead TO-92 and SOT-223(surfacemount) packages, 8-lead SO (sur-face mount) and mini dip packages.In 8-pin packages, a shutdown pin is

provided, allowing logic control ofsystem power. Pulling the shutdownpin to ground turns off the outputstage of the regulator and reducesthe supply current to 16µA, includ-ing the shutdown-pin bias current.The output pin may still be held highwithout reverse current flowing. Asshown in Figure 6, the output of thedevice draws only 15µA when it isheld high and the input is at zerovolts.

Figure 4. Ground-pin current versusoutput current for the LT1121

Figure 5. LT1121 Basic architecture Figure 6. Power supply with battery backup. Reverseoutput current (IR) with VCC = 0V is only 15µA


DESIGN FEATURES

LT1227, continued from page 1

Figure 2. Many-input video MUX cable driver

1227_1.eps

+IN VOUT

V+

V–

NULL

–IN3

SHUTDOWN8

TURNS OFF ALL CURRENTS

14kQ13

Q1

Q3

Q2

Q4

V–

V+

NULL5

2

Q7

Q5

Q8

Q6

1

Q9

Q11

Q12

V–

V+

Q10

6

7

4

Figure 1. Video amplifier simplified schematic

Composite Video Flat BW – 3dB BWMonochrome Composite 1MHz @ 1dB 3MHzColor Composite 3MHz @ 1dB 5MHz(home use)Color Composite 5MHz @ 0.05dB 50MHz(professional)HDTV (professional) 30MHz @ 0.1dB 120MHz

RGB Video Pixel Width – 3dB BWVGA, 640 x 480 40ns 35MHzSVGA, 800 x 600 30ns 50MHzWorkstation, 1024 x 768 15ns 90MHzWorkstation, 1200 x 900 9ns 150MHz

Table 1. Bandwidths of popular video formats

In most video circuits, the imped-ance levels are quite low, in order tomaximize frequency response. Typicalfeedback resistors are less than 2kand the most common load is a 75Ωcable. For this reason, video amplifiersmust deliver high output currents.Video amplifiers should be specifiedwhen driving one or more 150Ω loads(the equivalent of driving one properlyterminated 75Ω cable) as well as driv-ing higher values such as 1kΩ.

What Amplifier ParametersAffect Video Performance?

When selecting a video amplifier, thefirst consideration is bandwidth. Table1 lists several video formats and theapproximate bandwidth each requires.If there is more than one amplifier in thesignal path, each will need more band-width than is indicated in the table.

The next parameter to look at isoutput-drive capability. If the amplifieris driving internal circuitry, the load isunlikely to be a problem. Cable-drivingapplications require more current thantraditional op amps can deliver; be surethe amplifier can deliver enough cur-rent. Also, check to see how much theamplifier slows down when driving aheavy load.

Heavy loads do more than just slowdown amplifiers. Because the outputimpedance of real amplifiers is neitherzero nor constant, the open-loop char-acteristics change with outputcurrent. For video amplifiers, the main

concern is bandwidth changing withthe output DC level. For RGB video thisis not too important, since the changesin bandwidth are usually small andimpossible to detect, but color compos-ite video is very sensitive to this. Smallchanges in bandwidth affect the gainand phase of the color subcarrier, caus-ing changes in tint and color intensity.The specifications that describe thesechanges in the color subcarrier are“differential gain” and “differentialphase.” Most television sets have about2 to 4% differential gain and 2 to 4degrees of differential phase. Profes-sional equipment has to be much bet-ter because the signal may pass throughseveral pieces of equipment and theerrors can accumulate.

The LT1227 Video AmplifierThe LT1227 is a current-feedback

amplifier optimized for speed and AClinearity. The excellent performance ofthe LT1227 is due to simple circuitryand a very fast complementaryprocess. This LTC proprietary processcombines 36V low-capacitance,600MHz PNPs and NPNs on one chip.The excellent DC matching of the tran-sistors allows the signal path to be verysimple and very fast. The simplifiedschematic in Figure 1 shows that theinput stage consists of just four tran-sistors: Q1–Q4. For maximum speed,only two transistors make up eachcurrent mirror (Q5 & Q6 and Q7 &Q8). The output buffer (Q9–Q12) is as

simple as the input stage. Anotheradvantage of simple circuits is thatthey operate on low supply voltages aswell as on standard higher-voltagesupplies.

The LT1227’s output is capable ofdriving over 60mA at 25°C. Currentlimiting reduces this with increasingtemperature in order to limit powerdissipation during fault conditions.The current-limit transistors are notshown in Figure 1.

1227_2.eps

+

–

1.5k

LT1227

83

75Ω

2

CABLE

+

–

1.5k

LT1227

83

2

+

–

1.5k

LT1227

8VIN1

3

2

75Ω

75Ω

75Ω6

1/6 74C9061 ON

VIN2

2 ON

VINn

n ON

1/6 74C906

1/6 74C906

6

6

DESIGN FEATURES


1063_6b.epsfCUTOFF

V OS

(OUT

)

0

100Hz

+1mV

–1mV

1kHz 10kHz10Hz

8

7

6

54

3

2

1VIN LTC1063

1063_6a.eps

fCLOCK0.1µF WIMA

0.1µF WIMA

– 5V1µF

TANTALUM

1µF TANTALUM

VOS (OUT)

+5V

The amplifier has offset-adjust pins,so a single potentiometer can vary thegain of one of the current mirrors. Thischanges the inverting-input bias cur-rent to null the output-offset voltage.Driving these pins with another ampli-fier will adjust the output offset auto-matically.

Pulling pin 8 of the LT1227 lowturns off the amplifier. The easiest wayto drive pin 8 is with an open-collectoror open-drain logic gate. The amplifieroperates normally when pin 8 is openor at VCC. Pulling pin 8 more than 4Vbelow VCC guarantees shutdown. Thereis no need for a current-limiting resis-tor in series with pin 8 because aninternal FET limits the current to100µA. The amplifier is shut down byturning off all the bias currents, andthe total supply current drops to 120µA,including the current pulled from pin8. The input stage is off and the outputimpedance rises to over 200k in paral-lel with 12pF. The feedback resistorsare still there, so for minimum loadingby the out-put when the amplifier isshutdown, operate the LT1227 at unity

gain. In typical cable driving applica-tions, the input-to-output isolation is70dB at 10MHz. The input will isolatesignals up to 10V peak-to-peak; largersignals will feed through the zenerclamp across the input stage. Thezener clamp protects the input tran-sistors from excessive voltage duringfaults such as electrostatic discharge.

Parameter Gain Load VS = ±15V VS = ±5V

Bandwidth AV = +1 RL = 150Ω 275MHz 200MHzBandwidth AV = +2 RL = 150Ω 140MHz 85MHzBandwidth AV = –1 RL = 150Ω 77MHz 70MHzBandwidth AV = +4 RL = 150Ω 90MHz 70MHzBandwidth AV = +10 RL = 150Ω 65MHz 50MHzDifferential Gain AV = +2 RL = 150Ω 0.014% 0.02%Differential Phase AV = +2 RL = 150Ω 0.01° 0.14°Output Slew Rate AV = +10 RL = 400Ω 1100V/µs 600V/µsInput Slew Rate AV = +1 150V/µs 100V/µsMinimum Output Current 30mA 30mAMaximum Input Offset Voltage 10mV 10mVMaximum Inverting Input Current 60µA 60µAInput Noise Voltage AV = +10 3nV/√Hz 3nV/√HzMaximum Quiescent Supply Current 15mA 13mA

Table 2. LT1227 performance characteristics

ConclusionsSampled-data filter technology has

evolved considerably over the pastseveral years. The output offset of theLTC1063 has been lowered by overtwo order of magnitudes in compari-son to the first switched-capacitorfilters. At the same time the AC per-formance of the device is at the lead-

ing edge of filter technology. TheLTC1063 has nothing to envy from itsactive R,C active counterparts. Fu-ture switched-capacitor filters willchallenge the active R,C filter in allperformance criteria, including out-put offset voltage, maximum cutofffrequency, and dynamic range.

1063, continued from page 11

filter exhibiting no more than 200µVof total output-offset variation overfour decades of cutoff frequency. Acombination of a WIMA ceramic ca-pacitor and a solid tantalum capaci-tor is used for power-supplydecoupling. The LTC1063 was testedusing a well thought out and care-fully designed PC board.

Figure 2 shows how simple it is to con-figure a video MUX using the LT1227.

LT1227 PerformanceTable 2 lists the major performance

specifications of the LT1227 on ±5Vand ±15V supplies. Performance on±12V is essentially the same ason ±15V.

Figure 6b. Output offset vs fCUTOFF forFigure 6a's circuit

Figure 6a. LTC1063 operating as a clock-sweepable lowpass filter


DESIGN FEATURES

Cascaded 8th-Order Butterworth FiltersProvide Steep Roll-Off Lowpass Filter

by Philip Karantzalis andRichard Markell

Sometimes a design requires a fil-ter that exceeds the specifications ofthe standard “dash-number” filter.In this case, the requirement was alow-distortion (–70dB) filter with roll-off faster than that of an 8th-orderButterworth. An elliptic filter wasruled out because its distortion speci-fications are too high.Two low-powerLTC1164-5s were wired in cascade to

investigate the specifications thatcould be achieved with this architec-ture. The LTC1164-5 is a low-power(4 milliamperes with ±5 volt sup-plies), clock-tunable, 8th-order filter,which can be configured for a Butter-worth or Bessel response by strap-ping a pin. Figure 1 shows theschematic diagram of the two-filtersystem. The frequency response is

shown in Figure 2, where it can beseen that the filter's attenuation is80dB at 2.3 times the cutoff fre-quency. The distortion, as shown inFigure 3, is nothing less than spec-tacular. From 100Hz to 1kHz, the twofilters have less than –74dB distor-tion specifications. At the standardmeasurement frequency of 1kHz, thespecification is –78dB.

DESIGN IDEAS

FREQUENCY (Hz)

100–100

THD

(%)

–90

–75

–65

–50

–40

1k

–45

–60

–70

–80

–95

–55

–85

1.4VRMS IN

±5V SUPPLY

1164_3. eps

FREQUENCY (Hz)

100–90

GAIN

(dB)

–70

–50

–30

–10

10

5k

1164_2. eps

0

–20

–40

–60

–80

1k

Figure 1. Schematic diagram: low-power, 16th-order lowpass filter (two 8th-order butterworths cascaded)

Figure 2. Frequency response for fCLK = 20kHz Figure 3. Distortion performance: two LTC1164-5s, fCLK = 60kHz(57:1) pin 10 connected to V+

1164_1. eps

LTC1164-5

1

2

3

4

5

6

7

14

13

12

11

10

9

8

LTC1164-5

1

2

3

4

5

6

7

14

13

12

11

10

9

8

INV (C)

VIN

GND

V+

GND

LP (A)

RIN (A)

OUT (C)

R3 SHUNT

V–

fCLK

50/100

VOUT

NC

INV (C)

VIN

GND

V+

GND

LP (A)

RIN (A)

OUT (C)

R3 SHUNT

V–

fCLK

50/100

VOUT

NC

0.1µF

0.1µF

0.1µF

0.1µF

16.2k

V–

16.2k

V–

fCLK1kΩ

+7.5V

–7.5V

V+ V+

–7.5V

+7.5V

DESIGN FEATURES


The circuit uses a micropowerLT1110 switching-regulator IC as acontroller. The internal switch of theLT1110 furnishes base drive to Q1through the 220Ω resistors. Q1, inturn, supplies base drive to the powerswitch Q2. The Zetex ZTX849 NPNdevice is rated at 5 amps current andcomes in a TO-92 package. For sur-face-mount fans, the FZT-849, alsofrom Zetex, provides the same perfor-mance in an SOT-223 package. The16Ω resistor provides a turn-off pathfor Q2’s stored charge. When Q2 is on,current builds in L1. As Q2 turns off,its collector flies positive until D1 turnson. L1’s built-up current dischargesthrough D1 into C2 and the load. Thevoltage at VOUT is divided by R4 and R3and fed back into the FB pin of theLT1110, which controls Q2’s cyclingaction. Switch-current limit, which isnecessary to ensure saturation oversupply variations, is implemented byQ3–Q5. Q3, C1, R2, and the auxiliarygain block inside the LT1110, form a220mV reference point at the LT1110’s

LT1110 Supplies 6 Volts at 550mAfrom 2 AA NiCad Cells

The LT1110 micropower DC–DCconverter can provide 5V at 150mAwhen operating from two AA alkalinecells. The internal switch VCE(sat) setsthis power limit. Even with an externallow-drop switch, more power is notrealistically possible. The internal im-pedance (typically 200mΩ fresh and500mΩ at end-of-life) of alkaline AAcells limits peak obtainable batterypower. Conversely, nickel-cadmiumcells have a constant internal imped-ance (35–50mΩ) for AA size) that in-creases only when the cell is completelydischarged. This allows power to bedrawn from the cell at a far greaterrate. The circuit in Figure 1 uses twoAA NiCad cells to supply 6 volts at550mA. The circuit, developed for pag-ers with transmit capability, runs atfull output current for 15 minuteswith two Gates Millennium AA NiCadcells. With a 250mA load, the circuitruns for 36 minutes. Less heat isgenerated with a reduced load, result-ing in the watt-hour difference observ-able above.

SET pin. Transistors Q4 and Q5 forma common-base differential amplifier.Q5’s emitter monitors the voltageacross 50mΩ resistor R1. When thevoltage across R1 exceeds 220mV, Q4turns on hard, pulling current throughR5. When the voltage at the ILIM pin ofthe LT1110 reaches a diode drop be-low the VIN pin, the internal switchturns off. Thus, maximum switch cur-rent is maintained at 220mV/50mΩ,or 4.4A, over input variations andmanufacturing spread in the LT1110’son time and frequency.

The circuit’s output ripple measures200mVp-p, and efficiency is 78% atfull load with a 2.4V input. Outputpower can be scaled down for lessdemanding requirements. To reducepeak current, increase the value of R1.A 100mΩ resistor will limit current to2.2A. L1 should be increased in valuelinearly as current is reduced. For acurrent limit of 2.2A, L1 should be10µH. Base drive for Q2 can also bereduced by increasing the value of the10Ω resistor. These lower peak cur-rents are much easier on alkaline cellsand will dramatically increase alka-line battery life.

by Steve Pietkiewicz

Figure 1. Schematic diagram, 2 AA NiCad to +6 volt converter

DESIGN IDEAS

Figure 2. Operating time atILOAD = 550mA and 250mA

TIME (MINUTES)

00

BATT

ERY/

OUTP

UT V

OLTA

GE (V

)

1.0

2.0

3.5

4.5

6.57.0

10 20 30 40

1110_2.eps

6.05.55.0

4.0

3.02.5

1.5

0.5

5 15 25 35

ILOAD = 250mAILOAD = 550mA

ILOAD = 250mAILOAD = 550mA

OUTPUT BATTERY

L1 5µH

10T # 18GA MAGNETICS INC. 55-041-A2 CORE

1110_1.eps

GND SW2FB

SW1ILIM VIN

D1 1N5820

R5 1k

C1 1µF

+

6VOUT 550mA

LT1110

AO

2 x AA NICAD

+

220

220

Q5 2N3904

Q2 ZETEX ZTX-849

16ΩR4

394k 1%

R2 1k

Q4 2N3904

Q3 2N3906

SET

10k 300pF

R3 15k 1%

R1 50mΩ

Q1 2N4403

5 4

8

3

21

6

7

10Ω

516) 543-7100 619) 661-6322 412) 282-8282

ZETEX ( SANYO (

MAGNETICS INC. (

C2 220µF 10V SANYO 0S-CON

100µF 10V SANYO OS-CON


DESIGN FEATURESDESIGN SOFTWARE

SwitcherCAD ReleaseSlated for August

Linear Technology will release anew design software program calledSwitcherCAD in August. SwitcherCADis a menu-driven DC-DC converterdesign-aid program that runs on IBM-PCs and compatibles. The programselects the appropriate DC-DCconverter architecture and LinearTechnology switching-regulator IC fora given design specification. In addi-tion, it selects the other essential com-ponents for the converter, includingthe input and output capacitors, in-ductor, catch diode, and output filter,and calculates the operating condi-tions for these components.SwitcherCAD can cut days off thedesign cycle by eliminating the neces-sity of searching through multiple datasheets, application notes, and maga-zine cookbook articles for the answerto a design problem.

SwitcherCAD was designed to beused by the novice and expert de-signer alike. SwitcherCAD is orga-nized around several different windowsor screens. First, you enter your de-sign requirements in the Design Speci-fication Screen, shown in Figure 1.The Design Specification screen as-sumes that the design requirements,including input-voltage range, outputvoltage and current, output ripple,and maximum ambient temperature,are well defined. Once this screen is

by Brian Huffman

Figure 2. SwitcherCAD Topology and Device Selection Screen

Allowed topologiesPositive BoostFlyback

PartsLT1171LT1170LT1071LT1070LT1270LT1270ALT1271

PackagesTO-220-5TO-3TO-220-11J-8N-8CS-8

Part PeakIsw Im Freq ThetaJA ThetaJCLT1170 1.797A 5.00A 100kHz 50.0 2.0

Select from allowed Topology & Device lists

Schematic

Expert Mode

Close

Novice Mode

Schematic

completed, advance to the Topologyand Device Selection Screen, shownin Figure 2.

In the Topology and Device Selec-tion Screen, SwitcherCAD determineswhich DC-DC converter topologies andwhich Linear Technology IC’s will sat-isfy the design requirements. Theremay be several architectures to choosefrom. Trade-offs between efficiencyand circuit complexity will influencethe final selection. SwitcherCAD sup-ports several topologies, includingstep-up, step-down, inverting, andflyback converters.

SwitcherCAD selects the optimumLinear Technology switching regula-tor ICs based upon topology, input-voltage range, output voltage, andcurrent requirements. The user canselect any part listed in the partscolumn to compare designs based ondifferent parts or topologies. The useralso has a choice of several packagestyles, although not all of the listedpackages are available for all LTCparts. SwitcherCAD displays the ap-proximate peak switch current of theselected part along with other electri-cal and thermal characteristics.

Novice ModeFigure 3 shows the Novice Mode

screen. Novice Mode provides the userwith a schematic diagram and a list ofcomponents and operating conditions.A viable DC-DC converter can be builtand tested directly from the circuitdiagram to get a circuit “on the air.”This screen can either be printedin its entirety or scrolled throughon screen. SwitcherCAD uses conser-vative design practices to select allpower components. Components areselected from an extensive database

Switcher Design Spec.

Figure 1. SwitcherCAD Design Specification Screen

Output Voltage (V): Nom 6Input Voltage (V): Min 5 Nom 5 Max 5Load Current (A): Min 0.05 Nom 0.08 Max 1

Allowed Output Ripple Voltage mV(p-p): 120

Output #2 and #3 are for Multiple Output Switcher Only: Vout(V) IoutMin(A) IoutMax(A) Ripple (mV)Output #2: 0 0 0 0Output #3: 0 0 0 0

Max Temperature (°C): Ambient 60 IC 100 Diode 100

Fully Isolated Output: (•) No ( ) Yes

Next Close

DESIGN FEATURES


Figure 4. SwitcherCAD Expert Mode Screen

DESIGN SOFTWARE

I Output VoltageI Min Input VoltageI Nom Input VoltageI Max Input VoltageI Min Load CurrentI Nom Load CurrentI Max Load CurrentI Output Ripple VoltageI Max Ambient Temperature

IC CharacteristicsI Maximum Rated Switch VoltageI Rated Switch CurrentI Switch On ResistanceI Switch Offset Voltage Loss

I Min Input Voltage

Variable Watch List: O Power Dissipated in IC O Output Capacitor RMS Ripple Current O Input Capacitor RMS Ripple Current

6.05.05.05.00.0500.0801.000120.060

65.02.5000.4000.0

VVVVAAAmV p-p°C

VAΩV

WA RMSA RMS

0.320.6700.342

Positive Boost [LT1170]

Print

Help

Close

Watch

=========

====

<<Input

===

5.0

containing diodes, inductors and ca-pacitors. Electrical characteristics ofthe components are extracted fromthe database when operating charac-teristics are computed. The programprovides enough information aboutthe selected components to allow theuser to make substitutions.

SwitcherCAD calculates all operat-ing conditions and selects componentsbased on a minimum input voltage andmaximum output current. To vary theinput conditions to simulate worst-case conditions for each component,the user should advance to Expert Mode.

Expert ModeThe Expert Mode screen, shown in

Figure 4, displays a “spreadsheet” con-taining a detailed view of a converterdesign and its operating conditions.More important, this screen allows theuser to modify selected componentvalues and operating conditions andview the results on the screen or printthem in a report. This is useful forperforming “worst case” analysis andfor playing “what if” with componentvalues and operating conditions.

An on-line help feature providesquick information about operating con-ditions and component selections. Helpprovides detailed information on thesignificance of the highlighted param-eter or value, how it is calculated, and

its role in determining other valuesand parameters. Understanding howand why a particular component isselected is essential in designing areliable switching regulator circuit.

Because only a small portion of thespreadsheet can be displayed on thescreen at one time, SwitcherCAD isequipped with a “Watch List” at thebottom of the screen. This window

Operating Conditions: C1 Nichicon x1 UPL1A151MPH 150µF 10V ESR=0.350Ω Irms= 0.265A C2 United Chemi-Con x1 LXF16VB181M6X15LL 180µF 16V ESR=0.220Ω C3 Nichicon x1 UPL1A151MPH 150µF 10V ESR=0.350Ω Irms=0.265A L1 Gowanda 050AT1002 10µH DCR=0.010Ω L2 Hurricane HL-8015 3µH Ipk=3A D1 Motorola Schottky MBR115P If=1A Vr=15V Trr=0nS R1 4.74kΩ 1% R2 1.24kΩ 1% Rc Suggested Value: 1.0kΩ Cc Suggested Value: 1.0µF U1 LT1170 Im=5.0A Vs=0.0V Rsw=0.20Ω f=100kHz 1 Vout=6V VinMin=5V VinNom=5V VinMax=5V 2 IoMin=0.5A IoNom=0.08A IoMax=1A 3 Ripple=120mV TaMax=60°C TjMax=100°C TjDMax=100°C

O Operating Mode at Full Load Current = _Cont_ O Duty Cycle = 25.5 % O Maximum Rated Switch Current at this DC = 5.000 A

Print

Close

Positive Boost Operating Condition & Parts List

Figure 3. SwitcherCAD Novice Mode Screen

allows the user to view up to threeselected variables without having toscroll through the spreadsheet. Thisfeature is particularly useful for find-ing worst-case operating conditionfor the various components. For ex-ample, input-capacitor RMS ripplecurrent, output-capacitor RMS ripplecurrent, and IC power dissipation canbe monitored simultaneously as theinput voltage is changed.

As with any CAD program, Switch-erCAD performs computer aided de-sign, not computer generated design.At the core of SwitcherCAD are up to300 design equations for each topol-ogy. The equations quantify the oper-ating point for each circuit component,but they cannot guarantee that, onceassembled, the circuit will meet theuser’s every expectation. It is the re-sponsibility of the user to verifySwitcherCAD’s work by building thecircuit and evaluating its performanceby measuring component stresses un-der all expected operating conditions.

SwitcherCAD will be available in Au-gust from your local LTC sales office, orby calling the 800 number listed on thelast page of this magazine.


DESIGN FEATURES

LT1225, LT1226: New Ultra-High-Speed Op Amps for FastData-Acquisition Systems

The new LT1225 and LT1226 areultra-high-speed operational amplifi-ers with excellent DC performance.The output currents of the LT1225and LT1226 are typically 24mA mini-mum, making them capable of drivinga 500Ω load to ±12V with ±15V sup-plies and a 150Ω load to ±3V with ±5Vsupplies. The unique output stages ofthese devices allow them to remainstable while driving any amount ofcapacitive load, which makes themuseful in buffer or cable-driving appli-cations. The LT1225 is compensatedto be stable for gains of five or moreand provides a 150MHz gain-band-width product, 400V/µs slew rate, 90nanosecond settling time to 0.1% witha 10V step, and 20V/mV DC gain. TheLT1225 consumes only 7mA of supplycurrent. A key feature of the LT1225 isits precision offset voltage. The LT1225provides a 1mV maximum offset volt-age, typically 0.5mV. The LT1226 is alow-noise, ultra-high-speed opera-tional amplifier with excellent DC per-formance. The device is stable for gainsof 25 or more and provides a 1GHzgain-bandwidth product, 400V/µsslew rate, 50V/mV minimum large-signal voltage gain, and 100 nanosec-onds settling time to 0.1% with a 10Vstep. Key features of the LT1226 in-clude a maximum offset voltage of1mV and a mere 2.6 nanovolts/(Hz)1/

2 of noise. Applications include high-speed data-acquisition systems, cabledriving, RF amplifiers, and high-speedsample-and-hold circuits. Additionalapplications include input buffers forA/D converters, output buffers forD/A converters, and video-imagingcircuitry.

8-Pin 12-Bit CompleteData-Acquisition System

The LTC1291 is the latest 12-Bit A/D converter to join Linear Technology’sgrowing list of data-acquisition prod-

by LTC MarketingNew Device CameosNEW DEVICE CAMEOS

ucts. This device is a completedata-acquisition system packaged inan 8-pin DIP. It includes a serial-I/O,successive-approximation A/D thatuses switched-capacitor technology toperform 12-bit unipolar conversions.The analog front end has a two-chan-nel multiplexer and a sample-and-hold. The multiplexer can be configuredfor single-ended or differential inputs.With the sample-and-hold, fast-mov-ing signals with bandwidths of up to27kHz can be digitized. An externalreference is not required because theLTC1291 takes its reference from thepower supply pin (VCC). When the de-vice is not being used, it can be put intoa power-shutdown mode that reducesthe supply current to 5µA. The serial I/O allows the LTC1291 to be easilyinterfaced to most processors with asfew as three wires. Ease of use, smallpackage, and low power consumptionmake this device will suited for re-mote-sensing applications and por-table, hand-held instruments.

New LT1137A Advanced, Low-Power 5V RS232 Transceiverwith Small Capacitors

The LT1137A is a three-driver, five-receiver RS232 transceiver that is pincompatible with the industry-standardLT1137, and offers performance im-provements that permit operation withsmall (0.1 microfarad) storage capaci-tors. The LT1137A’s charge pump isdesigned for extended compliance andcan deliver over 40 milliamps of loadcurrent. Additional load-drive capa-bility is important for driving externalperipherals such as mice. Supply cur-rent is typically 12 milliamps, which iscompetitive with similar CMOS de-vices. An advanced driver output stageoperates to 120kbaud while drivingheavy capacitive loads.

The LT1137A is fully compliant withall EIA-RS232 specifications. Specialbipolar construction techniques pro-tect the drivers and receivers beyondthe fault conditions stipulated for

RS232. Driver outputs and receiverinputs can be shorted to ±30V withoutdamaging the device or the power-supply generator. In addition, theRS232 I/O lines are resilient to mul-tiple 5kV ESD strikes.

The transceiver has two shutdownmodes. One mode disables the driversand charge pump, while the othershuts down all circuitry. While shutdown, the drivers and receivers as-sume high impedance output states.

The new LT1137A is specified fortemperature ranges from military (MJ)to commercial (CN and CJ) and is alsoavailable in a 28-pin surface-mount-able package.

New LT1112 Dual Microvolt-Offset, Picoamp-Input-CurrentOp Amp In SO8 Package

The new LT1112 is a dual, inter-nally compensated, universal preci-sion operational amplifier. This parthas the lowest offset voltage of anydual non-chopped op amp. The lowestgrade has a guaranteed offset of 75microvolts. The LT1112 combinespicoampere bias currents, microvoltoffset voltage, low voltage and currentnoise and low power dissipation. TheLT1112 achieves precision operationon two NiCad batteries with 1.6mW ofpower dissipation. The part alsoachieves very high CMRR and PSRRspecifications, almost non-existent warm-up drift, and the abilityto deliver 5 milliamps load current.

The LT1112 is available in theindustry-standard dual SO8 opera-tional-amplifier pinout. The dual am-plifiers have a full set of matchingspecifications. These matching speci-fications make it easy to use the am-plifiers in applications that requireamplifier matching, such as two- andthree-amplifier instrumentation am-plifiers. The quad version of the LT1112is known as the LT1114. The LT1114features four operational amplifiers ina single package, with the same guar-anteed specifications as the LT1112.

DESIGN FEATURES


NEW DEVICE CAMEOS

Another set of specifications arefurnished at ±1V supply. Thesespecifications, plus the low 290-µA-per-amplifier supply current, allow theLT1112/LT1114 to be powered by twonearly discharged AA cells.

The LT1109A MicropowerStep-up DC–DC Converter

The LT1109A is the newest memberof LTC’s growing micropower DC–DCconverter family. Housed in a space-saving 8-pin DIP or SO package, theLT1109A-12 delivers 12 volts at over250mA from a 5 volt supply. Up to fourflash-memory chips can be pro-grammed simultaneously using theLT1109A for Vpp generation. Like itslittle brother the LT1109, the “A” ver-sion has a logic-level shutdown inputand high-speed 120kHz operation. A1.5A switch, on-chip switch-currentlimit, and a 65% duty cycle provideover twice the output power of theLT1109. The LT1109A operates over a2-to-9-volt input range and consumesjust 300µA quiescent current underno-load or shutdown conditions. Five-volt and adjustable versions are alsoavailable.

Surface Mount PowerDevices in DD PackagesLTC will soon have available severalpower devices in a new surface mountpower package, generically referred toas the DD package. This package hasbeen specially designed to allow voltageregulators to be surface mounted. Eversince LTC began offering the SO pack-age for surface mount applications,customers have been asking for linearand switching regulators in surfacemount packages. Believe it or not,otherwise sane and intelligent engi-neers will call up and ask for “anLT1084 in an SO package.” What wasrequired was a surface-mountablepackage designed especially to dissi-pate power. Now it’s here.

3 2 1

TOP VIEW

M PACKAGE 3 LEAD PLASTIC DD

DD3 • POI01

Figure 1. DD packages

3 2 1

TOP VIEW

Q PACKAGE 5 LEAD PLASTIC DD

DD5 • POI01

4 5

In the past, when designs employ-ing surface mount technology (SMT)required high-current voltage re-gulators, one was forced to either usestandard packages with the leadsformed for surface mounting or tohave holes drilled on an otherwiseSMT board. Neither choice is idealfrom cost and manufacturability view-points. The DD package is the solu-tion. The DD package is available in 3,5, and 7 lead versions and will allowLTC to offer everything from simple3-terminal regulators to full-featuredswitching regulators. The package suf-fix designators are as follows:3-Lead: M, 5-Lead: Q, 7-Lead: R.

An example of a typical part num-ber is “LT1171CQ”

The three package types are shownin Figure 1.

Mechanical and Thermal Details TheDD package resembles a TO-220 withthe tab cut off and the leads clippedand formed for surface mounting. Mea-surements verify about 2°C/W junc-tion-to-case thermal resistance. Witha surface-mounted device, most heatsinking is done through the board, solayout and copper pad size will con-tribute to the overall thermal resis-tance number. It is unlikely that alinear regulator operating at 3 amps

at high input/output voltage differen-tial will be a candidate for surfacemounting, but the DD package willallow many high-current switchingregulation tasks to be performed.

Part Types to be Offered The fol-lowing is a list of our initial productoffering in the DD package:

LT1171CQ 3A 100kHz switchingregulator

LT1271CQ 4A high-efficiencyswitching regulator

LT1076CR-5 5V, 2A step-downswitching regulator

For further information onany devices mentioned in this issue ofLinear Technology, use the readerservice card or call the LTC literatureservice number: (800) 637-5545. Askfor the pertinent data sheets andapplication notes.

Information furnished by Linear Technology Cor-poration is believed to be accurate and reliable.However, no responsibility is assumed for its use.Linear Technology makes no representation thatthe circuits described herein will not infringe onexisting patent rights.

421

TOP VIEW

R PACKAGE 7 LEAD PLASTIC DD

3 65 7


DESIGN FEATURES

DESIGN TOOLSApplications on DiskNOISE DISKThis IBM-PC (or compatible) progam allows the user tocalculate circuit noise using LTC op amps, determine thebest LTC op amp for a low noise application, display thenoise data for LTC op amps, calculate resistor noise, andcalculate noise using specs for any op amp.Available at no charge.

SPICE MACROMODEL DISKThis IBM-PC (or compatible) high density diskette containsthe library of LTC op amp SPICE macromodels. Themodels can be used with any version of SPICE for generalanalog circuit simulations. The diskette also contains work-ing circuit examples using the models, and a demonstrationcopy of PSPICETM by MicroSim.Available at no charge.

Technical BooksLinear Databook — This 1,400 page collection of datasheets covers op amps, voltage regulators, references,comparators, filters, PWMs, data conversion and interfaceproducts (bipolar and CMOS), in both commercial andmilitary grades. The catalog features well over 300 devices.$10.00

Linear Applications Handbook — 928 pages chock full ofapplication ideas covered in-depth through 40 ApplicationNotes and 33 Design Notes. This catalog covers a broadrange of “real world” linear circuitry. In addition to detailed,systems-oriented circuits, this handbook contains broadtutorial content together with liberal use of schematics andscope photography. A special feature in this edition in-cludes a 22-page section on SPICE macromodels.$20.00

Monolithic Filter Handbook — This 232 page book comeswith a disk which runs on PCs. Together, the book and diskassist in the selection, design and implementation of theright switched capacitor filter circuit. The disk containsstandard filter responses as well as a custom mode. Thehandbook contains over 20 data sheets, Design Notes andApplication Notes.$40.00

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(408) 432-1900Literature Department (800) 637-5545

© 1992 Linear Technology Corporation/ Printed in U.S.A./20K

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