REV. CInformation furnished by Analog Devices is believed to be accurate andreliable. However, no responsibility is assumed by Analog Devices for itsuse, nor for any infringements of patents or other rights of third partieswhich may result from its use. No license is granted by implication orotherwise under any patent or patent rights of Analog Devices.

aAD9764

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel:781/329-4700 World Wide Web Site: http://www.analog.com Fax:781/326-8703 © Analog Devices, Inc., 1999-2016

14-Bit, 125 MSPSTxDAC® D/A Converter

FUNCTIONAL BLOCK DIAGRAMFEATURES

Member of Pin-Compatible TxDAC Product Family

125 MSPS Update Rate

14-Bit Resolution

Excellent SFDR and IMD

Differential Current Outputs: 2 mA to 20 mA

Power Dissipation: 190 mW @ 5 V to 45 mW @ 3 V

Power-Down Mode: 25 mW @ 5 V

On-Chip 1.20 V Reference

Single +5 V or +3 V Supply Operation

Packages: 28-Lead SOIC and TSSOP

Edge-Triggered Latches

APPLICATIONS

Communication Transmit Channel:

Basestations

ADSL/HFC Modems

Instrumentation

PRODUCT DESCRIPTIONThe AD9764 is the 14-bit resolution member of the TxDACseries of high performance, low power CMOS digital-to-analogconverters (DACs). The TxDAC family, which consists of pincompatible 8-, 10-, 12-, and 14-bit DACs, is specifically opti-mized for the transmit signal path of communication systems.All of the devices share the same interface options, small outlinepackage and pinout, providing an upward or downward compo-nent selection path based on performance, resolution and cost.The AD9764 offers exceptional ac and dc performance whilesupporting update rates up to 125 MSPS.

The AD9764’s flexible single-supply operating range of 2.7 V to5.5 V and low power dissipation are well suited for portable andlow power applications. Its power dissipation can be furtherreduced to a mere 45 mW with a slight degradation in performanceby lowering the full-scale current output. Also, a power-downmode reduces the standby power dissipation to approximately25 mW.

The AD9764 is manufactured on an advanced CMOS process.A segmented current source architecture is combined with aproprietary switching technique to reduce spurious componentsand enhance dynamic performance. Edge-triggered inputlatches and a 1.2 V temperature compensated bandgap refer-ence have been integrated to provide a complete monolithicDAC solution. Flexible supply options support +3 V and +5 VCMOS logic families.

The AD9764 is a current-output DAC with a nominal full-scaleoutput current of 20 mA and > 100 kΩ output impedance.

TxDAC is a registered trademark of Analog Devices, Inc.

Differential current outputs are provided to support single-ended or differential applications. Matching between the twocurrent outputs ensures enhanced dynamic performance in adifferential output configuration. The current outputs may betied directly to an output resistor to provide two complemen-tary, single-ended voltage outputs or fed directly into a trans-former. The output voltage compliance range is 1.25 V.

The on-chip reference and control amplifier are configured formaximum accuracy and flexibility. The AD9764 can be drivenby the on-chip reference or by a variety of external referencevoltages. The internal control amplifier, which provides a wide(>10:1) adjustment span, allows the AD9764 full-scale currentto be adjusted over a 2 mA to 20 mA range while maintainingexcellent dynamic performance. Thus, the AD9764 may operateat reduced power levels or be adjusted over a 20 dB range toprovide additional gain ranging capabilities.

The AD9764 is available in 28-lead SOIC and TSSOP packages.It is specified for operation over the industrial temperature range.

PRODUCT HIGHLIGHTS1. The AD9764 is a member of the TxDAC product family that

provides an upward or downward component selection pathbased on resolution (8 to 14 bits), performance and cost.

2. Manufactured on a CMOS process, the AD9764 uses a pro-prietary switching technique that enhances dynamic perfor-mance beyond that previously attainable by higher power/costbipolar or BiCMOS devices.

3. On-chip, edge-triggered input CMOS latches readily interfaceto +3 V and +5 V CMOS logic families. The AD9764 cansupport update rates up to 125 MSPS.

4. A flexible single-supply operating range of 2.7 V to 5.5 V, anda wide full-scale current adjustment span of 2 mA to 20 mA,allows the AD9764 to operate at reduced power levels.

5. The current output(s) of the AD9764 can be easily config-ured for various single-ended or differential circuit topologies.

50pF

COMP1

+1.20V REF

AVDD ACOMREFLO

COMP2CURRENTSOURCEARRAY

0.1mF

+5V

SEGMENTEDSWITCHES

LSBSWITCHES

REFIO

FS ADJ

DVDD

DCOM

CLOCK

+5VRSET

0.1mF

CLOCK

IOUTA

IOUTB

0.1mF

LATCHES

AD9764

SLEEP

DIGITAL DATA INPUTS (DB13–DB0)

REV. C–2–

AD9764–SPECIFICATIONSDC SPECIFICATIONSParameter Min Typ Max Units

RESOLUTION 14 Bits

DC ACCURACY1

Integral Linearity Error (INL)TA = +25°C –4.5 ±2.5 +4.5 LSBTMIN to TMAX –6.5 +6.5 LSB

Differential Nonlinearity (DNL)TA = +25°C –2.5 ±1.5 +2.5 LSBTMIN to TMAX –4.5 +4.5 LSB

ANALOG OUTPUTOffset Error –0.025 +0.025 % of FSRGain Error (Without Internal Reference) –2 ±1 +2 % of FSRGain Error (With Internal Reference) –7 ±1 +7 % of FSRFull-Scale Output Current2 2.0 20.0 mAOutput Compliance Range –1.0 1.25 VOutput Resistance 100 kΩOutput Capacitance 5 pF

REFERENCE OUTPUTReference Voltage 1.08 1.20 1.32 VReference Output Current3 100 nA

REFERENCE INPUTInput Compliance Range 0.1 1.25 VReference Input Resistance 1 MΩSmall Signal Bandwidth (w/o CCOMP1)4 1.4 MHz

TEMPERATURE COEFFICIENTSOffset Drift 0 ppm of FSR/°CGain Drift (Without Internal Reference) ±50 ppm of FSR/°CGain Drift (With Internal Reference) ±100 ppm of FSR/°CReference Voltage Drift ±50 ppm/°C

POWER SUPPLYSupply Voltages

AVDD5 2.7 5.0 5.5 VDVDD 2.7 5.0 5.5 V

Analog Supply Current (IAVDD) 25 30 mADigital Supply Current (IDVDD)6 1.5 4 mASupply Current Sleep Mode (IAVDD) 5.0 8.5 mAPower Dissipation6 (5 V, IOUTFS = 20 mA) 133 170 mWPower Dissipation7 (5 V, IOUTFS = 20 mA) 190 mWPower Dissipation7 (3 V, IOUTFS = 2 mA) 45 mWPower Supply Rejection Ratio8—AVDD –0.4 +0.4 % of FSR/VPower Supply Rejection Ratio8—DVDD –0.025 +0.025 % of FSR/V

OPERATING RANGE –40 +85 °CNOTES1Measured at IOUTA, driving a virtual ground.2Nominal full-scale current, IOUTFS, is 32 × the IREF current.3Use an external buffer amplifier to drive any external load.4Reference bandwidth is a function of external cap at COMP1 pin and signal level.5For operation below 3 V, it is recommended that the output current be reduced to 12 mA or less to maintain optimum performance.6Measured at fCLOCK = 25 MSPS and fOUT = 1.0 MHz.7Measured as unbuffered voltage output with IOUTFS = 20 mA and 50 Ω RLOAD at IOUTA and IOUTB, fCLOCK = 100 MSPS and fOUT = 40 MHz.8± 5% Power supply variation.

Specifications subject to change without notice.

(TMIN to TMAX , AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, unless otherwise noted)

REV. C –3–

AD9764

DYNAMIC SPECIFICATIONSParameter Min Typ Max Units

DYNAMIC PERFORMANCEMaximum Output Update Rate (fCLOCK) 125 MSPSOutput Settling Time (tST) (to 0.1%)1 35 nsOutput Propagation Delay (tPD) 1 nsGlitch Impulse 5 pV-sOutput Rise Time (10% to 90%)1 2.5 nsOutput Fall Time (10% to 90%)1 2.5 nsOutput Noise (IOUTFS = 20 mA) 50 pA/√HzOutput Noise (IOUTFS = 2 mA) 30 pA/√Hz

AC LINEARITYSpurious-Free Dynamic Range to Nyquist

fCLOCK = 25 MSPS; fOUT = 1.00 MHz0 dBFS Output

TA = +25°C 75 82 dBcTMIN to TMAX 73 dBc

–6 dBFS Output 85 dBc–12 dBFS Output 77 dBc–18 dBFS Output 70 dBc

fCLOCK = 50 MSPS; fOUT = 1.00 MHz 80 dBcfCLOCK = 50 MSPS; fOUT = 2.51 MHz 77 dBcfCLOCK = 50 MSPS; fOUT = 5.02 MHz 70 dBcfCLOCK = 50 MSPS; fOUT = 20.2 MHz 58 dBc

Spurious-Free Dynamic Range within a WindowfCLOCK = 25 MSPS; fOUT = 1.00 MHz; 2 MHz Span

TA = +25°C 78 89 dBcTMIN to TMAX 76 dBc

fCLOCK = 50 MSPS; fOUT = 5.02 MHz; 2 MHz Span 84 dBcfCLOCK = 100 MSPS; fOUT = 5.04 MHz; 4 MHz Span 84 dBc

Total Harmonic DistortionfCLOCK = 25 MSPS; fOUT = 1.00 MHz

TA = +25°C –78 –74 dBcTMIN to TMAX –72 dBc

fCLOCK = 50 MHz; fOUT = 2.00 MHz –75 dBcfCLOCK = 100 MHz; fOUT = 2.00 MHz –75 dBc

Multitone Power Ratio (Eight Tones at 110 kHz Spacing)fCLOCK = 20 MSPS; fOUT = 2.00 MHz to 2.99 MHz

0 dBFS Output 73 dBc–6 dBFS Output 76 dBc–12 dBFS Output 73 dBc–18 dBFS Output 64 dBc

NOTES1Measured single-ended into 50 Ω load.

Specifications subject to change without notice.

(TMIN to TMAX , AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, Differential Transformer Coupled Output,50 V Doubly Terminated, unless otherwise noted)

REV. C

AD9764

–4–

DIGITAL SPECIFICATIONSParameter Min Typ Max Units

DIGITAL INPUTSLogic “1” Voltage @ DVDD = +5 V 3.5 5 VLogic “1” Voltage @ DVDD = +3 V 2.1 3 VLogic “0” Voltage @ DVDD = +5 V 0 1.3 VLogic “0” Voltage @ DVDD = +3 V 0 0.9 VLogic “1” Current –10 +10 µALogic “0” Current –10 +10 µAInput Capacitance 5 pFInput Setup Time (tS) 2.0 nsInput Hold Time (tH) 1.5 nsLatch Pulsewidth (tLPW) 3.5 ns

Specifications subject to change without notice.

0.1%

0.1%

tS tH

tLPW

tPD tST

DB0–DB13

CLOCK

IOUTAOR

IOUTB

Figure 1. Timing Diagram

ORDERING GUIDE

THERMAL CHARACTERISTICS Thermal Resistance28-Lead 300 mil SOIC

θJA = 71.4° C/WθJC = 23° C/W

28-Lead TSSOPθJA = 97.9° C/W θJC = 14.0° C/W

ABSOLUTE MAXIMUM RATINGS*

WithParameter Respect to Min Max Units

AVDD ACOM –0.3 +6.5 VDVDD DCOM –0.3 +6.5 VACOM DCOM –0.3 +0.3 VAVDD DVDD –6.5 +6.5 VCLOCK, SLEEP DCOM –0.3 DVDD + 0.3 VDigital Inputs DCOM –0.3 DVDD + 0.3 VIOUTA, IOUTB ACOM –1.0 AVDD + 0.3 VCOMP1, COMP2 ACOM –0.3 AVDD + 0.3 VREFIO, FSADJ ACOM –0.3 AVDD + 0.3 VREFLO ACOM –0.3 +0.3 VJunction Temperature +150 °CStorage Temperature –65 +150 °CLead Temperature

(10 sec) +300 °C

*Stresses above those listed under Absolute Maximum Ratings may cause perma-nent damage to the device. This is a stress rating only; functional operation of thedevice at these or any other conditions above those indicated in the operationalsections of this specification is not implied. Exposure to absolute maximum ratingsfor extended periods may affect device reliability.

CAUTIONESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readilyaccumulate on the human body and test equipment and can discharge without detection.Although the AD9764 features proprietary ESD protection circuitry, permanent damage mayoccur on devices subjected to high energy electrostatic discharges. Therefore, proper ESDprecautions are recommended to avoid performance degradation or loss of functionality.

WARNING!

ESD SENSITIVE DEVICE

1 RW = Small Outline IC, RU = TSSOP. 2 Z = RoHS Compliant Part.

REV. C

AD9764

–5–

PIN CONFIGURATION

14

13

12

11

17

16

15

20

19

18

10

9

8

1

2

3

4

7

6

5

TOP VIEW(Not to Scale)

28

27

26

25

24

23

22

21

AD9764

NC = NO CONNECT

(MSB) DB13

DB12

DB11

DB10

DB9

DB8

DB7

DB6

DB5

DB4

DB3

DB2

DB1

DB0

CLOCK

DVDD

DCOM

NC

AVDD

COMP2

IOUTA

IOUTB

ACOM

COMP1

FS ADJ

REFIO

REFLO

SLEEP

PIN FUNCTION DESCRIPTIONS

Pin No. Name Description

1 DB13 Most Significant Data Bit (MSB).2–13 DB12–DB1 Data Bits 1–12.14 DB0 Least Significant Data Bit (LSB).15 SLEEP Power-Down Control Input. Active High. Contains active pull-down circuit; it may be left unterminated if

not used.16 REFLO Reference Ground when Internal 1.2 V Reference Used. Connect to AVDD to disable internal reference.17 REFIO Reference Input/Output. Serves as reference input when internal reference disabled (i.e., Tie REFLO to

AVDD). Serves as 1.2 V reference output when internal reference activated (i.e., Tie REFLO to ACOM).Requires 0.1 µF capacitor to ACOM when internal reference activated.

18 FS ADJ Full-Scale Current Output Adjust.19 COMP1 Bandwidth/Noise Reduction Node. Add 0.1 µF to AVDD for optimum performance.20 ACOM Analog Common.21 IOUTB Complementary DAC Current Output. Full-scale current when all data bits are 0s.22 IOUTA DAC Current Output. Full-scale current when all data bits are 1s.23 COMP2 Internal Bias Node for Switch Driver Circuitry. Decouple to ACOM with 0.1 µF capacitor.24 AVDD Analog Supply Voltage (+2.7 V to +5.5 V).25 NC No Internal Connection.26 DCOM Digital Common.27 DVDD Digital Supply Voltage (+2.7 V to +5.5 V).28 CLOCK Clock Input. Data latched on positive edge of clock.

REV. C

AD9764

–6–

DEFINITIONS OF SPECIFICATIONSLinearity Error (Also Called Integral Nonlinearity or INL)Linearity error is defined as the maximum deviation of theactual analog output from the ideal output, determined by astraight line drawn from zero to full scale.

Differential Nonlinearity (or DNL)DNL is the measure of the variation in analog value, normalizedto full scale, associated with a 1 LSB change in digital inputcode.

Offset ErrorThe deviation of the output current from the ideal of zero iscalled offset error. For IOUTA, 0 mA output is expected when theinputs are all 0s. For IOUTB, 0 mA output is expected when allinputs are set to 1s.

Gain ErrorThe difference between the actual and ideal output span. Theactual span is determined by the output when all inputs are setto 1s minus the output when all inputs are set to 0s.

Output Compliance RangeThe range of allowable voltage at the output of a current-outputDAC. Operation beyond the maximum compliance limits maycause either output stage saturation or breakdown, resulting innonlinear performance.

Temperature DriftTemperature drift is specified as the maximum change from theambient (+25°C) value to the value at either TMIN or TMAX. Foroffset and gain drift, the drift is reported in ppm of full-scalerange (FSR) per degree C. For reference drift, the drift isreported in ppm per degree C.

Power Supply RejectionThe maximum change in the full-scale output as the suppliesare varied over a specified range.

Settling TimeThe time required for the output to reach and remain within aspecified error band about its final value, measured from thestart of the output transition.

Glitch ImpulseAsymmetrical switching times in a DAC give rise to undesiredoutput transients that are quantified by a glitch impulse. It isspecified as the net area of the glitch in pV-s.

Spurious-Free Dynamic RangeThe difference, in dB, between the rms amplitude of the outputsignal and the peak spurious signal over the specified bandwidth.

Total Harmonic DistortionTHD is the ratio of the sum of the rms value of the first sixharmonic components to the rms value of the measured outputsignal. It is expressed as a percentage or in decibels (dB).

Multitone Power RatioThe spurious-free dynamic range for an output containing mul-tiple carrier tones of equal amplitude. It is measured as thedifference between the rms amplitude of a carrier tone to thepeak spurious signal in the region of a removed tone.

50pF

COMP1

+1.20V REF

AVDD ACOMREFLO

COMP2PMOS

CURRENT SOURCEARRAY

0.1mF

SEGMENTED SWITCHESFOR DB13–DB5

LSBSWITCHES

REFIO

FS ADJ

DVDD

DCOM

CLOCK

+5VRSET2kV

0.1mF

DVDDDCOM

IOUTA

IOUTB

0.1mF

AD9764

SLEEP50V

RETIMEDCLOCK

OUTPUT*

LATCHES

DIGITALDATA

TEKTRONIXAWG-2021

LECROY 9210PULSE GENERATOR

CLOCKOUTPUT

50V20pF

50V20pF

100V

TO HP3589ASPECTRUM/NETWORKANALYZER50V INPUT

MINI-CIRCUITST1-1T

* AWG2021 CLOCK RETIMEDSUCH THAT DIGITAL DATATRANSITIONS ON FALLING EDGEOF 50% DUTY CYCLE CLOCK.

+5V

Figure 2. Basic AC Characterization Test Setup

REV. C

AD9764

–7–

Typical AC Characterization Curves(AVDD = +5 V, DVDD = +3 V, IOUTFS = 20 mA, 50 V Doubly Terminated Load, Differential Output, TA = +258C, SFDR up to Nyquist, unless otherwise noted)

FREQUENCY – MHz

SF

DR

– d

Bc

90

85

400.1 1 10010

55

45

50

65

60

80

70

75

5 MSPS

25 MSPS

50 MSPS

100 MSPS

Figure 3. SFDR vs. fOUT @ 0 dBFS

FREQUENCY – MHz

SF

DR

– d

Bc

90

85

400 5 2510 15 20

60

55

50

45

80

70

75

65

0dBFS

–6dBFS

–12dBFS

Figure 6. SFDR vs. fOUT @ 50 MSPS

AOUT – dBFS

SF

DR

– d

Bc

90

80

50–30 –25 0–20 –15 –10 –5

70

60

455kHz @ 5 MSPS

2.27MHz @ 25 MSPS

4.55MHz @ 50 MSPS

9.09MHz @ 100 MSPS

Figure 9. Single-Tone SFDR vs. AOUT

@ fOUT = fCLOCK/11

FREQUENCY – MHz

SF

DR

– d

Bc

90

85

400 0.5 2.51.0 1.5 2.0

60

55

50

45

80

70

75

65

0dBFS

–6dBFS

–12dBFS

Figure 4. SFDR vs. fOUT @ 5 MSPS

FREQUENCY – MHz

SF

DR

– d

Bc

90

85

400 10 5020 30 40

60

55

50

45

80

70

75

65

0dBFS

–6dBFS

–12dBFS

Figure 7. SFDR vs. fOUT @100 MSPS

AOUT – dBFS

SF

DR

– d

Bc

90

80

50–30 –25 0–20 –15 –10 –5

70

60

1MHz @ 5 MSPS

5MHz @ 25 MSPS

10MHz @ 50 MSPS

20MHz @ 100 MSPS

Figure 10. Single-Tone SFDR vs. AOUT @ fOUT = fCLOCK/5

FREQUENCY – MHz

SF

DR

– d

Bc

90

400 2 124 6 8 10

85

60

55

50

45

80

75

65

70

0dBFS

–6dBFS

–12dBFS

Figure 5. SFDR vs. fOUT @ 25 MSPS

FREQUENCY – MHz

90

85

500.0 2.0 10.04.0 6.0 8.0

70

65

60

55

80

75

5mA @ 3V

5mA @ 5V

10mA @ 3V

10mA @ 5V

20mA @ 3V

20mA @ 5V

SF

DR

– d

Bc

Figure 8. SFDR vs. fOUT andIOUTFS @ 25 MSPS and 0 dBFS

AOUT – dBFS

SF

DR

– d

Bc

90

80

50–30 –25 0–20 –15 –10 –5

70

60

0.675/0.725MHz @ 5 MSPS

3.38/3.63MHz @ 25 MSPS

13.5/14.5MHz @ 100 MSPS

6.75/7.25 @ 50 MSPS

Figure 11. Dual-Tone SFDR vs. AOUT

@ fOUT = fCLOCK/7

REV. C

AD9764

–8–

dBc

–70

–75

–95000.0E+0 40.0E+6 80.0E+6 120.0E+6

–80

–85

–90

2ND HARMONIC

3RD HARMONIC

4TH HARMONIC

Figure 12. THD vs. fCLOCK @ fOUT = 2 MHz

CODE

ER

RO

R –

LS

B

2.0

–2.00 160004000 8000 12000

1.5

0.0

–0.5

–1.0

–1.5

1.0

0.5

Figure 15. Typical INL

000.0E+0 7.5E+6 15.0E+6 22.5E+6

10dB

– D

iv

0

–10

–90

–50

–60

–70

–80

–20

–40

–30

fCLK = 50MSPS fOUT = 1.25MHz SFDR = 78dBc AMPLITUDE = 0dBFS

Figure 18. Single-Tone SFDR

fCLOCK – MSPS

SN

R –

dB

85

80

600 10 10020 30 40 50 60 70 80 90

75

70

65

IOUTFS = 5mA,DVDD = +5V

IOUTFS = 5mA,DVDD = +3V

IOUTFS = 10mA, DVDD = +5V

IOUTFS = 10mA,DVDD = +3V

IOUTFS = 20mA,DVDD = +5V

IOUTFS = 20mA,DVDD = +3V

Figure 13. SNR vs. fCLOCK @ fOUT = 2.0 MHz

CODE

ER

RO

R –

LS

B

2.0

–1.00 160004000 8000 12000

1.5

1.0

0.5

0.0

–0.5

Figure 16. Typical DNL

0E+0 25E+65E+6 10E+6 15E+6 20E+6

10dB

– D

iv

0

–10

–90

–50

–60

–70

–80

–20

–40

–30

fCLK = 50MSPS fOUT1 = 6.75MHz fOUT2 = 7.25MHz SFDR = 69dBc AMPLITUDE = 0dBFS

Figure 19. Dual-Tone SFDR

OUTPUT FREQUENCY – MHz

SF

DR

– d

Bc

80

70

501 10 100

60

IDIFF @ 0dBFS

IDIFF @ –6dBFS

IA @ 0dBFS

IA @ –6dBFS

Figure 14. Differential vs. Single-Ended SFDR vs. fOUT @ 50 MSPS

TEMPERATURE – 8CS

FD

R –

dB

c

80

75

50–40 –20 8060

70

65

60

55

40200

2.5MHz

10MHz

40MHz

Figure 17. SFDR vs. Temperature@ 100 MSPS, 0 dBFS

10dB

– D

iv

0

–70

–100

–10

–60

–80

–90

–40

–50

–20

–30

000.0E+0 7.5E+6 15.0E+6 22.5E+6

fCLK = 50MSPS fOUT1 = 6.25MHz fOUT2 = 6.75MHz fOUT3 = 7.25MHz fOUT4 = 7.75MHz SFDR = 66dBc AMPLITUDE = 0dBFS

Figure 20. Four-Tone SFDR

REV. C

AD9764

–9–

FUNCTIONAL DESCRIPTIONFigure 21 shows a simplified block diagram of the AD9764. TheAD9764 consists of a large PMOS current source array that iscapable of providing up to 20 mA of total current. The arrayis divided into 31 equal currents that make up the five mostsignificant bits (MSBs). The next four bits or middle bits consistof 15 equal current sources whose value is 1/16th of an MSBcurrent source. The remaining LSBs are binary weighted frac-tions of the middle bits current sources. Implementing themiddle and lower bits with current sources, instead of an R-2Rladder, enhances its dynamic performance for multitone or lowamplitude signals and helps maintain the DAC’s high outputimpedance (i.e., >100 kΩ).

All of these current sources are switched to one or the other ofthe two output nodes (i.e., IOUTA or IOUTB) via PMOS differen-tial current switches. The switches are based on a new architec-ture that drastically improves distortion performance. This newswitch architecture reduces various timing errors and providesmatching complementary drive signals to the inputs of the dif-ferential current switches.

The analog and digital sections of the AD9764 have separatepower supply inputs (i.e., AVDD and DVDD) that can operateindependently over a 2.7 volt to 5.5 volt range. The digitalsection, which is capable of operating up to a 125 MSPS clockrate, consists of edge-triggered latches and segment decodinglogic circuitry. The analog section includes the PMOS currentsources, the associated differential switches, a 1.20 V bandgapvoltage reference and a reference control amplifier.

The full-scale output current is regulated by the reference con-trol amplifier and can be set from 2 mA to 20 mA via an exter-nal resistor, RSET. The external resistor, in combination withboth the reference control amplifier and voltage referenceVREFIO, sets the reference current IREF, which is mirrored over tothe segmented current sources with the proper scaling factor.The full-scale current, IOUTFS, is 32 times the value of IREF.

DAC TRANSFER FUNCTIONThe AD9764 provides complementary current outputs, IOUTA

and IOUTB. IOUTA will provide a near full-scale current output,IOUTFS, when all bits are high (i.e., DAC CODE = 16383) whileIOUTB, the complementary output, provides no current. Thecurrent output appearing at IOUTA and IOUTB is a function ofboth the input code and IOUTFS and can be expressed as:

IOUTA = (DAC CODE/16384) × IOUTFS (1)

IOUTB = (16383 – DAC CODE)/16384 × IOUTFS (2)

where DAC CODE = 0 to 16383 (i.e., Decimal Representation).

As mentioned previously, IOUTFS is a function of the referencecurrent IREF, which is nominally set by a reference voltageVREFIO and external resistor RSET. It can be expressed as:

IOUTFS = 32 × IREF (3)

where IREF = VREFIO/RSET (4)

The two current outputs will typically drive a resistive loaddirectly or via a transformer. If dc coupling is required, IOUTA

and IOUTB should be directly connected to matching resistiveloads, RLOAD, that are tied to analog common, ACOM. Notethat RLOAD may represent the equivalent load resistance seen byIOUTA or IOUTB as would be the case in a doubly terminated50 Ω or 75 Ω cable. The single-ended voltage output appearingat the IOUTA and IOUTB nodes is simply:

VOUTA = IOUTA × RLOAD (5)

VOUTB = IOUTB × RLOAD (6)

Note that the full-scale value of VOUTA and VOUTB should notexceed the specified output compliance range to maintain speci-fied distortion and linearity performance.

The differential voltage, VDIFF, appearing across IOUTA andIOUTB is:

VDIFF = (IOUTA – IOUTB) × RLOAD (7)

Substituting the values of IOUTA, IOUTB and IREF; VDIFF can beexpressed as:

VDIFF = (2 DAC CODE – 16383)/16384 ×VDIFF = (32 RLOAD/RSET) × VREFIO (8)

These last two equations highlight some of the advantages ofoperating the AD9764 differentially. First, the differential op-eration will help cancel common-mode error sources associatedwith IOUTA and IOUTB such as noise, distortion and dc offsets.Second, the differential code-dependent current and subsequentvoltage, VDIFF, is twice the value of the single-ended voltageoutput (i.e., VOUTA or VOUTB), thus providing twice the signalpower to the load.

Note that the gain drift temperature performance for a single-ended (VOUTA and VOUTB) or differential output (VDIFF) of theAD9764 can be enhanced by selecting temperature trackingresistors for RLOAD and RSET due to their ratiometric relation-ship as shown in Equation 8.

DIGITAL DATA INPUTS (DB13–DB0)

50pF

COMP1

+1.20V REF

AVDD ACOMREFLO

COMP2PMOS

CURRENT SOURCEARRAY

0.1mF

+5V

SEGMENTED SWITCHESFOR DB13–DB5

LSBSWITCHES

REFIO

FS ADJ

DVDD

DCOM

CLOCK

+5VRSET2kV

0.1mF

IOUTA

IOUTB

0.1mF

AD9764

SLEEP

IREF

VREFIO

CLOCK

IOUTB

IOUTA

RLOAD50V

VOUTB

VOUTA

RLOAD50V

VDIFF = VOUTA – VOUTB

LATCHES

Figure 21. Functional Block Diagram

REV. C

AD9764

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REFERENCE OPERATIONThe AD9764 contains an internal 1.20 V bandgap referencethat can be easily disabled and overridden by an externalreference. REFIO serves as either an input or output, dependingon whether the internal or external reference is selected. IfREFLO is tied to ACOM, as shown in Figure 22, the internalreference is activated, and REFIO provides a 1.20 V output. Inthis case, the internal reference must be compensated externallywith a ceramic chip capacitor of 0.1 µF or greater from REFIOto REFLO. Also, REFIO should be buffered with an externalamplifier having an input bias current less than 100 nA if anyadditional loading is required.

50pF

COMP1

+1.2V REF

AVDDREFLO

CURRENTSOURCEARRAY

0.1mF

+5V

REFIO

FS ADJ

2kV

0.1mF

AD9764

ADDITIONALLOAD

OPTIONALEXTERNAL

REF BUFFER

Figure 22. Internal Reference Configuration

The internal reference can be disabled by connecting REFLO toAVDD. In this case, an external reference may then be appliedto REFIO as shown in Figure 23. The external reference mayprovide either a fixed reference voltage to enhance accuracy anddrift performance or a varying reference voltage for gain control.Note that the 0.1 µF compensation capacitor is not requiredsince the internal reference is disabled, and the high input im-pedance (i.e., 1 MΩ) of REFIO minimizes any loading of theexternal reference.

50pF

COMP1

+1.2V REF

AVDDREFLO

CURRENTSOURCEARRAY

0.1mF

AVDD

REFIO

FS ADJ

RSET

AD9764

EXTERNALREF

IREF =VREFIO/RSET

AVDD

REFERENCECONTROLAMPLIFIER

VREFIO

Figure 23. External Reference Configuration

REFERENCE CONTROL AMPLIFIERThe AD9764 also contains an internal control amplifier that isused to regulate the DAC’s full-scale output current, IOUTFS.The control amplifier is configured as a V-I converter, as shownin Figure 23, such that its current output, IREF, is determined bythe ratio of the VREFIO and an external resistor, RSET, as statedin Equation 4. IREF is copied over to the segmented currentsources with the proper scaling factor to set IOUTFS as stated inEquation 3.

The control amplifier allows a wide (10:1) adjustment span ofIOUTFS over a 2 mA to 20 mA range by setting IREF between62.5 µA and 625 µA. The wide adjustment span of IOUTFS

provides several application benefits. The first benefit relatesdirectly to the power dissipation of the AD9764, which is pro-portional to IOUTFS (refer to the Power Dissipation section). Thesecond benefit relates to the 20 dB adjustment, which is usefulfor system gain control purposes.

The small signal bandwidth of the reference control amplifier isapproximately 1.4 MHz and can be reduced by connecting anexternal capacitor between COMP1 and AVDD. The output ofthe control amplifier, COMP1, is internally compensated via a50 pF capacitor that limits the control amplifier small-signalbandwidth and reduces its output impedance. Any additionalexternal capacitance further limits the bandwidth and acts as afilter to reduce the noise contribution from the reference ampli-fier. Figure 24 shows the relationship between the externalcapacitor and the small signal –3 dB bandwidth of the refer-ence amplifier. Since the –3 dB bandwidth corresponds to thedominant pole, and hence the time constant, the settling time ofthe control amplifier to a stepped reference input response canbe approximated.

COMP1 CAPACITOR – nF

1000

10

0.10.1 10001

BA

ND

WID

TH

– k

Hz

10 100

Figure 24. External COMP1 Capacitor vs. –3 dB Bandwidth

The optimum distortion performance for any reconstructedwaveform is obtained with a 0.1 µF external capacitor installed.Thus, if IREF is fixed for an application, a 0.1 µF ceramic chipcapacitor is recommended. Also, since the control amplifier isoptimized for low power operation, multiplying applicationsrequiring large signal swings should consider using an externalcontrol amplifier to enhance the application’s overall large signalmultiplying bandwidth and/or distortion performance.

There are two methods in which IREF can be varied for a fixedRSET. The first method is suitable for a single-supply system inwhich the internal reference is disabled, and the common-modevoltage of REFIO is varied over its compliance range of 1.25 Vto 0.10 V. REFIO can be driven by a single-supply amplifier orDAC, thus allowing IREF to be varied for a fixed RSET. Since theinput impedance of REFIO is approximately 1 MΩ, a simple,low cost R-2R ladder DAC configured in the voltage modetopology may be used to control the gain. This circuit is shownin Figure 25 using the AD7524 and an external 1.2 V reference,the AD1580.

REV. C

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The second method may be used in a dual-supply system inwhich the common-mode voltage of REFIO is fixed, and IREF isvaried by an external voltage, VGC, applied to RSET via an ampli-fier. An example of this method is shown in Figure 26 in whichthe internal reference is used to set the common-mode voltageof the control amplifier to 1.20 V. The external voltage, VGC, isreferenced to ACOM and should not exceed 1.2 V. The value ofRSET is such that IREFMAX and IREFMIN do not exceed 62.5 µAand 625 µA, respectively. The associated equations in Figure 26can be used to determine the value of RSET.

50pF

COMP1 AVDDREFLO

CURRENTSOURCEARRAY

AVDD

REFIO

FS ADJ

RSET AD9764IREF

OPTIONALBANDLIMITING

CAPACITOR

VGC

1mF

IREF = (1.2–VGC)/RSET

WITH VGC < VREFIO AND 62.5mA # IREF # 625A

+1.2V REF

Figure 26. Dual-Supply Gain Control Circuit

In some applications, the user may elect to use an externalcontrol amplifier to enhance the multiplying bandwidth,distortion performance and/or settling time. External amplifierscapable of driving a 50 pF load such as the AD817 are suitablefor this purpose. It is configured in such a way that it is inparallel with the weaker internal reference amplifier as shown inFigure 27. In this case, the external amplifier simply overdrivesthe weaker reference control amplifier. Also, since the internalcontrol amplifier has a limited current output, it will sustain nodamage if overdriven.

50pF

COMP1

+1.2V REF

AVDDREFLO

CURRENTSOURCEARRAY

AVDD

REFIO

FS ADJ

RSET AD9764

VREFINPUT

EXTERNALCONTROL AMPLIFIER

Figure 27. Configuring an External Reference Control Amplifier

ANALOG OUTPUTSThe AD9764 produces two complementary current outputs,IOUTA and IOUTB, which may be configured for single-endor differential operation. IOUTA and IOUTB can be converted intocomplementary single-ended voltage outputs, VOUTA andVOUTB, via a load resistor, RLOAD, as described in the DACTransfer Function section by Equations 5 through 8. Thedifferential voltage, VDIFF, existing between VOUTA and VOUTB

can also be converted to a single-ended voltage via a transformeror differential amplifier configuration.

Figure 28 shows the equivalent analog output circuit of theAD9764 consisting of a parallel combination of PMOS differen-tial current switches associated with each segmented currentsource. The output impedance of IOUTA and IOUTB is determinedby the equivalent parallel combination of the PMOS switchesand is typically 100 kΩ in parallel with 5 pF. Due to the na-ture of a PMOS device, the output impedance is also slightlydependent on the output voltage (i.e., VOUTA and VOUTB) and, toa lesser extent, the analog supply voltage, AVDD, and full-scalecurrent, IOUTFS. Although the output impedance’s signal depen-dency can be a source of dc nonlinearity and ac linearity (i.e.,distortion), its effects can be limited if certain precautions arenoted.

AD9764

AVDD

IOUTA IOUTB

RLOAD RLOAD

Figure 28. Equivalent Analog Output Circuit

IOUTA and IOUTB also have a negative and positive voltage compli-ance range. The negative output compliance range of –1.0 V isset by the breakdown limits of the CMOS process. Operationbeyond this maximum limit may result in a breakdown of theoutput stage and affect the reliability of the AD9764. The posi-tive output compliance range is slightly dependent on the full-scale output current, IOUTFS. It degrades slightly from its nominal

1.2V50pF

COMP1

+1.2V REF

AVDDREFLO

CURRENTSOURCEARRAY

AVDD

REFIO

FS ADJ

RSET AD9764IREF =VREF/RSET

AVDD OPTIONALBANDLIMITING

CAPACITOR

VREF

VDDRFB

OUT1

OUT2

AGND

DB7–DB0

AD7524AD1580

0.1V TO 1.2V

Figure 25. Single-Supply Gain Control Circuit

REV. C

AD9764

–12–

1.25 V for an IOUTFS = 20 mA to 1.00 V for an IOUTFS = 2 mA.Operation beyond the positive compliance range will induceclipping of the output signal which severely degrades theAD9764’s linearity and distortion performance.

For applications requiring the optimum dc linearity, IOUTA and/or IOUTB should be maintained at a virtual ground via an I-V opamp configuration. Maintaining IOUTA and/or IOUTB at a virtualground keeps the output impedance of the AD9764 fixed, signifi-cantly reducing its effect on linearity. However, it does notnecessarily lead to the optimum distortion performance due tolimitations of the I-V op amp. Note that the INL/DNL speci-fications for the AD9764 are measured in this manner usingIOUTA. In addition, these dc linearity specifications remainvirtually unaffected over the specified power supply range of2.7 V to 5.5 V.

Operating the AD9764 with reduced voltage output swings atIOUTA and IOUTB in a differential or single-ended output configu-ration reduces the signal dependency of its output impedancethus enhancing distortion performance. Although the voltagecompliance range of IOUTA and IOUTB extends from –1.0 V to+1.25 V, optimum distortion performance is achieved when themaximum full-scale signal at IOUTA and IOUTB does not exceedapproximately 0.5 V. A properly selected transformer with agrounded center-tap will allow the AD9764 to provide the re-quired power and voltage levels to different loads while main-taining reduced voltage swings at IOUTA and IOUTB. DC-coupledapplications requiring a differential or single-ended output con-figuration should size RLOAD accordingly. Refer to Applying theAD9764 section for examples of various output configurations.

The most significant improvement in the AD9764’s distortionand noise performance is realized using a differential outputconfiguration. The common-mode error sources of both IOUTA

and IOUTB can be substantially reduced by the common-moderejection of a transformer or differential amplifier. Thesecommon-mode error sources include even-order distortionproducts and noise. The enhancement in distortion performancebecomes more significant as the reconstructed waveform’sfrequency content increases and/or its amplitude decreases.This is evident in Figure 14, which compares the differentialvs. single-ended performance of the AD9764 at 50 MSPS for0.0 and –6.0 dBFS single tone waveforms over frequency.

The distortion and noise performance of the AD9764 is alsoslightly dependent on the analog and digital supply as well as thefull-scale current setting, IOUTFS. Operating the analog supply at5.0 V ensures maximum headroom for its internal PMOS currentsources and differential switches leading to improved distortionperformance as shown in Figure 8. Although IOUTFS can be setbetween 2 mA and 20 mA, selecting an IOUTFS of 20 mA willprovide the best distortion and noise performance also shown inFigure 8. The noise performance of the AD9764 is affected bythe digital supply (DVDD), output frequency, and increaseswith increasing clock rate as shown in Figure 13. Operating theAD9764 with low voltage logic levels between 3 V and 3.3 Vwill slightly reduce the amount of on-chip digital noise.

In summary, the AD9764 achieves the optimum distortion andnoise performance under the following conditions:

(1) Differential Operation.

(2) Positive voltage swing at IOUTA and IOUTB limited to +0.5 V.

(3) IOUTFS set to 20 mA.

(4) Analog Supply (AVDD) set at 5.0 V.

(5) Digital Supply (DVDD) set at 3.0 V to 3.3 V with appro-priate logic levels.

Note that the ac performance of the AD9764 is characterizedunder the above mentioned operating conditions.

DIGITAL INPUTSThe AD9764’s digital input consists of 14 data input pins and aclock input pin. The 14-bit parallel data inputs follow standardpositive binary coding where DB13 is the most significant bit(MSB), and DB0 is the least significant bit (LSB). IOUTA pro-duces a full-scale output current when all data bits are at Logic1. IOUTB produces a complementary output with the full-scalecurrent split between the two outputs as a function of the inputcode.

The digital interface is implemented using an edge-triggeredmaster slave latch. The DAC output is updated following therising edge of the clock as shown in Figure 1 and is designed tosupport a clock rate as high as 125 MSPS. The clock can beoperated at any duty cycle that meets the specified latch pulse-width. The setup and hold times can also be varied within theclock cycle as long as the specified minimum times are met,although the location of these transition edges may affect digitalfeedthrough and distortion performance. Best performance istypically achieved when the input data transitions on the fallingedge of a 50% duty cycle clock.

The digital inputs are CMOS-compatible with logic thresholds,VTHRESHOLD, set to approximately half the digital positive supply(DVDD) or

VTHRESHOLD = DVDD/2 (±20%)

The internal digital circuitry of the AD9764 is capable of operatingover a digital supply range of 2.7 V to 5.5 V. As a result, thedigital inputs can also accommodate TTL levels when DVDD isset to accommodate the maximum high level voltage of the TTLdrivers VOH(MAX). A DVDD of 3 V to 3.3 V will typically ensureproper compatibility with most TTL logic families. Figure 29shows the equivalent digital input circuit for the data and clockinputs. The sleep mode input is similar with the exception thatit contains an active pull-down circuit, thus ensuring that theAD9764 remains enabled if this input is left disconnected.

DVDD

DIGITALINPUT

Figure 29. Equivalent Digital Input

REV. C

AD9764

–13–

Since the AD9764 is capable of being updated up to 125 MSPS,the quality of the clock and data input signals are important inachieving the optimum performance. Operating the AD9764with reduced logic swings and a corresponding digital supply(DVDD) will result in the lowest data feedthrough and on-chipdigital noise. The drivers of the digital data interface circuitryshould be specified to meet the minimum setup and hold timesof the AD9764 as well as its required min/max input logic levelthresholds.

Digital signal paths should be kept short and run lengthsmatched to avoid propagation delay mismatch. The insertionof a low value resistor network (i.e., 20 Ω to 100 Ω) betweenthe AD9764 digital inputs and driver outputs may be helpful inreducing any overshooting and ringing at the digital inputs thatcontribute to data feedthrough. For longer run lengths and highdata update rates, strip line techniques with proper terminationresistors should be considered to maintain “clean” digital inputs.

The external clock driver circuitry should provide the AD9764with a low jitter clock input meeting the min/max logic levelswhile providing fast edges. Fast clock edges will help minimizeany jitter that will manifest itself as phase noise on a recon-structed waveform. Thus, the clock input should be driven bythe fastest logic family suitable for the application.

Note, that the clock input could also be driven via a sine wave,which is centered around the digital threshold (i.e., DVDD/2)and meets the min/max logic threshold. This will typically resultin a slight degradation in the phase noise, which becomes morenoticeable at higher sampling rates and output frequencies.Also, at higher sampling rates, the 20% tolerance of the digitallogic threshold should be considered since it will affect the effec-tive clock duty cycle and, subsequently, cut into the requireddata setup and hold times.

SLEEP MODE OPERATIONThe AD9764 has a power-down function that turns off theoutput current and reduces the supply current to less than8.5 mA over the specified supply range of 2.7 V to 5.5 V andtemperature range. This mode can be activated by applying alogic level “1” to the SLEEP pin. This digital input also con-tains an active pull-down circuit that ensures the AD9764 re-mains enabled if this input is left disconnected. The SLEEPinput with active pull-down requires <40 µA of drive current.

The power-up and power-down characteristics of the AD9764are dependent upon the value of the compensation capacitorconnected to COMP1. With a nominal value of 0.1 µF, theAD9764 takes less than 5 µs to power down and approximately3.25 ms to power back up. Note, the SLEEP MODE should notbe used when the external control amplifier is used as shown inFigure 27.

POWER DISSIPATIONThe power dissipation, PD, of the AD9764 is dependent onseveral factors, including: (1) AVDD and DVDD, the powersupply voltages; (2) IOUTFS, the full-scale current output; (3)fCLOCK, the update rate; and (4) the reconstructed digital inputwaveform. The power dissipation is directly proportional to theanalog supply current, IAVDD, and the digital supply current,IDVDD. IAVDD is directly proportional to IOUTFS, as shown inFigure 30, and is insensitive to fCLOCK.

IOUTFS – mA

30

02 204 6 8 10 12 14 16 18

25

20

15

10

5

I AV

DD

– m

A

Figure 30. IAVDD vs. IOUTFS

Conversely, IDVDD is dependent on both the digital input wave-form, fCLOCK, and digital supply DVDD. Figures 31 and 32show IDVDD as a function of full-scale sine wave output ratios(fOUT/fCLOCK) for various update rates with DVDD = 5 V andDVDD = 3 V, respectively. Note, how IDVDD is reduced by morethan a factor of 2 when DVDD is reduced from 5 V to 3 V.

RATIO – fOUT/fCLK

18

16

00.01 10.1

I DV

DD

– m

A

8

6

4

2

12

10

14

5MSPS

25MSPS

50MSPS

100MSPS

125MSPS

Figure 31. IDVDD vs. Ratio @ DVDD = 5 V

RATIO – fOUT/fCLK

8

00.01 10.1

I DV

DD

– m

A

6

4

2

5MSPS

25MSPS

50MSPS

100MSPS

125MSPS

Figure 32. IDVDD vs. Ratio @ DVDD = 3 V

REV. C

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–14–

APPLYING THE AD9764OUTPUT CONFIGURATIONSThe following sections illustrate some typical output configura-tions for the AD9764. Unless otherwise noted, it is assumedthat IOUTFS is set to a nominal 20 mA. For applications requir-ing the optimum dynamic performance, a differential outputconfiguration is suggested. A differential output configurationmay consist of either an RF transformer or a differential op ampconfiguration. The transformer configuration provides the opti-mum high frequency performance and is recommended for anyapplication allowing for ac coupling. The differential op ampconfiguration is suitable for applications requiring dc coupling, abipolar output, signal gain and/or level shifting.

A single-ended output is suitable for applications requiring aunipolar voltage output. A positive unipolar output voltage willresult if IOUTA and/or IOUTB is connected to an appropriatelysized load resistor, RLOAD, referred to ACOM. This configura-tion may be more suitable for a single-supply system requiring adc coupled, ground referred output voltage. Alternatively, anamplifier could be configured as an I-V converter, thus convert-ing IOUTA or IOUTB into a negative unipolar voltage. This con-figuration provides the best dc linearity since IOUTA or IOUTB ismaintained at a virtual ground. Note, IOUTA provides slightlybetter performance than IOUTB.

DIFFERENTIAL COUPLING USING A TRANSFORMERAn RF transformer can be used to perform a differential-to-single-ended signal conversion as shown in Figure 33. Adifferentially coupled transformer output provides the optimumdistortion performance for output signals whose spectral contentlies within the transformer’s passband. An RF transformer suchas the Mini-Circuits T1-1T provides excellent rejection of com-mon-mode distortion (i.e., even-order harmonics) and noiseover a wide frequency range. It also provides electrical isolationand the ability to deliver twice the power to the load. Trans-formers with different impedance ratios may also be used forimpedance matching purposes. Note that the transformerprovides ac coupling only.

RLOADAD9764

22

21

MINI-CIRCUITST1-1T

OPTIONAL RDIFF

IOUTA

IOUTB

Figure 33. Differential Output Using a Transformer

The center tap on the primary side of the transformer must beconnected to ACOM to provide the necessary dc current pathfor both IOUTA and IOUTB. The complementary voltages appear-ing at IOUTA and IOUTB (i.e., VOUTA and VOUTB) swing sym-metrically around ACOM and should be maintained with thespecified output compliance range of the AD9764. A differentialresistor, RDIFF, may be inserted in applications in which theoutput of the transformer is connected to the load, RLOAD, via apassive reconstruction filter or cable. RDIFF is determined by thetransformer’s impedance ratio and provides the proper sourcetermination that results in a low VSWR. Note that approxi-mately half the signal power will be dissipated across RDIFF.

DIFFERENTIAL USING AN OP AMPAn op amp can also be used to perform a differential-to-single-ended conversion as shown in Figure 34. The AD9764 is con-figured with two equal load resistors, RLOAD, of 25 Ω. Thedifferential voltage developed across IOUTA and IOUTB is con-verted to a single-ended signal via the differential op amp con-figuration. An optional capacitor can be installed across IOUTA

and IOUTB, forming a real pole in a low-pass filter. The additionof this capacitor also enhances the op amp’s distortion perfor-mance by preventing the DAC’s high slewing output from over-loading the op amp’s input.

AD9764

22IOUTA

IOUTB 21COPT

500V

225V

225V

500V

25V25V

AD8047

Figure 34. DC Differential Coupling Using an Op Amp

The common-mode rejection of this configuration is typicallydetermined by the resistor matching. In this circuit, the differ-ential op amp circuit using the AD8047 is configured to providesome additional signal gain. The op amp must operate from adual supply since its output is approximately ±1.0 V. A highspeed amplifier capable of preserving the differential perform-ance of the AD9764 while meeting other system level objectives(i.e., cost, power) should be selected. The op amps differentialgain, its gain setting resistor values and full-scale output swingcapabilities should all be considered when optimizing this circuit.

The differential circuit shown in Figure 35 provides the neces-sary level-shifting required in a single supply system. In thiscase, AVDD, which is the positive analog supply for both theAD9764 and the op amp, is also used to level-shift the differ-ential output of the AD9764 to midsupply (i.e., AVDD/2). TheAD8041 is a suitable op amp for this application.

AD9764

22IOUTA

IOUTB 21COPT

500V

225V

225V

1kV25V25V

AD8041

1kV

AVDD

Figure 35. Single-Supply DC Differential Coupled Circuit

SINGLE-ENDED UNBUFFERED VOLTAGE OUTPUTFigure 36 shows the AD9764 configured to provide a unipolaroutput range of approximately 0 V to +0.5 V for a doubly termi-nated 50 Ω cable since the nominal full-scale current, IOUTFS, of20 mA flows through the equivalent RLOAD of 25 Ω. In this case,RLOAD represents the equivalent load resistance seen by IOUTA orIOUTB. The unused output (IOUTA or IOUTB) can be connected toACOM directly or via a matching RLOAD. Different values of

REV. C

AD9764

–15–

IOUTFS and RLOAD can be selected as long as the positive compli-ance range is adhered to. One additional consideration in thismode is the integral nonlinearity (INL) as discussed in the Ana-log Output section of this data sheet. For optimum INL perfor-mance, the single-ended, buffered voltage output configurationis suggested.

AD9764

IOUTA

IOUTB 21

50V

25V

50V

VOUTA = 0 TO +0.5VIOUTFS = 20mA

22

Figure 36. 0 V to +0.5 V Unbuffered Voltage Output

SINGLE-ENDED BUFFERED VOLTAGE OUTPUTCONFIGURATIONFigure 37 shows a buffered single-ended output configuration inwhich the op amp U1 performs an I-V conversion on theAD9764 output current. U1 maintains IOUTA (or IOUTB) at avirtual ground, thus minimizing the nonlinear output impedanceeffect on the DAC’s INL performance as discussed in the Ana-log Output section. Although this single-ended configurationtypically provides the best dc linearity performance, its ac distor-tion performance at higher DAC update rates may be limited byU1’s slewing capabilities. U1 provides a negative unipolaroutput voltage and its full-scale output voltage is simply theproduct of RFB and IOUTFS. The full-scale output should be setwithin U1’s voltage output swing capabilities by scaling IOUTFS

and/or RFB. An improvement in ac distortion performance mayresult with a reduced IOUTFS since the signal current U1 will berequired to sink will be subsequently reduced.

AD9764

22IOUTA

IOUTB 21

COPT

200V

U1 VOUT = IOUTFS 3 RFB

IOUTFS = 10mA

RFB200V

Figure 37. Unipolar Buffered Voltage Output

POWER AND GROUNDING CONSIDERATIONSIn systems seeking to simultaneously achieve high speed andhigh performance, the implementation and construction of theprinted circuit board design is often as important as the circuitdesign. Proper RF techniques must be used in device selection,placement and routing and supply bypassing and grounding.Figures 42–47 illustrate the recommended printed circuit boardground, power and signal plane layouts that are implemented onthe AD9764 evaluation board.

Proper grounding and decoupling should be a primary objectivein any high speed, high resolution system. The AD9764 featuresseparate analog and digital supply and ground pins to optimizethe management of analog and digital ground currents in a

system. In general, AVDD, the analog supply, should be decoupledto ACOM, the analog common, as close to the chip as physi-cally possible. Similarly, DVDD, the digital supply, should bedecoupled to DCOM as close as physically as possible.

For those applications requiring a single +5 V or +3 V supplyfor both the analog and digital supply, a clean analog supplymay be generated using the circuit shown in Figure 38. Thecircuit consists of a differential LC filter with separate powersupply and return lines. Lower noise can be attained using lowESR type electrolytic and tantalum capacitors.

100mFELECT.

10-22mFTANT.

0.1mFCER.

TTL/CMOSLOGIC

CIRCUITS

+5V OR +3VPOWER SUPPLY

FERRITEBEADS

AVDD

ACOM

Figure 38. Differential LC Filter for Single +5 V or +3 V Applications

Maintaining low noise on power supplies and ground is criticalto obtain optimum results from the AD9764. If properlyimplemented, ground planes can perform a host of functions onhigh speed circuit boards: bypassing, shielding current trans-port, etc. In mixed signal design, the analog and digital portionsof the board should be distinct from each other, with the analogground plane confined to the areas covering the analog signaltraces, and the digital ground plane confined to areas coveringthe digital interconnects.

All analog ground pins of the DAC, reference and other analogcomponents should be tied directly to the analog ground plane.The two ground planes should be connected by a path 1/8 to1/4 inch wide underneath or within 1/2 inch of the DAC tomaintain optimum performance. Care should be taken to ensurethat the ground plane is uninterrupted over crucial signal paths.On the digital side, this includes the digital input lines runningto the DAC as well as any clock signals. On the analog side, thisincludes the DAC output signal, reference signal and the supplyfeeders.

The use of wide runs or planes in the routing of power lines isalso recommended. This serves the dual role of providing a lowseries impedance power supply to the part, as well as providingsome “free” capacitive decoupling to the appropriate groundplane. It is essential that care be taken in the layout of signal andpower ground interconnects to avoid inducing extraneous volt-age drops in the signal ground paths. It is recommended that allconnections be short, direct and as physically close to the pack-age as possible in order to minimize the sharing of conductionpaths between different currents. When runs exceed an inch inlength, strip line techniques with proper termination resistorsshould be considered. The necessity and value of this resistorwill be dependent upon the logic family used.

For a more detailed discussion of the implementation and con-struction of high speed, mixed signal printed circuit boards,refer to Analog Devices’ application notes AN-280 and AN-333.

REV. C

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–16–

MULTITONE PERFORMANCE CONSIDERATIONS ANDCHARACTERIZATIONThe frequency domain performance of high speed DACs hastraditionally been characterized by analyzing the spectral outputof a reconstructed full-scale (i.e., 0 dBFS), single-tone sine waveat a particular output frequency and update rate. Although thischaracterization data is useful, it is often insufficient to reflect aDAC’s performance for a reconstructed multitone or spread-spectrum waveform. In fact, evaluating a DAC’s spectralperformance using a full-scale, single tone at the highest specifiedfrequency (i.e., fH) of a bandlimited waveform is typicallyindicative of a DAC’s “worst-case” performance for that givenwaveform. In the time domain, this full-scale sine wave repre-sents the lowest peak-to-rms ratio or crest factor (i.e., VPEAK/Vrms) that this bandlimited signal will encounter.

MA

GN

ITU

DE

– d

Bm

FREQUENCY – MHz

–10

–70

–1102.19 2.812.25 2.31 2.38 2.44 2.50 2.56 2.63 2.69 2.75

–20

–60

–80

–100

–40

–50

–90

–30

Figure 39a. Multitone Spectral Plot

TIME

1.0000

0.8000

–1.0000

VO

LTS

–0.2000

–0.4000

–0.6000

–0.8000

0.2000

0.0000

0.4000

0.6000

Figure 39b. Time Domain “Snapshot” of the MultitoneWaveform

However, the inherent nature of a multitone, spread spectrum,or QAM waveform, in which the spectral energy of the wave-form is spread over a designated bandwidth, will result in ahigher peak-to-rms ratio when compared to the case of a simplesine wave. As the reconstructed waveform’s peak-to-averageratio increases, an increasing amount of the signal energy isconcentrated around the DAC’s midscale value. Figure 39a isjust one example of a bandlimited multitone vector (i.e., eighttones) centered around one-half the Nyquist bandwidth (i.e.,fCLOCK/4). This particular multitone vector, has a peak-to-rms

ratio of 13.5 dB compared to a sine waves peak-to-rms ratio of3 dB. A “snapshot” of this reconstructed multitone vector in thetime domain as shown in Figure 39b reveals the higher signalcontent around the midscale value. As a result, a DAC’s“small-scale” dynamic and static linearity becomes increas-ingly critical in obtaining low intermodulation distortion andmaintaining sufficient carrier-to-noise ratios for a given modula-tion scheme.

A DAC’s small-scale linearity performance is also an importantconsideration in applications where additive dynamic range isrequired for gain control purposes or “predistortion” signalconditioning. For instance, a DAC with sufficient dynamicrange can be used to provide additional gain control of itsreconstructed signal. In fact, the gain can be controlled in6 dB increments by simply performing a shift left or right on theDAC’s digital input word. Other applications may intentionallypredistort a DAC’s digital input signal to compensate fornonlinearities associated with the subsequent analog compo-nents in the signal chain. For example, the signal compressionassociated with a power amplifier can be compensated for bypredistorting the DAC’s digital input with the inverse nonlineartransfer function of the power amplifier. In either case, theDAC’s performance at reduced signal levels should be carefullyevaluated.

A full-scale single tone will induce all of the dynamic and staticnonlinearities present in a DAC that contribute to its distortionand hence SFDR performance. Referring to Figure 3, as thefrequency of this reconstructed full-scale, single-tone waveformincreases, the dynamic nonlinearities of any DAC (i.e., AD9764)tend to dominate thus contributing to the rolloff in its SFDRperformance. However, unlike most DACs, which employ an R-2Rladder for the lower bit current segmentation, the AD9764 (aswell as other TxDAC members) exhibits an improvement indistortion performance as the amplitude of a single tone is re-duced from its full-scale level. This improvement in distortionperformance at reduced signal levels is evident if one comparesthe SFDR performance vs. frequency at different amplitudes(i.e., 0 dBFS, –6 dBFS and –12 dBFS) and sample rates asshown in Figures 4 through 7. Maintaining decent “small-scale”linearity across the full span of a DAC transfer function is alsocritical in maintaining excellent multitone performance.

Although characterizing a DAC’s multitone performance tendsto be application-specific, much insight into the potential per-formance of a DAC can also be gained by evaluating the DAC’sswept power (i.e., amplitude) performance for single, dual andmultitone test vectors at different clock rates and carrier frequen-cies. The DAC is evaluated at different clock rates when recon-structing a specific waveform whose amplitude is decreased in3 dB increments from full-scale (i.e., 0 dBFS). For each specificwaveform, a graph showing the SFDR (over Nyquist) perfor-mance vs. amplitude can be generated at the different testedclock rates as shown in Figures 9–11. Note that the carrier(s)-to-clock ratio remains constant in each figure. In each case, animprovement in SFDR performance is seen as the amplitude isreduced from 0 dBFS to approximately –9.0 dBFS.

A multitone test vector may consist of several equal amplitude,spaced carriers each representative of a channel within a definedbandwidth as shown in Figure 39a. In many cases, one or moretones are removed so the intermodulation distortion performance

REV. C

AD9764

–17–

of the DAC can be evaluated. Nonlinearities associated with theDAC will create spurious tones of which some may fall back intothe “empty” channel thus limiting a channel’s carrier-to-noiseratio. Other spurious components falling outside the band ofinterest may also be important, depending on the system’s spectralmask and filtering requirements.

This particular test vector was centered around one-half theNyquist bandwidth (i.e., fCLOCK/4) with a passband of fCLOCK/16.Centering the tones at a much lower region (i.e., fCLOCK/10)would lead to an improvement in performance while centeringthe tones at a higher region (i.e., fCLOCK/2.5) would result in adegradation in performance. Figure 40a shows the SFDR vs.amplitude at different sample rates up to the Nyquist frequencywhile Figure 40b shows the SFDR vs. amplitude within thepassband of the test vector. In assessing a DAC’s multitoneperformance, it is also recommended that several units be testedunder exactly the same conditions to determine any performancevariability.

AOUT – dBFS

SF

DR

– d

Bc

80

50

–20 –15 0–10 –5

70

60

10 MSPS

100 MSPS

20 MSPS

50 MSPS

40

30

75

65

55

45

35

Figure 40a. Multitone SFDR vs. AOUT (Up to Nyquist)

SF

DR

– d

Bc

50

70

60

40

75

65

55

45

80

10 MSPS20 MSPS

50 MSPS

100 MSPS

AOUT – dBFS–20 –15 0–10 –5

Figure 40b. Multitone SFDR vs. AOUT (Within Multitone Passband)

AD9764 EVALUATION BOARDGeneral DescriptionThe AD9764-EB is an evaluation board for the AD9764 14-bitDAC converter. Careful attention to layout and circuit design,combined with a prototyping area, allows the user to easily andeffectively evaluate the AD9764 in any application where highresolution, high speed conversion is required.

This board allows the user the flexibility to operate the AD9764in various configurations. Possible output configurations includetransformer coupled, resistor terminated, inverting/noninvertingand differential amplifier outputs. The digital inputs are designedto be driven directly from various word generators with theonboard option to add a resistor network for proper load termi-nation. Provisions are also made to operate the AD9764 witheither the internal or external reference or to exercise the power-down feature.

Refer to the application note AN-420, Using the AD9760/AD9764/AD9764-EB Evaluation Board for a thorough description andoperating instructions for the AD9764 evaluation board.

REV. C

AD9764

–18–

109

87

65

43

2

1

R4

109

87

65

43

2

1

R7

DV

DD

109

87

65

43

2

1

R3

109

87

65

43

2

1 DV

DD

R6

2 4 6 810 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40

1 3 5 7 9 11 13 15 17 19 21 23 25 27 29 31 33 35 37 39

P1

109

87

65

43

2

1

R5

DV

DD

109

87

65

43

2

1

R1

16 15 14 13 12 11 109

1 2 3 4 5 6 7 8

C19

C1

C2

C25

C26

C27

C28

C29

16 P

IND

IPR

ES

PK 16 15 14 13 12 11 10

1 2 3 4 5 6 7

C30

C31

C32

C33

C34

C35

C36

16 P

IND

IPR

ES

PK

1 2 3 4 5 6 7 8 910 11 12 13 14

28 27 26 25 24 23 22 21 20 19 18 17 16 15

DB

13D

B12

DB

11D

B10

DB

9D

B8

DB

7D

B6

DB

5D

B4

DB

3D

B2

DB

1D

B0

CLO

CK

DV

DD

DC

OM

NC

AV

DD

CO

MP

2IO

UT

AIO

UT

BA

CO

MC

OM

P1

FS

AD

JR

EF

IOR

EF

LOS

LEE

P

U1

AD

9764

AV

DD

CT

1

A1

A

R15

49.9

V

CLK

JP1

AB 3

21

J1T

P1

EX

TC

LK

C7

1mF

C8

0.1m

F

AV

DD

A

C9

0.1m

F

TP

8

2

AV

DD

TP

11

C11

0.1m

F

TP

10T

P9

R16

2kV TP

14

JP4

C10

0.1m

F

OU

T 1

OU

T 2

TP

13

R17

49.9

V

PD

INJ2

AA

AA

VD

D

3JP

2

TP

12

TP

7

AC

610

mF

AV

CC

B6

TP

6

AC

510

mF

AV

EE

B5

TP

19

A

AG

ND

B4 TP

18

TP

5C

410

mF

TP

4

AV

DD

B3

TP

2

DG

ND

B2

C3

10m

F

TP

3

DV

DD

B1

R20

49.9

V

J3

C12

22pF

AA

R14 0

A

4 5 6

13 T1

J7

R38

49.9

V

J4 AA

JP6A

JP6B

A

R13

OP

EN

C13

22pF

C20 0

R12

OP

EN

A

B A

JP7B

B A

JP7A

R10

1kV

B A

JP8

R9

1kV

A B

A

R35

1kV

JP9

R18

1kV

A

37

6

24

AD

8047

C21

0.1m

F

A

C22

1mF

R36

1kV C

230.

1mF

A

C24

1mF

AV

EE

AV

CC

R37

49.9

V

J6 A

37

6

24

12

3

JP5

C15

0.1m

F

A

AV

EE

R46

1kV

C17

0.1m

F

A123

JP3

AB

AV

CC

A

CW

R43

5kV

R45

1kV

C14

1mF

A

R44

50V

EX

TR

EF

INJ5 A

R42

1kV

C16

1mF

A

AV

CC

C18

0.1m

F

U7

62

4A

VIN

VO

UT

GN

D

RE

F43

98

76

54

32

1

R2

10

A

109

87

65

43

2

1 DV

DD

R8

U6

A

AD

8047

OU

T2

OU

T1

U4

Figure 41. Evaluation Board Schematic

REV. C

AD9764

–19–

Figure 42. Silkscreen Layer—Top

Figure 43. Component Side PCB Layout (Layer 1)

REV. C

AD9764

–20–

Figure 44. Ground Plane PCB Layout (Layer 2)

Figure 45. Power Plane PCB Layout (Layer 3)

REV. C

AD9764

–21–

Figure 46. Solder Side PCB Layout (Layer 4)

Figure 47. Silkscreen Layer—Bottom

REV. C

AD9764

–22–

OUTLINE DIMENSIONS

D13783-0-1/16(C)

PR

INT

ED

IN U

.S.A

.

28-Lead Standard Small Outline Package [SOIC_W]Wide Body

(RW-28)Dimensions shown in millimeters and (inches)

COMPLIANT TO JEDEC STANDARDS MS-013-AECONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FORREFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.

18.10 (0.7126)17.70 (0.6969)

0.30 (0.0118)0.10 (0.0039)

2.65 (0.1043)2.35 (0.0925)

10.65 (0.4193)10.00 (0.3937)

7.60 (0.2992)7.40 (0.2913)

0.75 (0.0295)0.25 (0.0098) 45°

1.27 (0.0500)0.40 (0.0157)

COPLANARITY0.10 0.33 (0.0130)

0.20 (0.0079)0.51 (0.0201)0.31 (0.0122)

SEATINGPLANE

8°0°

28 15

141

1.27 (0.0500)BSC

06-0

7-20

06-A

28-Lead Thin Shrink Small Outline Package [TSSOP](RU-28)

Dimensions shown in millimeters

COMPLIANT TO JEDEC STANDARDS MO-153-AE

28 15

141

8°0°SEATING

PLANECOPLANARITY

0.10

1.20 MAX

6.40 BSC

0.65BSC

PIN 1

0.300.19

0.200.09

4.504.404.30

0.750.600.45

9.809.709.60

0.150.05