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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 6, JUNE 2014 2891
Design of Polarization Reconfigurable Antenna
Using MetasurfaceH. L. Zhu, S. W. Cheung, Senior Member, IEEE, X. H. Liu, and T. I. Yuk, Member, IEEE
AbstractA planar polarization-reconfigurable metasurfaced
antenna (PRMS) designed using metasurface (MS) is proposed.The PRMS antenna consists of a planar MS placed atop of and indirect contact with a planar slot antenna, both having a circularshape with a diameter of 78 mm (0.9 ), making it compact andlow profile. By rotating the MS around the center with respect tothe slot antenna, the PRMS antenna can be reconfigured to linear
polarization, left-hand and right-hand circular polarizations. Anequivalent circuit is used to explain the reconfigurability of theantenna. The PRMS antenna is studied and designed to operateat around 3.5 GHz using computer simulation. For verification ofsimulation results, the PRMS antenna is fabricated and measured.
The antenna performance, in terms of polarization reconfigura-
bility, axial-ratio bandwidth, impedance bandwidth, realizedboresight gain and radiation pattern, is presented. Results showthat the PRMS antenna in circular polarizations achieves anoperating bandwidth of 3.33.7 GHz (i.e., fractional bandwidth11.4%), a boresight gain of above 5 dBi and high-polarization
isolation of larger than 15 dB. While the PRMS antenna in linearpolarization achieves a gain of above 7.5 dBi with cross-polariza-tion isolation larger than 50 dB.
Index TermsMetasurface (MS), metasurfaced antenna, polar-ization reconfiguration, source antenna.
I. INTRODUCTION
RECONFIGURABLE antenna generally has frequency,radiation pattern, or polarization tunability [1][4].
Since the postures of mobile communications devices in use
are often dynamically changing which creates difficulties
to have good polarization matching for antennas with only
single polarization, it is desirable to have antennas to be able
to work in different polarizations such as linear polarization
(LP) and circular polarization (CP). Study on antennas with
polarization-reconfigurable features has been attracting much
attention [4][20]. In these studies, RF switches such as PIN
diode switches [5][13], [15][20] and MEMS switches [4],
[14] were used to electrically switch polarization of the an-
tennas, thus direct-current (DC) biasing circuits were needed to
bias the PIN or MEMS switches. These switches and biasingcircuits made the whole antennas bulky and high cost. More-
over, the antenna operation would depend on the reliability
of the electronic components used to implement the switches
Manuscript received October 25, 2013; revised January 23, 2014; acceptedFebruary 27, 2014. Date of publication March 06, 2014; date of current version
May 29, 2014.Theauthorsare with theDepartmentof Electrical andElectronicEngineering,
The University of Hong Kong, Hong Kong, China (e-mail: [email protected];[email protected]; [email protected]; [email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2014.2310209
and require DC sources to drive the switches. The electronic
components and circuits may also have adverse effects on the
antennas performances. Electrically reconfigurable antennas
usually have narrow axial-ratio bandwidths (ARBWs) such as
those in [16][19] which had the fractional ARBWs of less
than 5%.
Compared to electrical reconfiguration, mechanical reconfig-
uration is way less popular. To the best knowledge of the au-
thors, there has not been any mechanically operated polariza-
tion reconfigurable antenna proposed. One of the main reasons
is that mechanically reconfigurable antennas require movable
parts. If not designed well, the actuator used to produce the me-chanical movements will be very complicated and occupy much
space, which lead to a bulky and expensive structure. More-
over, change of size and/or shape during tuning is the common
problem for most mechanically reconfigurable antennas. For
these reasons, the polarization reconfigurable antennas studied
so far were all electrical tuning [4][19], [21].
MS, a two-dimensional equivalent of metamaterial, is essen-
tially a surface distribution of electrically small scatterers [22].
With its succinct planar structure and low cost, MS has wide
applications and one of which is on the design of planar an-
tennas with improved performances. For example in [23], [24],
a MS was placed atop of a simple patch/slot antenna to not onlyimprove the gain and return loss bandwidth, but also convert
polarization from LP to CP. The patch/slot antenna (known as
the source antenna in such design) together with the MS was
called a metasurfaced antenna (MS antenna) [23], [24]. In these
designs, there was an air gap between the MS and the source
antenna, which substantially increased the volume of the MS
antenna.
In this paper, as a significant improvement to the work in [23],
[24], a novel polarization-reconfigurable metasurfaced (PRMS)
antenna constructed by placing a MS atop of a slot antenna is
proposed. To make the PRMS antenna compact and low profile,
the slot antenna and MS are placed together in direct contact,
thus eliminating the air gap between them. Results of studies
show that by rotating the MS around the center and relative
to the patch antenna, polarization of the PRMS antenna can
be reconfigured to operate in LP, left-hand circular polarization
(LHCP) and right-hand circular polarizations (RHCP).Compact
size, low cost and simple construction are the apparent advan-
tages of the proposed design. For easy mechanical operation,
the shapes of the slot antenna and the MS are made circular with
the same size. For verification of simulated results, the PRMS
antenna is fabricated and measured using the antenna measure-
ment equipment, Starlab system. Simulated and measured re-
sults show good agreements.
0018-926X 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
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Fig. 1. Geometries of (a) MS and (b) slot antenna.
II. DESIGN OF POLARIZATION-RECONFIGURABLE
METASURFACED (PRMS) ANTENNA
The polarization-reconfigurable metasurfaced (PRMS) an-
tenna proposed here consists of a slot antenna (as the source
antenna) and a metasurface (MS). The MS is composed of
corner-truncated square unit cells as shown in Fig. 1(a) on a
single-sided substrate. The same unit cell was used to design a
switchable microstrip antenna in [5] and a 2 2 sequentially
rotated patch antenna array in [25]. Here it is chosen to design
the MS because it has a simple structure and can be specified
using only a few parameters, significantly simplifying the de-
sign work for the PRMS antenna. The slot antenna is designed
on a double-sided substrate as shown in Fig. 1(b) and the MS
are designed to have a circular shape with the same size for
easy mechanical operation. The MS is similar to the MS used
in [24] with the edges of the unit cells at the boundary truncatedto make it circular. Of course, if needed, the PRMS antenna
could also be designed using a rectangular-shaped MS, but this
would be another design. Here, our design purpose is to use
a MS with circular shape for easy mechanical operation. As
will be seen later, polarization reconfigurablity of the antenna
can be accomplished by rotating the MS around the center
relative to the slot antenna. The rotation angle is measured
from the -axis as shown in Fig. 1(a). Studies have shown that
with and 90 , the PRMS antenna is left-hand cir-
cular polarization (LHCP) and right-hand circular polarization
(RHCP), respectively. Since the MS at is a mirror
image of itself at , the performances of the antennain RHCP and LHCP should be identical. With and
135 , the PRMS antenna is linear polarization (LP) along the
-axis, same with that of the source slot antenna. Fig. 2 shows
polarization of the antenna at , 45 , 90 , and 135 .
In assembling the PRMS antenna, the non-copper side of the
MS is placed atop the slot antenna and is in direct contact with
the feed-line (top side) of it as shown in Fig. 3. This leads to a
very compact and low profile structure. An SMA connector is
used to feed to the feed-line through the ground plane and sub-
strate material of the slot antenna. The PRMS antenna together
with the SMA connector is studied and designed using the EM
simulation tool CST on a Rogers substrate RO4350B, having a
thickness of 1.524 mm and a dielectric constant of .
The design procedure can be divided into the following steps:
Fig. 2. Antenna polarization at different rotation angles .
Fig. 3. Assembly schematic of PRMS antenna.
TABLE IDIMENSIONS OFFRMS ANTENNA (Unit: mm)
1) Design the slot antenna to have a wide operating band at
around 3.5 GHz.
2) Add the MS atop the slot antenna (which will change the
characteristic of the antenna) and then optimize the PRMS
antenna in terms of S11 using the dimensions of the radi-
ator and feed line.
3) Optimize the ARBW using the dimension of the MS.
The dimensions of thefinal design are listed in Table I which
is used to fabricate the PRMS antenna shown in Fig. 4 for mea-
surement using the antenna measurement equipment, Satimo
Starlab system.
III. ANALYSIS OF PRMS ANTENNA
A. Parametric Study
As mentioned previously, the truncated-corner unit cell used
to design our MS can be specified using only a few parameters.
This can significantly simplify the design work of the PRMS.
Computer simulation has shown that there are only three pa-
rameters in the unit cell of Fig. 1(a) having significantly effects
on the antenna performance. They are the width of the unit
cell, the side of the triangle, and the spacing between the
unit cells. Thus parametric studies of these parameters on the
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Fig. 4. (a) Top view of prototyped slot antenna, (b) bottom view of prototyped
slot antenna, and (c) prototyped metasurface.
axial ratio (AR) are carried out on the proposed PRMS antenna
with LHCP. The results are shown in Fig. 5 where the green
lines show the AR of the proposed PRMS antenna using the
parameters of Table I. It can be seen that the proposed PRMS
antenna has a AR bandwidth (ARBW) for dB from 3.3
to 3.7 GHz, two dips of 0.9 and 0.5 dB at 3.4 and 3.65 GHz,
respectively, and a peak of 1.7 dB at 3.53 GHz between the two
dips.As decreases from 20 mm to 18.5 mm, Fig. 5(a) shows
that the high- and low-frequency dips shift up from 3.18 and
3.56 GHz to 3.4 and 3.65 GHz, respectively, with the frequency
spacing between the two dips reduced from 380 to 250 MHz.
The peak between two dips shifts from 3.4 to 3.53 GHz with
amplitude dropped from 4.7 to 1.7 dB, respectively, resulting
in a large ARBW of 400 MHz. When decreases further to
17 mm, the two dips merge together with a minimum AR of
0.7 dB at 3.68 GHz. The antenna has an AR frequency band (for
dB) from 3.5 to 3.82 GHz, with the ARBW reduced
to 320 MHz. Thus, the parameter can be used to adjust the
frequency band of AR. Fig. 5(b) shows the AR with differentvalues of . It can be seen that as increases from 4.6 mm to
5, 5.3, and 5.6 mm, the AR at the frequency of 3.5 GHz drops
continuously. The two dips of AR move towards each other and
thus the ARBW reduces. When continues to increase to 6 mm,
the two dips merge together with a minimum AR of 1.2 dB at
around 3.5 GHz. Note that as changes, the frequency band for
dB is not changed much, with the center frequency
remained at about 3.5 GHz, unlike the case of changing in
Fig. 5(a). Thus the parameter can be used to slightly tradeoff
the value of AR against ARBW. TheAR with different values of
is shown in Fig.5(c).It can be seenthat has the similar effects
to , but in an opposite way. When increases from 0.5 mm to
0.75, 1 and 1.5 mm, the two dips in AR move towards each other
and the peak between them drops. The ARBW decreases with
Fig. 5. Simulated AR with different (a) , (b) , and (c) .
the AR within the AR frequency band reduced. Thus, can also
be used to slightly tradeoff the value of AR against ARBW.
Since the MS at and 90 for RHCP and LHCP,
respectively, of the MS are the mirror images of each other, the
results of parametric study on AR for RHCP should be the same
as those of LHCP.
B. Equivalent Circuit of MS
Here we use an equivalent circuit to explain how theproposed
PRMS antenna can have different polarizations when the MS
is rotated. In Fig. 1(a), the slot antenna is linear polarization
along the -axis. To start this, the MS in Fig. 1(a) is redrawn in
Fig. 6(a) where the pattern enclosed by the blue square can be
regarded as a new unit cell on the same MS. For convenience in
description, the new unit cell is enlarged and shown in Fig. 6(b).
When the MS is placed atop the slot antenna which is linear
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Fig. 6. (a) MS, (b) new unit cell with diagonal corner truncated, and (c) new
unit cell without truncation.
polarization along the -axis, an -field will be developed along
the -axis as shown in Fig. 6(b). The -field can be resolved
into two orthogonal components and as shown in the
same figure. If the diagonal corners of the unit cell were not
truncated as shown in Fig. 6(c), due to the symmetrical structure,
the orthogonal components and would see the MS as
an identical RLC circuit shown in Fig. 6(c) with an impedance
given by
(1)
where and are the inductance and resistance, respectively,
of each patch, is the capacitance created by the gaps between
two adjacent and opposite patches. However, if the diagonal cor-
ners are truncated as in our proposed MS shown in Fig. 6(b),
and will see two different impedances and , respec-
tively, which can be written as
(2)
(3)
Truncating the corners widens the gaps between the adjacent
and opposite patches and hence increases the value of in
(2), making less capacitive than . Thus, we can use the di-
mensions of the truncated corners (which is determined by the
parameter in Table I) to vary the phase difference between
and . If the MS is designed such that and
, then and .
will lead by 90 and the resultant -field throughthe MS
will be LHCP and rotating in the clockwise direction as indi-
cated by the yellow arrowed arc in Fig. 6(b). To reconfigure the
antenna to RHCP, we can simply rotate the MS by , so
that the value of in (3) becomes larger and will lead
by 90 instead. When the rotation angle is or 135 ,
the unit cell is symmetrical along the and axes as shown in
Fig. 7. Thus, both and see an identical impedance and
so have the same amplitude and phase after going through the
MS. The PRMS antenna remains LP.
IV. SIMULATION ANDMEASUREMENT RESULTS
A. Reflection Coef ficient S11
The final design of proposed PRMS antenna has been studied
using computer simulation and measurement. The simulated
Fig. 7. E-field for (a) and (b) .
and measured S11 of the antenna with different rotation angles
are shown in Fig. 8. It can be seen that the simulated and
measured results agree well. Since the MS at and 90
are the mirror images of each other, the simulated S11 for LHCP
and RHCP are identical as shown in Fig. 8(a) and (c), respec-
tively. The simulated frequency band (for dB) are
more than from 34 GHz. The measured S11 at these two rota-
tion angles are not exactly the same because of the fabrication
and measurement tolerances. At for LP, Fig. 8(b)
shows that the frequency band shifts down slightly and is from3 to 3.76 GHz for simulation and 3.74 GHz for measurement.
However, at , Fig. 8(d) shows that the S11 frequency
band shifts up slightly, from 3.28 to 4 GHz for simulation and
from 3.3 to 4.2 GHz for measurement.
B. Axial Ratio (AR)
The simulated and measured ARs in the boresight (along
axis as shown in Fig. 1) of the PRMS antennas are shown in
Fig. 9, where it can be seen that the simulated and measured re-
sults agree well. Again, because of mirror images of each other,
the simulated ARs with and 90 are identical but
with orthogonal polarizations as shown in Figs. 9(a) and (c), re-spectively. The simulated and measured ARs are less than 3 dB
from 3.33.7 GHz for both LHCP and RHCP, with an ARBW
of 400 MHz (or a fractional bandwidth of 11.4%). The ARBW
is narrower than the impedance bandwidth dB in
CP showed in Fig. 8(a) and (c), and thus becomes the operating
bandwidth of the antenna. With and 135 when the
antenna is operated in LP, Fig. 9(b) and (d) show that the simu-
lated ARs are larger than 40 dB from 3 to 4 GHz, indicating very
high linear polarization purity. Since the antenna measurement
equipment, the Satimo Starlab System, can measure the AR of
CP antennas only up to 20 dB, the measured result is therefore
a horizontal straight line at 20 dB in Fig. 9(b) and (d).
C. Ef ficiency and Realized Gain
The efficiencies of the PRMS antenna with different are
shown in Fig. 10. It can be seen that the simulated and measured
results agree well. The simulated and measured efficiencies of
the antenna at all rotation angles tested are above 80% across
the operating bandwidth (3.33.7 GHz).
The simulated and measured realized boresight gains of the
PRMS antenna are shown in Fig. 11. Again, good agreements
between simulated and measured results in the co-polarization
and cross-polarization can be seen.
Fig. 11(a) and (c) shows that the realized boresight gains
of the antenna in LHCP and RHCP in the operating band-
width of 3.33.7 GHz are above 5 dBi, with cross-polarization
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Fig. 8. Simulated and measured S11 for (a) , (b) , (c)
, and (d) .
of less than 10 dBi. This suggests high cross-polarization
isolation (XPI) of larger than 15 dB, i.e., good polarization
Fig. 9. Simulated and measured ARs for different rotation angles.
purity. Fig. 11(b) and (d) show that when the MS is rotated to
and 135 , (i.e., the PRMS antenna is LP), the realized
boresight gains are higher than 7.5 and 6 dBi, respectively,
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Fig. 10. Simulated and measured efficiencies for different rotation angles.
at the operating bandwidth for both simulated and measured
results. Since XPI for LP are less than 50 dB, it is too small
to be shown in the same figures
Fig. 11. Simulated and measured realized gain for different .
D. Radiation Pattern
The simulated and measured radiation patterns of the PRMS
antenna at 3.5 GHz for different are shown in Fig. 12. For
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Fig. 12. Simulated and measured radiation patterns at 3.5 GHz for different
. (a) -plane; (b) yz-plane; (c) -plane;(d) yz-plane; (e) -plane; (f) -plane; (g)
xz-plane; (h) -plane.
comparison, the radiation patterns of the slot antenna alone are
also shown in the same figures. It can be see that the antenna has
unidirectional radiation patterns, instead of bidirectional radia-
tion pattern typically for single slot antenna. This phenomenon
was already found in [24]. For the co-polarization patterns in
CP in Fig. 12(a), (b), (e), and (f), the front-to-back ratios (FBRs)
are all large than 12 dB. For the radiation patterns in LP shown
in Fig. 12(c), (d), (g) and (h), the FBRs are larger than 15 dB,
where the cross-polarizations are too small to be shown and so
omitted. The results in Fig. 12 show that the MS receives the LP
signal from the source slot antenna and reradiates the signal in
CP ( or 90 ) or LP ( or 135 ) to the other side
in the opposite direction (along axis as shown in Fig. 1).
V. CONCLUSIONS
A PRMS antenna designed using a slot antenna and a MS has
been presented. The polarization of the antenna can be mechan-
ically reconfigured to LHCP, RHCP and LP by rotating the MS
around the center with respect to the slot. The polarization-re-
configurable property has been analyzed and explained using an
equivalent circuit. The simulated and measured performances in
terms of polarization reconfigurability, efficiency, gain and radi-
ation pattern, have been presented. Results have shown that the
polarization reconfiguration can be achieved at around 3.5 GHz
with a fractional operating bandwidth 11.4%.
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H. L. Zhureceived the B.Eng. degree in information
engineering, and the Masters degree in electromag-
netic field and microwave engineering from Beijing
Institute of Technology, Beijing, China, in 2009
and 2011, respectively. He is currently working
toward the Ph.D. degree in electrical and electronic
engineering at the University of Hong Kong, Hong
Kong, China.
His research interests include antenna design and
study of metasurface.
S. W. Cheung (M89SM02) received the B.Sc.degree with First Class Honours in Electrical and
Electronic Engineering from Middlesex University,
London, U.K. in 1982 and the Ph.D. degree fromLoughborough University of Technology, Lough-
borough, U.K., in 1986.From 1982 to 1986, he was a Research Assistant
in the Department of Electronic and Electrical En-gineering, Loughborough University of Technology,
where he collaborated with the Rutherford AppletonLaboratory and many U.K. universities to work a
project fo r new generations of satellite systems. He is an Associate Professor at
the University of Hong Kong and in charge of the Microwave, RF Frequencyand Telecom Laboratories. His current research interests include antenna
designs, 2G, 3G, and 4G mobile communications systems, MIMO systems,and satellite communications systems.
Dr. Cheung has been serving the IEEE in Hong Kong for the past twentyyears. In 2009 and 2010, he was the Chairman of the IEEE Hong Kong Joint
Chapter on Circuits and Systems and Communications. He was the Honorary
Treasurer and currently the Chair-Elect of the IEEE Hong Kong.
X. H. Liu received B.Eng. degree in photoelectricinformation engineering from Shenzhen University,
Shenzhen, China, in 2008 and the M.Sc. degreein electrical and electronic engineering (communi-
cations engineering stream) from The Universityof Hong Kong, Hong Kong, China, in 2013. His
research interests include antenna design and the
study of metasurface
T. I. Yukreceived the B.S. degree from Iowa State
University, Ames, IA, USA, in 1978 and the M.S.and
Ph.D. degrees from Arizona State University, Tempe,
AZ, USA, in 1980 and 1986, respectively.
Since 1986, he has been teaching at the Universityof Hong Kong. His current research interests include:
wireless communications, and antenna designs.
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