IEEE TRANSACTIONS ON MAGNETICS, VOL. 25, NO. 5 , SEPTEMBER 1989 3985

UNITY POWER FACTOR HIGH CURRENT SLEW RATE MAGNET POWER SUPPLY

F.P. Dawson Department of Electrical Engineering

University of Toronto Toronto, Ontario

Canada MSS-lA4

Abstract This paper presents an alternative design approach for a unity

power factor high current slew rate magnet power supply. The pro- posed scheme utilizes a conventional AC to DC n pulse bridge and a DC side commutation network. The addition of a commutator addresses the following four concerns; converter device latching current problems are overcome; a current slew rate can be prepro- grammed within certain limitations; magnet current can be com- mutated safely in the event of a line voltage collapse; the main converter can be designed to operate at rated load with a unity power factor. The major focus of this paper is to develop a simple analytical model of the converter commutator and magnet system suitable for design purposes. The model is subsequently verified experimentally. Finally practical limitations of this circuit are described.

INTRODUCTION Applications such as plasma confinement require large mag-

netic field intensities. The plasma confinement configurations are typically toroidal in shape. Large field intensities are created by surrounding the torus with many turns of low conductivity wire. The coiled wire is operated at current densities which are limited by structural and heating concerns. The latter concern is addressed by pulsing the current. Consequently total losses in the system per unit operating time are minimized if one can provide an instan- taneous current turn on and turn off capability. This is not possible in practice, consequently a requirement on current slew rate is typ- ically imposed.

The power supplies used to provide a direct current are either 6 pulse, 12 pulse, or in some instances, 24 pulse AC to DC con- verters. These converters are also commonly, comprised of thyris- tor power switches, referred to as line commutated converters. One very important requirement of all thyristor switches is that a gating pulse should be applied until the latching current of the device has been exceeded. [ 11 This becomes increasingly difficult as the mag- net coil inductance gets higher, given a fixed ac bus voltage. In practice this problem is alleviated by seriesing converters in order to provide a satisfactory current slew rate. However once current is flowing it is no longer necessary to utilize the full voltage capabil- ity of the supply. Moreover, even if current regulation were to be attempted, with all bridges seriesed, we would operate at a steady state input line power factor significantly less than one. This power factor problem can be eliminated, if we provide a bypass for all but one of the converters. An illustration of this process is shown in Figure 1. [2] However even this approach is expensive in that there are many additional devices of which the bypass devices have to be rated for full dc rated conditions.

This paper presents an alternative power supply which over- comes the latching current problem, operates at unity power factor at rated load, and can be designed to operate for a range of current slew rates. Furthermore, the proposed power supply can also be used to remove the current from the magnet in the event of a line voltage collapse.

The presentation of this paper is divided into the following sections. Section 2 describes the operation of the proposed power

supply. Section 3 presents the equivalent circuit models for the power supply commencing with the load current initiation and concluding with the final steady state operating condition. More- over, the operation of the auxiliary assist turn off phase is described. Section 4 presents the experimental verification of the proposed equivalent circuit models. Finally, section 5 presents concluding remarks on the limitations of the proposed power supply.

60 Hr 39

L

Fig. 1: Traditional AC/DC converter for a magnet power supply.

2.POWER SUPPLY OPERATION Figure 2 shows a pictorial representation of the proposed

power supply. A brief description of the component functions fol- lows. The bridge denoted by S 11 is activated by a command sig- nal ON/OFF. The activation level is selected to be at a value of magnet current suitable to the user. A clearer interpretation of this meaning will follow shortly.

The initial current in the magnet is created by gating thyristor T11. Consequently capacitor C1 will discharge through the mag- net. The discharge current profile will be sinusoidal and can be characterized by a peak current and a half period. Peak current is predetermined by the size of C1 and the initial precharge voltage level applied across capacitor C1. On the other hand the half period is primarily dependent on the value of C1. Thus, the precharge voltage level and the value of capacitor C1 determine jointly the current slew rate.

The activation level for converter S 11 can be adjusted over a wide range. The lower bound on the activation level is associated with the minimum value of latching current for converter S11. Increased activation levels correspond to an increasing current slew rate. However this process is limited by the fact that the max- imum activation level is limited by the component values for L 1, C 1, and the precharge voltage on capacitor C 1.

The role of the commutation circuit can also be reversed if one wishes to commutate the load current independent of the line voltage. In this case the gating to S11 is removed and thrystor Ti1 is gated. Moreover it is necessary to provide a clamping circuit,

0018-9464/89/0900-3985$01.000 1989 IEEE

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across capacitor C 1, comprised of diode D 1 and voltage source VC+ . [3-41 Failure to do so will result in an excessive overvol- tage across capacitor C1 . Moreover the severity of the overvol- tage increases with increasing values of magnet inductance. The following section develops a simplified analytical model, suitable for design purposes.

ONIOFF

I__-, r - - sl*- - - I

I I

I I I I 1 I I I L-- --- _ - _ _ 1

Fig. 2: Power circuit configuration for unity power factor operation and high current skwrate.

3. ANALYSIS The following simplifying assumptions are made:

(a) The converter output terminals can be modelled as an effective DC voltage with a series inductance. This DC voltage is approximately equal to the open circuit out- put voltage of a six pulse converter. The total equivalent line inductance is L2. The load inductance is appreciably larger than the equivalent line inductance or stated mathematically we have: L 1 3 L 2 .

Under these aforementioned conditions the circuit models shown in Figure 3 are valid. The turn-on sequence is characterized by three modes. These modes are described as follows: Mode 1: Circuit model prior to the activation of ON/OFF on

bridge S 1 1 , and subsequent to the gating of T11. Mode2: Circuit model prior to the commutation of thyristor

Tl1.

Mode 3: Circuit model subsequent to the extinction of current in thyristor T11.

Relevant equations for each mode appear in Appendix A. Figure 4 shows selected currents and voltages which are valid for modes 1 through mode 3. Units for currents voltages and time are all normalized. Moreover specific normalization definitions appear in Appendix A. The following section presents the experimental verification of the aforementioned mathematical model.

(b)

4. EXPERIMENTAL VERIFICATION Various circuit parameters were chosen and varied systemati-

cally. In each case the experimental waveforms matched those of the computer simulation. A particular example is shown in Figure 5 and relates directly with the computer simulation result shown in Figure 4.

5. CONCLUSION This paper has presented an alternative design approach for a

unity power factor high current slew rate magnet power supply. Equations developed as part of the analysis can , through some manipulation, be utilized for design purposes. Specific details of the design process can be quite varied and are as such not part of the scope of this paper. Nevertheless salient limitations of a design employing the proposed power circuit are stated as follows:

L2 Model

T I 1

R1 'i I I I

Mode 2

I

Mode 3

Fig. 3: Equivalent circuit modes of operation.

(1) The maximum recharge voltage level for capacitor Cl is governed by the peak reverse voltage rating of the converter devices. The capacitor voltage subsequent to mode 1 should be lower than the equivalent applied voltage Veq. This condition is necessary in order that thyristor Tll will have sufficient turn off time margin.

(3) Capacitor value C1 and precharge voltage level are parameters which influence the current slew rate be it rate of rise, or rate of decline. The proposed power circuit can be utilized for the turn off process if the precharge voltage on capacitor C1 is somewhat larger than the peak line to line voltage.

(2 )

(4)

6. REFERENCES (1) Sarab K. Ghandi, Semiconductor Power Devices, New York:

Wiley Interscience, 1977. (2) Edward W. Kimbark, Direct Current Transmission; Vol. I,

New York: Wiley Interscience, 1971. (3) F.P. Dawson, S.B. Dewan, "Commutation Configurations for

the Protection of a Line Commutated Converter", Power Electronics Specialists Conference, 1988, pp. 245-253.

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0 0

Scales are in per unit value. "i :jj - M;i -- ;e2 I I

,I-b Mode3 ti > 0 I 1

II 1

I1

0

> . O O 0 . 4 0 0 . 8 p 1.20 1.60 2 . 0 0 0

4 I I r-

H U

0 0 I 1 0 I I I I I I I ,o. 00 0 . 4 0 0.80 1.20 1 . 6 0 2 . 0 0

I1 I1

0

9 . 0 0 0.40 1.60 2 . 0 0 0

-1

0 ' I 1 1 1 - 9 . 0 0 0 . 4 0 0.810

I 0

II r- -1 II

1.20 1 . 6 0 2 . 0 0

n I I I: I I I - ,o. 00 0 . 4 0 1 . 6 0 2 . 0 0 0 .- I I

0 0 I 0 ' I I I I

0 . 0 0 0 . 4 0 0 . 8 0 1.20 1.60 2 . 0 0 WT IN RAD

Fig. 4: Simulation waveforms for the circuit of Figure 2 and the modes of operation as depicted in Figure 3.

(4) F.P. Dawson, S.B. Dewan, "Analysis of Commutation Configurations for the Protection of a Line Commutated Con- verter", Conference Record Industry Application Society, IEEE IAS, 1988, pp. 987-995.

APPEND1X.A Base auantities

Horiz = 25 mec I div

Fig. 5 Experimental waveforms for; the circuit in Figure 2; the verification of the simulated waveforms in Figure 4.

Parameters L1 = .3H R I = 12.32R Vcl(0) =-100V

L2 = 1mH

c1= 18,000 p.F

Veq = 125V

I,, (w) = 10.15A Nomenclature

z = W t (P.U.)

Equations Mode 1: Initial Conditions [T11 on; S11 off; icl(0) = ' 0 vcl(o) = Vcl] Period of Validity: 0 < z I z l

dic 1 - = +iCl - vcl az

Period of Validity z1 < z 5 2 2

(3)

(4)