ÇUKUROVA UNIVERSITY INSTITUTE OF NATURAL AND …Endüksiyon Fırını, Pasif Filtreler, Melez Aktif...

ÇUKUROVA UNIVERSITY

INSTITUTE OF NATURAL AND APPLIED SCIENCES

MSc THESIS

Adnan TAN

MODELING AND ANALYSIS OF POWER QUALITY COMPENSATION

SYSTEMS FOR CURRENT SOURCE INVERTER BASED INDUCTION

FURNACE

DEPARTMENT OF ELECTRICAL AND ELECTRONICS ENGINEERING

ADANA, 2011





FURNACE

Adnan TAN

MSc THESIS


We certify that the thesis titled above was reviewed and approved for the award of degree

of the Master of Science by the board of jury on 09/08/2011.

Asst. Prof. Dr. K. Çağatay BAYINDIR

SUPERVISOR

Prof. Dr. Mehmet TÜMAY

MEMBER

Assoc. Prof. Dr. Zekeriya TÜFEKÇĠ

MEMBER

This MSc Thesis is written at the Department of Institute of Natural And Applied Sciences

of Çukurova University.

Registration Number:

Prof. Dr. İlhami YEĞİNGİL

Director

Institute of Natural and Applied Sciences

Note: The usage of the presented specific declarations, tables, figures, and photographs either in this thesis

or in any other reference without citation is subject to "The law of Arts and Intellectual Products"

number of 5846 of Turkish Republic.

To My Brother Levent TAN

Who Passed Away in October 2010

I

ABSTRACT

MSc THESIS



FURNACE

Adnan TAN




Supervisor : Asst. Prof. Dr. K. Çağatay BAYINDIR

Year : 2011, Pages: 193

Jury : Asst. Prof. Dr. K. Çağatay BAYINDIR

: Prof. Dr. Mehmet TÜMAY

: Assoc. Prof. Dr. Zekeriya TÜFEKÇĠ

With the developments in the power electronics technology, high-power converters

have become available in many of industrial applications. High-power converters provide

efficient electric energy utilizing and advance control of process in many of industrial

applications. Besides the advantages of these power converters, they show nonlinear load

characteristic which causes various power quality (PQ) problems in electric power systems.

Induction steel scrap melting furnaces are one of the examples of industrial

applications which use high power converters in power supplies. In high power induction

furnaces, power supply is commonly formed from 12-pulse or higher pulse fully controlled

rectifier, current source inverter and parallel resonant tank circuit. Besides the advantages of

current source inverter based induction furnaces (CSI-IF) in melting process, CSI-IFs causes

serious PQ problems such as; time varying harmonics and interharmonics. Moreover, when

these high power furnaces are supplied from weak power systems, voltage fluctuations and

flicker problems occur in the power system.

In this thesis, the CSI-IF is modeled in PSCAD/EMTDC simulation program in

order to investigate the time varying harmonics and interharmonics of CSI-IF and find

solutions to these PQ problems. For the solution of PQ problems of CSI-IF, passive filtering

and active filtering methods are investigated. In passive filtering methods; single tuned filter,

C-type filter and broad band filter are designed for the PQ problems of CSI-IF. In active

filtering methods, hybrid active power filters (HAPF) which use shunt active power filter

(APF) - shunt passive filter topology and APF in series with shunt passive filters topology

are designed for the PQ problems of CSI-IF. The compensation performances of these

passive filters and HAPFs are investigated by modeling in PSCAD/EMTDC simulation

program.

Key Words: Harmonics, Interharmonics, Current Source Inverters, Induction Furnace,

Passive Filters, Hybrid Active Power Filters

II

ÖZ

YÜKSEK LİSANS TEZİ

AKIM KAYNAKLI ÇEVİRGECE DAYALI ENDÜKSİYON FIRINI İÇİN

GÜÇ KALİTESİ KOMPANZASYON SİSTEMLERİNİN MODELLENMESİ

VE ANALİZİ

Adnan TAN

ÇUKUROVA ÜNİVERSİTESİ

FEN BİLİMLERİ ENSTİTÜSÜ

ELEKTRİK ELEKTRONİK MÜHENDİSLİĞİ ANABİLİM DALI

Danışman : Yrd. Doç. Dr. K. Çağatay BAYINDIR

Yıl :2011, Pages 193

Jüri : Yrd. Doç. Dr. K. Çağatay BAYINDIR

: Prof. Dr. Mehmet TÜMAY

: Doç. Dr. Zekeriya TÜFEKÇĠ

Güç elektroniği teknolojisindeki gelişmelerle birlikte yüksek güçlü çevirgeçler

birçok endüstriyel uygulamada kullanılmaktadır. Bu yüksek güçlü çevirgeçler, birçok

endüstriyel uygulamada elektrik enerjisinin verimli kullanılmasını ve önemli proses kontrol

özellikleri sağlamaktadırlar. Çevirgeçlerin bu avantajlarının yanında, bu çevirgeçler elektrik

güç sistemlerinde çeşitli güç kalitesi problemlerine yol açan lineer olmayan yük

karakteristiği göstermektedirler.

Endüksiyonla hurda çelik eritme fırınları yüksek güçlü çevirgeçlerin kullanıldığı

endüstriyel uygulamalara bir örnektir. Yüksek güçlü endüksiyon fırınlarında kullanılan güç

ünitesi genellikle 12 veya daha yüksek darbeli tam kontrollü doğrultucudan, akım kaynaklı

çevirgeçten ve paralel bağlı rezonans tank devresinden oluşmaktadır. Akım kaynaklı

çevirgece dayalı endüksiyon fırının (AKÇ-EF) avantajlarının yanında bu fırınlar zamanla

değişen harmonik ve araharmonik gibi güç kalitesi sorunlarına yol açmaktadır. Ayrıca bu

yüksek güçlü fırınlar zayıf elektrik şebekelerine bağlandıklarında, elektrik şebekesinde

gerilim dalgalanmalarına ve kırpışma sorunlarına sebep olmaktadırlar.

Bu tez çalışmasında, AKÇ-EF’ nin zamanla değişen harmoniklerini ve

araharmoniklerini incelemek ve bu güç kalitesi problemlerine çözüm bulmak için fırının

PSCAD/EMTDC simülasyon programında modeli oluşturulmuştur. Fırının güç kalitesi

problemlerinin çözümü için pasif ve aktif filtreleme yöntemleri incelenmiştir. Pasif

filtreleme yöntemlerinde, AKÇ-EF’ nin güç kalitesi problemlerinin çözümü için tek frekansa

ayarlanmış pasif filtre, C-Tipi filtre ve geniş band geçiren filtre tasarlanmıştır. Aktif

filtreleme yöntemlerinde ise AKÇ-EF’ nin güç kalitesi problemlerinin çözümü için paralel

aktif güç filtresi (AGF)-paralel pasif filtre topolojisi ve paralel pasif filtre ile seri bağlanmış

AGF topolojisi kullanan melez AGF’ leri tasarlanmıştır. Bu pasif filtrelerin ve melez

AGF’ lerinin kompanzasyon performansları PSCAD/EMTDC simülasyon programında

modellenerek incelenmiştir.

Anahtar Kelimeler: Harmonikler, Araharmonikler, Akım Kaynaklı Çevirgeçler,

Endüksiyon Fırını, Pasif Filtreler, Melez Aktif Güç Filtreleri

III

ACKNOWLEDGEMENTS

I would like to express my deepest gratitude to my supervisor Asst. Prof. Dr.

K. Çağatay BAYINDIR for his guidance, advice, criticism, encouragements and

insight throughout this research.

I am also grateful to Prof. Dr. Mehmet TÜMAY, head of the Department, for

his help and support during my study.

I am also grateful to Dr. Ahmet TEKE, Dr. M. Uğraş CUMA, Lütfü

SARIBULUT, Murat FURAT, Tahsin KÖROĞLU and M. Mustafa SAVRUN for

their support and friendship.

I wish to express my special thanks to Dr. Alper TERCĠYANLI and Ġlker

YILMAZ for sharing their knowledge and experience on power quality analysis and

active power filters.

I also wish to express my deepest gratitude to my mother Dilber TAN, my

brother Mehmet Serkan TAN and my extended family for their endless support,

encouragement and patience.

Finally, I owe my deepest gratitude to my lovely wife for her love, persistent

confidence in me, patience and endless support during this heavy work.

IV

CONTENTS PAGE

ABSTRACT .................................................................................................................. I

ÖZ ................................................................................................................................ II

ACKNOWLEDGEMENTS ....................................................................................... III

CONTENTS ............................................................................................................... IV

LIST OF TABLES ............................................................................................................. VI

LIST OF FIGURES ........................................................................................................ VIII

LIST OF SYMBOLS .............................................................................................. XVI

LIST OF ABBREVATIONS ............................................................................... XXIV

1. INTRODUCTION .................................................................................................... 1

1.1. Background and Research Motivation .............................................................. 1

1.2. Scope of Thesis .................................................................................................. 2

1.3. Outline of Thesis ............................................................................................... 6

2. INDUCTION HEATING ......................................................................................... 7

2.1. History of Induction Heating ............................................................................. 7

2.2. Basic Principles of Induction Heating ............................................................... 8

2.3. Applications of Induction Heating .................................................................. 10

2.3.1. Heating ................................................................................................... 10

2.3.2. Heat Treating .......................................................................................... 11

2.3.3. Melting ................................................................................................... 14

2.3.4. Welding, Brazing and Soldering ............................................................ 18

2.3.5. Other Applications ................................................................................. 19

2.4. Power Supplies of Induction Furnaces ............................................................ 20

3. HARMONICS AND INTERHARMONICS ......................................................... 29

3.1. Harmonics ........................................................................................................ 29

3.2. Interharmonics ................................................................................................. 37

3.3. Harmonic and Interharmonic Mitigation Techniques ..................................... 39

3.3.1. Passive Filters ......................................................................................... 40

3.3.2. Active Power Filters ............................................................................... 44

3.4. Harmonic and Interharmonic Extraction Methods .......................................... 51

V

4. MODELING AND ANALYSIS OF CSI-IF .......................................................... 53

4.1. Operating Principles of CSI – IF Power Supply ............................................. 54

4.1.1. Parallel Resonant Tank Circuit ............................................................... 54

4.1.2. CSI .......................................................................................................... 63

4.1.3. 12-Pulse Fully Controlled Rectifier ....................................................... 66

4.2. Power Quality Problems of CSI-IF ................................................................. 68

4.3. Modeling of Current Source Induction Furnace .............................................. 71

4.3.1. Power Circuit Parameters ....................................................................... 71

4.3.2. Controller of CSI-IF ............................................................................... 72

4.3.3. Simulation Results of CSI-IF ................................................................. 75

4.3.4. Comparison of Simulation Results with Real Power Quality

Measurements of Induction Furnace ...................................................... 92

5. MODELING AND ANALYSIS OF PQ COMPENSATION SYSTEMS FOR CSI-

IF ............................................................................................................................ 95

5.1. Passive Filters .................................................................................................. 95

5.1.1. Single Tuned Passive Filters .................................................................. 95

5.1.2. C - Type Passive Filters ........................................................................ 107

5.1.3. Broad - Band Passive Filters ................................................................ 119

5.2. Hybrid Active Power Filters .......................................................................... 134

5.2.1. Shunt Active Power Filter and Parallel Passive Filter Combination .... 134

5.2.1.1. Power Circuit Configuration of Proposed HAPF System ........ 134

5.2.1.2. Control Method of APF Modules in Proposed HAPF System 137

5.2.1.3. Simulation Results of Proposed HAPF System ....................... 143

5.2.2. Active Power Filter Series with Passive Filter Combination: SHAPF 157

5.2.2.1. Power Circuit Configuration of Proposed SHAPF System ..... 157

5.2.2.2. Control Method of SHAPF Modules in Proposed SHAPF System

.................................................................................................. 162

5.2.2.3. Simulation Results of Proposed SHAPF System ..................... 165

6. CONCLUSIONS .................................................................................................. 179

REFERENCES ......................................................................................................... 185

BIOGRAPHY .......................................................................................................... 193

VI

LIST OF TABLES PAGE

Table 3.1. Harmonic Extraction Methods ........................................................... 51

Table 4.1. Power Circuit Parameters of CSI-IF Model ....................................... 72

Table 5.1. Power Circuit Parameters of Single Tuned Passive Filter Model ...... 99

Table 5.2. Power Circuit Parameters of C-Type Filter Model .......................... 109

Table 5.3. Initial Values for Power Circuit Parameters of LLCL Type Broad-

Band Filter Model............................................................................. 122

Table 5.4. Final Values for Power Circuit Parameters of LLCL Type Broad-Band

Filter Model ...................................................................................... 123

Table 5.5. Power Circuit Parameters of HAPF System ..................................... 145

Table 5.6. Power Circuit Parameters of SHAPF System .................................. 167

Table 6.1. Comparison of Investigated Compensation Systems ....................... 183

VIII

LIST OF FIGURES PAGE

Figure 1.1. General Block Diagram of Induction Furnaces .................................... 1

Figure 1.2. 3-D Drawing of a High Power Induction Melting Furnace .................. 4

Figure 2.1. Principles of Induction Heating ............................................................ 9

Figure 2.2. Steel Billet Heating Induction Furnace .............................................. 11

Figure 2.3. Variety of Induction Hardening Patterns Obtained Using Variations in

Frequency, Heat Time, and Coil Power ............................................. 12

Figure 2.4. Induction Spin Hardening Process ..................................................... 13

Figure 2.5. Gap by Gap Hardening Operation of Gears ....................................... 13

Figure 2.6. Cross - Section View of Channel Type Induction Furnace ................ 16

Figure 2.7. Cross - Section View of Coreless Induction Furnace ......................... 17

Figure 2.8. Induction Welding of Tubular Products ............................................. 18

Figure 2.9. Power - Frequency Diagram of Induction Heating Applications ....... 22

Figure 2.10. Three Phase Uncontrolled Rectifier ................................................... 22

Figure 2.11. Three Phase Fully Controlled Rectifier .............................................. 23

Figure 2.12. Three Phase Uncontrolled Rectifier with SMR .................................. 23

Figure 2.13. Power-Frequency Diagram of Solid State Devices Used in Inverters of

Induction Heating Power Supplies ..................................................... 25

Figure 2.14. VSI with Series Resonance Load ....................................................... 26

Figure 2.15. VSI with Series Resonant Circuit Connected to Parallel Resonant

Tank .................................................................................................... 27

Figure 2.16. VSI with Parallel Resonance Load ..................................................... 28

Figure 3.1. Single Line Diagram of Power System with Potential Parallel

Resonance Problems ........................................................................... 33

Figure 3.2. Equivalent Circuit of Power System with Potential Parallel Resonance

Problems ............................................................................................. 33

Figure 3.3. Power System Impedance in Parallel Resonance Problems ............... 33

Figure 3.4. Single Line Diagram of Power System with Potential Parallel and

Series Resonance Problem ................................................................. 34

IX

Figure 3.5. Equivalent Circuit of Power System with Potential Parallel and Series

Resonance Problems ........................................................................... 35

Figure 3.6. Power System Impedance in Parallel and Series Resonance Problems .

............................................................................................................ 35

Figure 3.7. Single Line Diagram of Power System with Linear, Nonlinear and

Compensation Loads .......................................................................... 36

Figure 3.8. Equivalent Circuit of Power System with Linear, Nonlinear and

Compensation Loads .......................................................................... 36

Figure 3.9. Effects of Resistive Loads on Parallel Resonance ............................. 36

Figure 3.10. Series Passive Filter ............................................................................ 41

Figure 3.11. Shunt Passive Filter Topologies ......................................................... 42

Figure 3.12. Shunt Passive filters Impedance - Frequency Curves ......................... 43

Figure 3.13. Broad-Band Passive Filter .................................................................. 43

Figure 3.14. Block Diagram of Power Circuit of APF ........................................... 45

Figure 3.15. Block Diagram of Shunt APF ............................................................. 46

Figure 3.16. Block Diagram of Series APF ............................................................ 47

Figure 3.17. Block Diagram of UPQC.................................................................... 48

Figure 3.18. Hybrid APF Topologies ..................................................................... 49

Figure 3.19. Converter Based on Classification of APFs ....................................... 50

Figure 4.1. Power Circuit Topology of the Coreless CSI-IF in The Steel Mill .... 53

Figure 4.2. Equivalent Circuit of Transformer ..................................................... 55

Figure 4.3. Equivalent Circuit of Coreless Induction Furnace ............................. 55

Figure 4.4. Coreless Induction Furnace Equivalent Circuit Referred to Primary

Side ..................................................................................................... 56

Figure 4.5. Ideal Parallel Resonant Circuit ........................................................... 56

Figure 4.6. Impedance-Frequency Curve of Ideal Parallel Resonant Circuit ....... 57

Figure 4.7. Practical Parallel Resonant Circuit ..................................................... 58

Figure 4.8. Practical Parallel Resonant Circuit Converted To Ideal Parallel

Resonant Circuit Form ....................................................................... 58

Figure 4.9. CSI of IF with Parallel Resonant Tank Circuit .................................. 64

X

Figure 4.10. Voltage and Current Waveforms of Parallel Resonant Tank Circuit

Operating Above the Resonance Frequency ...................................... 64

Figure 4.11. Operation of Load Commutated CSI .................................................. 65

Figure 4.12. Load Commutation of CSI ................................................................. 65

Figure 4.13. Circuit Diagram of 12-Pulse Rectifier ................................................ 67

Figure 4.14. Generated Harmonics Related with Operating Frequency of CSI-IF 70

Figure 4.15. Power Circuit Model of CSI-IF .......................................................... 71

Figure 4.16. Block Diagram of DC Link Controller of CSI-IF .............................. 73

Figure 4.17. Block Diagram of CSI of CSI-IF........................................................ 74

Figure 4.18. Frequency Trend of Modeled Induction Furnace According To

Operating Scenario ............................................................................. 76

Figure 4.19. Trend of Phase Difference between Voltage and Current of Resonant

Load .................................................................................................... 77

Figure 4.20. Trend of CSI Angular Operating Frequency ...................................... 77


Load during Entire Simulation ........................................................... 78

Figure 4.22. Trend of CSI Angular Operating Frequency during Entire Simulation .

............................................................................................................ 78

Figure 4.23. Phase of the Operating Frequency of CSI .......................................... 79

Figure 4.24. Thyristor Gate Pulses of CSI .............................................................. 79

Figure 4.25. Current and Voltage Waveform between Terminals of Parallel

Resonant Tank Circuit ........................................................................ 80

Figure 4.26. Current Waveform of Furnace Coil .................................................... 80

Figure 4.27. Trend of Firing Angle of 12-Pulse Rectifier ...................................... 81

Figure 4.28. Trend of DC Link Voltage of CSI-IF ................................................. 82

Figure 4.29. Trend of DC Link Current of CSI-IF ................................................. 82

Figure 4.30. Trend of DC Link Voltage of CSI-IF during Entire Simulation ........ 83

Figure 4.31. Trend of DC Link Current of CSI-IF during Entire Simulation ......... 83

Figure 4.32. Ripple on DC Link Current of CSI-IF ................................................ 84

Figure 4.33. Supply Current Waveform of CSI-IF ................................................. 84

Figure 4.34. Supply Voltage Waveform of CSI-IF................................................. 85

XI

Figure 4.35. Trend of Active and Reactive Power of CSI-IF ................................. 85

Figure 4.36. Angular Operating Frequency at the 1st Second of the Simulation .... 86

Figure 4.37. Generated Harmonics When CSI-IF Operates at 175 Hz ................... 87

Figure 4.38. Harmonic Spectrum of CSI-IF Current when CSI-IF operates at 175

Hz ....................................................................................................... 88

Figure 4.39. Angular Operating Frequency at the 5th

Second of the Simulation .... 89

Figure 4.40. Generated Harmonics When CSI-IF Operates at 235 Hz ................... 90

Figure 4.41. Harmonic Spectrum of CSI-IF Current when CSI-IF operates at 235

Hz ....................................................................................................... 91

Figure 4.42. Harmonic Spectrum Obtained by the PQ Measurements in the Steel

Mill ..................................................................................................... 93

Figure 4.43. Interharmonic Spectrum Obtained by the PQ Measurements in the

Steel Mill ............................................................................................ 93

Figure 5.1. Single Tuned Passive Filter ................................................................ 96

Figure 5.2. Impedance Frequency Curve of Single Tuned Filter ......................... 96

Figure 5.3. Impedance-Frequency Curve of 5th

, 7th

, 11th

and 13th

Harmonic Single

Tuned Filters ....................................................................................... 98

Figure 5.4. Single Line Diagram and Equivalent Circuit of CSI-IF with Single

Tuned Passive Filter ........................................................................... 99

Figure 5.5. Impedance-Frequency Curve of Single Tuned Filter Designed for CSI-

IF....................................................................................................... 100

Figure 5.6. Power Circuit Model of Single Tuned Filter .................................... 101

Figure 5.7. CSI-IF Current Waveform in Single Tuned Filter Simulation ......... 102

Figure 5.8. Source Current Waveform in Single Tuned Filter Simulation ......... 102

Figure 5.9. Single Tuned Filter Current Waveform in Single Tuned Filter

Simulation......................................................................................... 103

Figure 5.10. Reactive Powers Drawn From source, Drawn By CSI-IF and Drawn

by Single Tuned Filter in Single Tuned Filter Simulation ............... 103

Figure 5.11. Harmonic Spectrum of Lower Order Harmonics of CSI-IF Current

and Source Current in Single Tuned Filter Simulation .................... 105

XII

Figure 5.12. Harmonic Spectrum of Higher Order Harmonics of CSI-IF Current

and Source Current in Single Tuned Filter Simulation .................... 106

Figure 5.13. Power Circuit Topology of C-Type Filter ........................................ 107

Figure 5.14. Single Line Diagram and Equivalent Circuit of CSI-IF with C-Type

Filter ................................................................................................. 109

Figure 5.15. Impedance-Frequency Curve of C-Type Filter Designed for CSI-IF ....

.......................................................................................................... 110

Figure 5.16. Impedance-Frequency Curve of C-Type Filter with Different Quality

Factor ................................................................................................ 111

Figure 5.17. Impedance-Frequency Curve of C-Type Filter with Different Reactive

Power Ratings ................................................................................... 112

Figure 5.18. Power Circuit Model of C-Type Filter ............................................. 113

Figure 5.19. CSI-IF Current Waveform in C-Type Filter Simulation .................. 114

Figure 5.20. Source Current Waveform in C-Type Filter Simulation .................. 114

Figure 5.21. C-Type Filter Current Waveform in C-Type Filter Simulation ....... 115


by C-Type Filter in C-Type Filter Simulation .................................. 115


and Source Current in C-Type Filter Simulation ............................. 117


and Source Current in C-Type Filter Simulation ............................. 118

Figure 5.25. Broad Band Passive Filter Topologies ............................................. 119

Figure 5.26. Single Line Diagram and Equivalent Circuit of CSI-IF with LLCL

Type Broad Band Filter .................................................................... 123

Figure 5.27. Impedance-Frequency Curve of LLCL Type Broad-Band Filter

Designed for CSI-IF ......................................................................... 124

Figure 5.28. Filtering Characteristic of LLCL Type Broad-Band Filter Designed

for CSI-IF ......................................................................................... 125

Figure 5.29. Power Circuit Model of LLCL Type Broad-Band Filter.................. 125

Figure 5.30. CSI-IF Current Waveform in LLCL-Type BBF Filter Simulation .. 126

Figure 5.31. Source Current Waveform in LLCL-Type BBF Filter Simulation .. 127

XIII

Figure 5.32. Current Waveform of Shunt Branch of LLCL Type Filter in LLCL-

Type BBF Filter Simulation ............................................................. 127

Figure 5.33. Source Voltage Waveform in LLCL-Type BBF Filter Simulation .. 128

Figure 5.34. CSI-IF Voltage Waveform in LLCL-Type BBF Filter Simulation .. 128

Figure 5.35. Reactive Powers Drawn From source and Drawn By CSI-IF in LLCL

Type Filter Simulation ...................................................................... 129


and Source Current in LLCL Type Broad-Band Passive Filter ....... 131


and Source Current in LLCL Type Broad-Band Passive Filter

Simulation......................................................................................... 132

Figure 5.38. Harmonic Spectrum of CSI-IF Voltage LLCL Type Broad-Band

Passive Filter Simulation .................................................................. 133

Figure 5.39. Proposed HAPF System for CSI-IF ................................................. 135

Figure 5.40. Power Circuit Configuration of APF Modules ................................. 136

Figure 5.41. Control Method of Shunt APF Modules in Proposed HAPF System137

Figure 5.42. Harmonic Extraction Method in Shunt APF Modules ..................... 138

Figure 5.43. Structure of EPLL............................................................................. 139

Figure 5.44. Distorted Input signal, Extracted Fundamnetal Signal and Extracted

Harmonics of EPLL .......................................................................... 140

Figure 5.45. DC Link Voltage Control of Shunt APF Modules ........................... 142

Figure 5.46. Current Control of Shunt APF Modules ........................................... 143

Figure 5.47. Power Circuit Model of Proposed HAPF System ............................ 144

Figure 5.48. CSI-IF Current Waveform in Proposed HAPF Simulation .............. 146

Figure 5.49. CSI-IF and STF Current Waveform in Proposed HAPF Simulation .....

.......................................................................................................... 146

Figure 5.50. Source Current Waveform in Proposed HAPF Simulation .............. 147

Figure 5.51. STF Currents in Proposed HAPF Simulation ................................... 147

Figure 5.52. High Voltage Side APFs Currents in Proposed HAPF Simulation .. 148


and Source Current in Proposed HAPF Simulation ......................... 149

XIV


and Source Current in Proposed HAPF Simulation ......................... 150

Figure 5.55. CSI-IF Voltage Waveform in Proposed HAPF Simulation ............. 151


By Single Tuned Filter in HAPF Simulation ................................... 152

Figure 5.57. Injected Current Waveform of APF1 in Proposed HAPF Simulation ...

.......................................................................................................... 153

Figure 5.58. Injected Current Waveform of APF2 in Proposed HAPF Simulation ...

.......................................................................................................... 154

Figure 5.59. DC Link Voltage Waveform of APF1 in Proposed HAPF Simulation ..

.......................................................................................................... 155

Figure 5.60. DC Link Voltage Waveform of APF2 in Proposed HAPF Simulation ..

.......................................................................................................... 155

Figure 5.61. Reference Current and Injected Current Waveforms of APF1 in

Proposed HAPF Simulation ............................................................. 156


Proposed HAPF Simulation ............................................................. 156

Figure 5.63. Proposed SHAPF System for CSI-IF ............................................... 158

Figure 5.64. Single Line Equivalent Circuit of SHAPF ....................................... 159

Figure 5.65. Power Circuit Configuration of SHAPF Modules............................ 161

Figure 5.66. Filtering Characteristics of a Single SHAPF Module with Different

SHAPFK Values .................................................................................. 162

Figure 5.67. Filtering Characteristics of Proposed SHAPF System with Different

SHAPFK Values ................................................................................... 162

Figure 5.68. Control Method of Shunt APF Modules in Proposed SHAPF System ..

.......................................................................................................... 163

Figure 5.69. DC Link Voltage Control of SHAPF Modules ................................ 164

Figure 5.70. Voltage Control of SHAPF Modules ............................................... 165

Figure 5.71. Power Circuit Model of Proposed SHAPF System .......................... 166

Figure 5.72. CSI-IF Current Waveform in Proposed SHAPF Simulation ............ 168

Figure 5.73. Source Current Waveform in Proposed SHAPF Simulation ............ 168

XV

Figure 5.74. High Voltage Side APFs Currents in Proposed SHAPF Simulation ......

.......................................................................................................... 169


and Source Current in Proposed SHAPF Simulation ....................... 171


and Source Current in Proposed SHAPF Simulation ....................... 172

Figure 5.77. CSI-IF Voltage Waveform in Proposed SHAPF Simulation ........... 173

Figure 5.78. Reactive Powers Drawn From Source, Drawn By CSI-IF and Drawn

By SHAPF Modules in SHAPF Simulation ..................................... 174

Figure 5.79. Injected Current Waveform of SAPF1 in Proposed SHAPF Simulation

.......................................................................................................... 175


.......................................................................................................... 175

Figure 5.81. DC Link Voltage Waveform of SHAPF1 in Proposed SHAPF

Simulation......................................................................................... 176


Simulation......................................................................................... 176

Figure 5.83. Reference Voltages and Triangular Wave Waveforms of SHAPF1

Controller in Proposed SHAPF Simulation ...................................... 177


Controller in Proposed SHAPF Simulation ...................................... 177

XVI

LIST OF SYMBOLS

fA : Amplitude of fundamental component

C : Capacitance

BBFC : Capacitance of LC type broad band passive filter capacitor

,BBF Y sC : Capacitance of wye connected capacitors of LLCL type broad

band passive filter

,BBF sC : Capacitance of delta connected capacitors of LLCL type broad

band passive filter

,1CtypeC : Capacitance of auxiliary capacitor of C-type passive filter

,2CtypeC : Capacitance of main capacitor of C-type passive filter

DCC : DC link capacitance

,dc APFC : DC link capacitance of active power filter

,dc SHAPFC : DC link capacitance of shunt hybrid active power filter

fC : Induction furnace resonant tank circuit capacitance

PC : Capacitance of parallel resonant load

SHAPFC : Capacitance of series connected capacitor of shunt hybrid active

power filter

,STF YC : Capacitance of wye connected capacitors of single tuned passive

filter

,STFC : Capacitance of delta connected capacitors of single tuned passive

filter

f : Frequency

if : Generated harmonics and interharmonics frequency

pf : Parallel resonance frequency

,p BBFf : Parallel resonance tuning frequency of LLCL type broad band

passive filter

XVII

PFf : Tuning frequency of passive filter

rf : Resonance frequency

sf : Series resonance frequency

,s BBFf : Series resonance tuning frequency of LLCL type broad band

passive filter

STFf : Tuning frequency of single tuned passive filter

Ctypef : Tuning frequency of C-type passive filter

mf : Maximum frequency of practical parallel resonant load

of : Operating frequency of the inverter

uf : Unity power factor frequency of practical parallel resonant load

1f : Fundamental supply frequency

h : Order of harmonics

I : Current

( , , )a b cI : Phase a, b, c currents

( , , ),a b c APFI : Phase a, b, c currents pf active power filter

( , , )_ ,a b c capref APFI : Generated three phase current references of DC link capacitor of

active power filter

( , , ),a b c CSI IFI : Phase a, b, c currents of CSI-IF

( , , ),a b c CtypeI : Phase a, b, c currents of C-type passive filter

( , , )_ ,a b c ref APFI : Generated three phase current references of active power filter

( , , ),a b c sourceI : Phase a, b, c currents of source

( , , ),a b c hI : Phase a, b, c harmonic currents

( , , ),a b c STFI : Phase a, b, c currents of single tuned passive filter

APFI : Injected current of active power filter

PCI : Current flow from the capacitance of parallel resonant load

XVIII

,capref APFI : Generated current reference of DC link capacitor of active power

filter

CSI IFI : CSI-IF current

,CSI IF harI : Harmonic components of CSI-IF current

DCI : DC link current

,DC setI : Set value of DC link current

fI : Fundamental component of current

fur coilI : Induction furnace coil current

hI : Harmonic component of current

,h pI : Magnified current harmonic components caused by parallel

resonance

,h sI : Voltage harmonics caused by series resonance

iI : Input current signal of EPLL

loadI : Load current

,load hI : Harmonic components of load current

PLI : Current flow from the inductance of parallel resonant load

PFI : Current drawn by passive filter

PFI : Harmonic components of current drawn by passive filter

,ref APFI : Reference current of active power filter

res tankI : Resonant tank circuit current

SHAPFI : Current of shunt hybrid active power filter

sourceI : Source current

,source hI : Harmonic components of source current

AK : Peak detector gain of EPLL

APFK : Feedback gain of active power filter

IK : Low-pass filter integrator gain of EPLL

XIX

PK : Low-pass filter proportional gain of EPLL

SHAPFK : Feedback gain of shunt hybrid active power filter

L : Inductance

APFL : Inductance of smoothing inductor of active power filter

SHAPFL : Inductance of series connected inductor of shunt hybrid active

power filter

BBFL : Inductance of LC type broad band passive filter

,BBF sL : Inductance of shunt branch inductor of LLCL type broad band

passive filter

,1CtypeL : Inductance of C-type passive filter

DCL : DC link inductance

,i BBFL : Inductance of input inductor of LLCL type broad band passive

filter

LL : Inductance of practical parallel resonant load

PL : Inductance of parallel resonant load

,o BBFL : Inductance of output inductor of broad band passive filter

STFL : Inductance of single tuned passive filter

strayL : Stray Inductance

sysL : Power system inductance

coilN : Turn number of induction furnace coil

PN : Primary winding turn number of transformer

SN : Secondary winding turn number of transformer

1p

: Pulse number of the rectifier

2p

: Pulse number of the inverter

Q : Reactive power

BBFQ : Reactive power rating of broad band passive filter

XX

CtypeQ : Reactive power rating of single tuned passive filter

PFQ : Reactive power rating of passive filter

STFQ : Reactive power rating of single tuned passive filter

BBFQF : Quality factor of broad band passive filter

CtypeQF : Quality factor of C-type passive filter

PQF : Quality factor of parallel resonant load

PFQF : Quality factor of passive filter

STFQF : Quality factor of single tuned passive filter

R : Resistance

,BBF sR : Resistance of shunt branch resistor of LLCL type broad band

filter

CR : Core resistance of transformer

chR : Resistance of charge metal of induction furnace

,ch refR : Resistance of charge metal of induction furnace referred to

primary side

CtypeR : Resistance of C-type filter

fR : Equivalent induction furnace resistance

fcR : Resistance of induction furnace coil

LR : Resistance of practical parallel resonant load

LoadR : Load resistance

PR : Resistance of parallel resonant load

PrR : Resistance of primary winding of transformer

SR : Resistance of secondary winding of transformer

STFR : Resistance of single tuned filter

LoadS : Apparent Power Rating of Load

SCS : Short circuit power of power system

XXI

( , , )a b cV : Phase a, b, c voltages

( , , )_ ,a b c capref SHAPFV : Generated three phase voltage references of DC link capacitor of

shunt hybrid active power filter

( , , )_ ,a b c ref SHAPFV : Generated three phase voltage references of shunt hybrid active

power filter

( , , ),a b c sourceV : Phase a, b, c source voltages

APFV : Injected voltage of active power filter

,cap APFV : DC link capacitor voltage of active power filter

,cap SHAPFV : DC link capacitor voltage of shunt hybrid active power filter

,capref SHAPFV : Generated voltage reference of DC link capacitor of shunt hybrid

active power filter

,capset APFV : DC link capacitor voltage set value of active power filter

,capset SHAPFV : DC link capacitor voltage set value of shunt hybrid active power

filter

,cappi APFV : PI controller output signal of DC link controller of active power

filter

,cappi SHAPFV : PI controller output signal of DC link controller of shunt hybrid

active power filter

DCV : DC link voltage

fV : Fundamental component of voltage

hV : Harmonic component of voltage

,h pV : Magnified voltage harmonic components caused by parallel

resonance

,h sV : Voltage harmonics caused by series resonance

,L L rmsV : Line to line RMS value of power system voltage

mV : Peak value of supply voltage rectifier

XXII

meanV : Mean value of DC link voltage of rectifier

PV : Voltage across the terminals of parallel resonant load

res tankV : Voltage between terminals of resonant tank circuit

,ref SHAPFV : Generated reference voltage of shunt hybrid active power filter

%SCV : Short circuit voltage of transformer in percentage

SHAPFV : Shunt hybrid active power filter voltage

sourceV : Source voltage

CX : Reactance of capacitor

chX : Reactance of charge metal of induction furnace

,ch refX : Reactance of charge metal of induction furnace referred to

primary side

fX : Equivalent induction furnace reactance

fcX : Reactance of induction furnace coil

gapX : Reactance of insulation material and gap between induction

furnace coil and inside of induction furnace crucible

,gap refX : Referred reactance of insulation material and gap between

induction furnace coil and inside of induction furnace crucible

LLX : Reactance of practical parallel resonant load inductance

PLX : Reactance of parallel resonant load inductance

PCX : Reactance of parallel resonant load capacitance

LoadX : Load reactance

MX : Magnetizing reactance of transformer

PrX : Leakage reactance of primary winding of transformer

SX : Leakage reactance of secondary winding of transformer

sourceX : Reactance of source

TRX : Total leakage reactance of transformer

XXIII

PY : Admittance of parallel resonant load

Z : Impedance

EqZ : Equivalent impedance

Filter SystemZ : Equivalent impedance of passive filter and system impendance

PZ : Impedance of parallel resonant load

PmZ : Maximum impedance of parallel resonant load

PFZ : Impedance of passive filter

sysZ : Impedance of power system

: firing angle of rectifier

: Phase angle between voltage and current of parallel resonant tank

circuit of induction furnace

error : Phase angle error between voltage and current of parallel resonant

tank circuit of induction furnace

set : Phase angle set value between voltage and current of parallel

resonant tank circuit of induction furnace

a : Phase of phase A voltage

CSI : Operating phase of current source inverter of induction furnace

f : Phase of fundamental component

_a sourceV : Phase of phase a voltage of source

CSI : Angular operating frequency of current source inverter of

induction furnace

,CSI i : Initial angular operating frequency of current source inverter of

induction furnace

XXIV

LIST OF ABBREVATIONS

AC : Alternating Current

APF : Active Power Filter

ASD : Adjustable Speed Drive

BBF : Broad-Band Filter

CSI : Current Source Inverter

CSI-APF : Current Source Inverter Based Active Power Filter

CSI-IF : Current Source Inverter Based Induction Furnace

DC : Direct Current

DFT : Discrete Fourier Transform

EAF : Electric Arc Furnace

EHV : Extra High Voltage

FFT : Fast Fourier Transform

GTO : Gate Turn off Thyristor

HAPF : Hybrid Active Power Filter

HV : High Voltage

HVDC : High Voltage Direct Current

IEC : International Electrotechnical Commission

IEEE : Institute of Electrical and Electronics Engineers

IF : Induction Furnace

IHD : Individual Harmonic Distortion

IGBT : Insulated Gate Bipolar Transistor

IGCT : Integrated Gate Commutated Thyristor

GTO : Gate Turn-off Thyristor

LV : Low Voltage

MOSFET : Metal Oxide Semiconductor Field Effect Transistor

MV : Medium Voltage

PI : Proportional Integral

PLL : Phase Lock Loop

PQ : Power Quality

XXV

PSCAD/EMTDC : Power System Computer Aided Design / Electromagnetic

Transient DC Program

PU : Per Unit

PWM : Pulse Width Modulation

RDFT : Recursive Discrete Fourier Transform

RMS : Root Mean Square

SCR : Silicon Controlled Rectifier

SHAPF : Shunt Hybrid Active Power Filter

SMPS : Switched Mode Power Supply

SMR : Switched Mode Regulator

SRF : Synchronous Reference Frame

STF : Single Tuned Filter

TDD : Total Demand Distortion

THD : Total Harmonic Distortion

UPQC : Unified Power Quality Conditioner

UPS : Uninterruptible Power Supply

VSI : Voltage Source Inverter

VSI-APF : Voltage Source Inverter Based Active Power Filter

VSI-IF : Voltage Source Inverter Based-Induction Furnace

1. INTRODUCTION Adnan TAN

1

1. INTRODUCTION

1.1. Background and Research Motivation

Induction furnaces (IFs) are used in many of mill and foundry processes such

as; heating, melting, hardening, tempering and welding. With the development of

power electronics technology, the capacity and the efficiency of IFs get higher and

become an alternative technology to electric arc furnaces (EAFs) in scrap steel

melting process. Although EAFs have major advantages in high capacity production

and melting speed, IFs are preferred in low production capacity mills because of the

low installation and operating cost and high heating efficiency.

The power supplies of IFs are formed from four main parts; rectifier, dc link,

inverter and resonance tank as shown in Figure 1.1. According to the inverter and the

resonance tank of IFs, they are divided into two type of IFs; voltage source inverter

based induction furnace (VSI-IF) which uses a series resonant circuit and current

source inverter based induction furnace (CSI-IF) which uses a parallel resonant

circuit.

3 PHASE SUPPLY

RECTIFIER DC LINK INVERTER RESONANCE TANK

Figure 1.1. General Block Diagram of Induction Furnaces

Besides the advantages of IFs in the installation and operating cost and

efficiency, IFs cause power quality problems because of the power supplies. Both

VSI-IFs and CSI-IFs generate harmonics related to pulse number of rectifier circuit

but, CSI-IFs generate specific power quality problems. Because of the current source

inverter, CSI-IFs generate time varying harmonics and interharmonics in addition to

their rectifier harmonics. In high power applications such as metal melting, current

source inverter (CSI) and parallel resonant tank circuit configuration are preferred in


2

IFs because of the advantages of parallel resonant circuit in high power applications.

High power CSI-IFs are commonly connected to distribution network through a

power transformer. Commonly, the power ratings of CSI-IFs are relatively large

compared to the distribution network so, the time varying harmonics and

interharmonics generated by CSI-IFs cause other power quality problems such as;

voltage harmonics, flicker and resonance problems in distribution networks.

To overcome harmonics, passive filters and APFs have been proposed in

literature. Passive filters are the economical way of reducing the harmonics and

meeting the reactive power requirements of nonlinear loads but, they have limited

performance on suppressing harmonics and cause resonance problems with system

impedance. The drawbacks of passive filters have led to the development of APFs.

Since 1970s, various type APF configurations have been proposed in literature to

achieve harmonic filtering, damping, isolation and termination, reactive-power

control for power factor correction and voltage regulation, load balancing, voltage-

flicker reduction, and/or their combinations (Akagi, 2005).

In recent years, the usage of high power coreless melting CSI-IFs become

widespread in steel mills. In order to prevent the effects of nonlinear load

characteristics of these CSI-IFs on power systems, appropriate compensation systems

must be applied to the CSI-IFs.

1.2. Scope of Thesis

The developments in power electronics technology perform the use of high

power converters in many of industrial applications. Besides the advantages in

efficient electric energy utilizing of these power electronics based converters, they

show nonlinear load characteristic which causes various power quality problems in

electric power systems. Induction steel scrap melting furnaces is one of the examples

of industrial applications which use high power converters in power supplies.

Induction melting is a well-known process and it has been used in many of mills

since the beginning of 1900s. Early induction melting furnaces used transformers and

motor-generator sets in the power supplies. These type power supplies cannot


3

completely meet the requirements of furnaces and cause inefficient melting operation

especially in high power melting applications. With the developments in solid state

technology, high power converters are developed and used in the power supplies of

induction melting furnaces. These power electronics based power converters provide

efficient and controllable melting operation. When these advantages of induction

power supplies are combined with the advantages of induction melting process in

high production quality, low environmental contamination and, low initial and

operating costs, induction melting furnaces become popular in metal melting process.

Today, the capacity of induction melting furnaces is reached to 50 tons and power of

these furnaces exceeds 25 MVA. With these specifications, induction melting

furnaces become a rival to electric arc furnaces in steel scrap melting process.

The 3-D drawing of a high power induction melting furnace is shown in

Figure 1.2. As shown in Figure 1.2, the power supply of high power induction

furnace is formed from transformer, rectifier, DC link, inverter and resonance tank

circuit which is formed from capacitor banks and furnace coil in the crucible. In the

power supplies of high power induction melting furnaces, the combination of 12 or

higher pulse rectifiers, current source inverters and parallel resonant tank circuits

which are formed from parallel connected capacitor bank with furnace coil are

preferred. The reason of preferring the 12-pulse or higher pulse rectifier is to

decrease the harmonic content of current drawn by the furnace. The cause of

preferring current source inverter with parallel resonance tank circuit is advantages

of parallel resonant tank circuit. When the parallel resonant tank circuit is operated at

the resonance frequency by the current source inverter, the current source inverter

supplies only the small percentage of excessive coil current of parallel resonant

circuit which is necessary for creating eddy currents in the workpieces. The coil

current is produced by the capacitors of resonant circuit in operation under resonance

frequencies. Besides the advantages of current source inverter based power supplies

of induction melting furnaces, these power supplies create unusual power quality

problems such as time varying harmonics and interharmonics. Moreover, these time

varying harmonics and interharmonics cause resonance problems and other power


4

quality problems such as voltage fluctuations and flicker in weak transmission and

distribution networks.

FURNACE

CRUCIBLE

SCRAP

METAL

POWER

CONVERTERRECTIFIER

DC LINK

INVERTER

TRANSFORMER

RESONANT TANK

CIRCUIT

CAPACITORS

WATER

COOLING

SYSTEM

Figure 1.2. 3-D Drawing of a High Power Induction Melting Furnace

(Otto-Junker GmhB)

These power quality problems of CSI-IF are investigated in a steel mill which

is located in Payas Industrial Zone in Turkey. This steel mill has two 10 MVA

coreless steel melting CSI-IFs and these furnaces are directly connected to 31.5kV

busbar which also supplies other industrial plants and domestic loads in Payas Town.

After the commissioning of CSI-IFs in the steel mill, power quality (PQ) problems

such as; voltage fluctuations, flicker and resonance problems began to appear in

power system which the steel mill is connected. These PQ problems caused damage

in reactive power compensation systems, microprocessor based controllers and

adjustable speed drives (ASD) of some other industrial plants which were supplied

from the same busbar with the steel mill. Moreover, many of home appliances

supplied from the line of domestic loads in Payas Town suffered from these PQ

problems. In order to find the source of these problems, power quality measurements

are performed at the substation which supplies the steel mill, other industrial plants


5

and domestic loads. When the power quality measurements are investigated, it is

observed that the CSI-IFs in the steel mill draw time varying harmonics and

interharmonics in wide frequency spectrum. After the detailed investigations of

power quality measurements, power supply of CSI-IFs and installed passive filtering

system at 31.5kV for CSI-IF which is formed from 5th

,7th

,11th

and 13th

harmonic

filters, it is seen that time varying harmonics and interharmonics of CSI-IFs cause

parallel resonance between the installed passive filtering system and the system

impedance. Because of the improper passive filtering compensation system, high

voltage distortion occurs in the 31.5kV busbar. Moreover, these time varying

harmonics and interharmonics of CSI-IF causes resonance problems with reactive

power compensation systems of other industrial plants and, voltage fluctuations and

flicker problems because of the weak power system. Because of these, design and

implementation of an appropriate compensation system is necessary for CSI-IFs in

order to prevent these PQ problems of CSI-IFs.

In this thesis, a CSI-IF is modeled in PSCAD/EMTDC simulation program by

using the power circuit parameters of CSI-IFs in the steel mill and knowledge

obtained during the power quality measurements of CSI-IFs. In the simulation model

of the CSI-IF, the PQ problems of CSI-IF are investigated and the simulation results

are compared with power quality measurements of CSI-IFs in the steel mill. In order

to find solutions to the PQ problems of CSI-IF, passive and active filtering methods

are modeled and investigated in PSCAD/EMTDC. In the passive filtering methods,

the characteristics of single tuned filter, C-type filter and LLCL type broadband filter

are investigated and performances of these filters on compensating PQ problems of

CSI-IF are demonstrated in PSCAD/EMTDC. In the active filtering methods, HAPF

which use shunt APF - shunt passive filter topology and APF in series with shunt

passive filters topology are proposed for the PQ problems of CSI-IF and

performances of these HAPFs on compensating PQ problems of CSI-IF are

demonstrated in PSCAD/EMTDC.


6

1.3. Outline of Thesis

After the introduction section, outline of the thesis is organized as follows;

The second chapter introduces general information about induction heating.

In this chapter, history, basic principles and applications of induction heating are

presented and general power supply topologies of induction furnaces are described.

In the third chapter, the general definitions about harmonics and

interharmonics are presented and their impacts on power systems are discussed. In

addition, harmonic and interharmonic mitigation techniques and extraction methods

are introduced.

In the fourth chapter, firstly the operating principles of CSI-IF are

investigated. Secondly, power quality problems of CSI-IF are presented and finally

the model of CSI-IF is introduced and the power quality problems of CSI-IF are

demonstrated by simulation results.

The fifth chapter presents probable passive and active filter based

compensation systems for the power quality problems of CSI-IF. In this chapter

firstly passive filtering methods are investigated and their compensation

performances on the PQ problems of CSI-IF are demonstrated in simulation results.

Afterwards, HAPF systems are proposed for the power quality problems of CSI-IF.

Their power circuit configurations and control methods are presented in detail.

Finally, the compensation performances of proposed HAPF systems on the PQ

problems of CSI-IF is demonstrated by the simulation results.

In the sixth chapter, conclusions of the thesis are given and future work

studies are discussed.

Finally all references used in this thesis study are presented.

2. INDUCTION HEATING Adnan TAN

7

2. INDUCTION HEATING

Induction heating has very important roles in many of mill and foundry

applications. The reason of popularity of induction heating is to have advanced

control in heating operation and fast heating feature in many of metalworking

processes. These features of induction heating can be achieved by application

specific coil design and applying necessary power and frequency to these specific

coils. Before several decades from today, the precision control of electric power

could not be implemented and capabilities of induction furnaces were limited.

However today, induction furnaces work under wide power and frequency range by

the help of static power converter based power supplies.

2.1. History of Induction Heating

In the beginning of 1900s, induction furnaces were firstly used in the metal

melting applications. The first induction furnace was formed from cylindrical

crucible and spark gap power supply. However, extensive application of first

induction furnaces was limited by the power attainable from spark-gap generators

(Zinn et al., 2002). In the beginning of 1920s, the motor-generator type power

supplies were invented for induction furnaces. After the development of motor-

generator based power supplies of induction furnaces, the induction furnaces were

widely used in metal melting and alloy production applications. At the motor-

generator type furnaces, a motor is connected to utility and rotates a generator which

is connected to the same shaft with motor. The generator output voltage frequency is

related to motor rotation speed and pole number of generator. When a constant

furnace frequency is employed, as is the case with motor-generators, the inductance

of the secondary circuit varies throughout the melting period, resulting in wide

changes of power factor. To prevent excessive current being drawn from the motor-

generator, switched capacitor banks are placed in parallel with the furnace coil

(Sieveking, 1940). The drawback of motor-generator type power supply is that the

desired operating conditions could not be reached by switching capacitors so the


8

efficient operating of these furnaces could not be obtained. After induction furnaces

proved themselves in metal melting processes, the attention of scientists and

engineers turned in another direction. Since the depth of the current penetration in a

given metal varies with the material electric resistivity, magnetic permeability, and

frequency, it is possible to heat specific areas of a piece of metal without heating

others. The first use of this knowledge shows itself in hardening applications of

steels (Rudnev et al., 2003).

After 1970s, the development of solid state power devices influenced to

change the motor-generator power supplies to solid state power supplies. This

change causes major advantages in induction heating technology. Today induction

heating has very wide application area by the help of solid state technology.

2.2. Basic Principles of Induction Heating

The principles of induction heating can be explained easily by the

electromagnetic theory. When the alternating voltage is applied to the coil of the

induction furnace, alternating current flows from coil. According to Ampere’s Law, a

conductor that carries a time varying current creates a time varying magnetic field so

the furnace coil produces a magnetic field at the same frequency with current passes

through coil as shown at Fig. 2(a). Moreover, a time varying magnetic field can

create an electric field with respect to Faraday’s Law. This electric field created in

the furnace coil produces eddy currents on the work pieces at shown at Fig. 2(b). As

a result, these eddy currents pieces heat up the work pieces according to Joule effect

(Rudnev et al., 2003; Zinn et al., 2002).


9

MagneticFlux

i

tEddyCurren

ii

i

(a) (b)

MagneticFlux

Figure 2.1. Principles of Induction Heating

(a) Effect of Current Carrying Conductor - Ampere's Law

(b) Effect of Magnetic Field on Metal Piece

In practice the magnetic field in the crucible and the current distribution in the

work piece are not uniform because of the several electromagnetic phenomena such

as skin effect, proximity effect and ring effect. These electromagnetic phenomena

influence directly the furnace coil design and furnace operating frequency related to

type of work pieces (Rudnev et al., 2003).

Induction heating has advantages over other type heating techniques such as

(Zinn et al.,2002; EPRI,1993);

Quick heating.

Fast start up.

Less scale loss.

High production rate.

Ease of automation and control.

Reduced floor space requirements.

Quite, safe and clean working conditions.

Low installation cost.

Low operation and maintenance requirements.


10

2.3. Applications of Induction Heating

As introduced in the history of induction heating, induction heating was

firstly used in metal melting applications. After the invention of specific area and

volume heating properties of induction heating, it was taking place in heat treatment

applications. Today, induction furnaces can work in wide power and frequency

ranges with advanced controls of power and frequency by the help of power

electronics based power supplies so induction furnaces take place in most of the mill

and foundry applications such as heating, hardening, annealing, coating, welding and

melting processes. Also induction furnaces find places in special applications in

paper industry, wood industry, wool industry, food industry, semiconductor

production. The major applications of induction furnaces are described in below

sections.

2.3.1. Heating

Induction heating is used to heat various metals to temperatures that will

allow hot or warm forming applications such as forging, rolling, extrusion and

heading. It is readily adapted to through heating of steels, aluminum alloys, and

specialty metals such as titanium and nickel-base alloys. Frequently, the workpieces

in these types of applications consist of round, square, or round-cornered square bar

stock (Zinn et al.,2002). Induction heating is also used for coating applications such

as; curing of points and vanishes, thermal spraying and galvanizing. Induction

heating has great advantage in raising the workpieces to the specified temperature

with required uniform temperature distribution. Besides the temperature uniformity,

the other advantage of induction heating is to provide the maximum production rate

at which the metal can be processed. High powers from hundreds kilowatts to several

megawatts and relatively low frequencies typically in the range of 200 Hz to 30 kHz

are most commonly used for induction heating applications (Rudnev et al., 2003).

The steel billet preheating induction furnaces are shown in Figure 2.2(a) and 2.2(b).


11

(a) (b) Figure 2.2. Steel Billet Heating Induction Furnace

(a) Short Billet Heating Induction Furnace (Rudnev, 2011)

(b) Long Billet Heating Induction Furnace (Rudnev, 2008)

2.3.2. Heat Treating

In physical metallurgy, the three main variables that are considered chemical

composition of the metal, any mechanical prior treatment and prior thermal

treatment. Any of these variables can be changed in order to produce a metal with

certain desired properties such as hardness, strength, ductility, toughness, corrosion

and wear resistance. Heat treatment deals with the effect of temperature and the rate

of heating and cooling of the metal in order to arrive at a specific microstructure and

properties (Rudnev et al., 2003). In heat treating applications, the primary advantage

of induction furnaces is the ability to control the area of material that is heat treated.

The most common heat treating applications are hardening, tempering, and

annealing.

Hardening is the most common induction heat treating operation that

improves the strength, wear resistance and fatigue properties of metals. Hardening

operation is divided into through hardening and surface hardening. The goal in

through hardening is to harden the workpiece throughout its entire cross-section. It


12

can be achieved by heating the workpiece as uniformly as possible to above the

transformation temperature and then to quench it to ambient temperature. Selection

of the correct induction heating frequency is very important to achieve uniform

“surface to core” temperature in the shortest time with the highest heating efficiency.

Through hardening may be needed for parts requiring high strength such as springs,

chain links, truck bed frames, certain fasteners, gears etc. By the help of skin effect

and some other electromagnetic phenomena, it is possible to induce power in

selected areas of the workpiece where the hardening is required. The goal in surface

hardening is to provide hardness and wear resistance on the specific outer areas of

work piece while allow the remainder of the part to be soft and ductile. Surface

hardening may be needed for parts such as gears, shafts, valves etc. The different

hardening patterns of gears which are obtained by induction hardening with different

frequencies and power is shown in Figure 2.3 and hardening operation of a gear is

shown in Figure 2.4. In Figure 2.4(a), narrow surface of gear is hardened and in

Figure 2.4(b) wide surface of gear is hardened. In some applications, tooth and holes

of gear can be separately hardened. In gap hardening of gears, special designed coils

which are coated by ceramic plates are converged with gaps between teeth of gear

and heat is applied to gaps by induction as shown in Figure 2.5. The tooth hardening

process is the same with gap hardening. This time, special design coils apply heat

treatment to each tooth (Rudnev et al., 2003; Zinn et al.,2002).

Figure 2.3. Variety of Induction Hardening Patterns Obtained Using Variations in

Frequency, Heat Time, and Coil Power (Rudnev, 2009)


13

(a) (b)

Figure 2.4. Induction Spin Hardening Process

(a) Narrow Surface Hardening Inductotherm (Rudnev et al., 2008)

(b) Wide Surface Hardening (Rudnev, 2007)

(a) (b)

Figure 2.5. Gap by Gap Hardening Operation of Gears

(a) (Rudnev et al., 2008)

(b) (Inductoheat Inc., 2011)

Tempering is often confused with annealing and sometimes referred to as

stress relieving. Tempering is a lower temperature process used to increase the

toughness and ductility of material to relieve the internal stress and eliminate the


14

brittleness caused by cold working or hardening. For example in surface hardening

only the surface layer of the workpiece is heated. The surface is raised to relatively

high temperature in a short period of time. A significant surface to core temperature

difference and the metal’s transformation phenomena upon quenching result in the

buildup of the internal residual stresses. Reheating the steel for tempering after

hardening and quenching leads to decrease or relaxation of these internal stresses. In

other words, because of tempering it is possible to improve the mechanical properties

of the workpiece and to reduce the stresses caused by the previous heat treatment

stage without losing too much of the hardness. For a particular application,

tempering can provide the optimal combination of hardness, strength and toughness.

(Rudnev et al., 2003; Zinn et al.,2002)

Annealing is the process of heating a metal and providing slow controlled

cooling to soften the metal. There are two basic types of annealing; full annealing

and process annealing. The purpose of full annealing is to decrease the hardness of

material, to maintain the full ductility of material and to create the material’s

homogenization and uniform structure. In full annealing, the material is allowed to

cool very slowly. It can take one day or longer. Process annealing is a short heat

treatment cycle than full annealing, that restores some of the ductility to a workpiece

allowing it to be worked further without breaking. Full annealing is generally

processed in gas fired or electric furnaces, induction furnaces are frequently used in

process annealing processes. (Rudnev et al., 2003; Zinn et al.,2002)

2.3.3. Melting

Electric furnaces have wide application area in various type metals melting

process. Electric furnaces which are used in metal melting process are induction

furnaces, electric arc furnaces and electric resistance furnaces. Electric resistance

furnaces have very narrow application area in metal melting process and they are

generally used in low temperature melting processes such as glass, plastic, etc.

Electric arc furnaces which heat/melt the metals by means of electric arc are mostly

preferred in steel production processes. Induction furnaces are commonly used in


15

melting of metals and production of high quality metal alloys. There are three type

induction furnaces; channel type induction furnace, coreless induction furnace and

vacuum induction furnace.

The channel type induction furnaces are used mainly for holding molten

metals as storage vessel for supplying process continuously and pouring of molten

metals which are melted previously other type furnaces such as electric arc furnaces

and coreless induction furnaces. If the channel type induction furnace is used as

melting furnace, pre-melted metal will be required for the start-up of furnace. As

shown in Figure 2.6, channel type induction furnace has a small channel of molten

metal passing through the magnetic core which has a coil wound around it. This

channel of molten metal act as like the secondary of a short circuited transformer

causing current to flow through the metal in the channel and heat the metal by the

Joule effect (Rudnev et al., 2003). The channel of furnace forms a continuous loop

with the molten metal in the main part of the furnace body. By convection, the hot

molten metal in the channel circulates into the main body of the charge in the furnace

envelope to be replaced by colder molten metal (Zinn et al.,2002). The channel type

induction furnaces are high efficiency furnaces and they work commonly under the

line frequency (Bakee, 2011). They are suitable for when high metal volumes are

desired, power outages are not expected and temperature uniformity is not critical in

the furnace vessel (Zinn et al.,2002). When channel type furnaces are compared to

the coreless furnaces; the power density of channel type furnaces are low, mixing

property of channel type furnace is weak, channel furnace cannot be started easily

when it is used as primary melting furnace, accurate input power does not necessarily

mean accurate metal temperature control throughout furnace vessel in the channel

type furnaces because of the poorer mixing (Zinn et al.,2002)..


16

Crucible

Refractory

Coil

Magnetic

Core

Channel

Figure 2.6. Cross - Section View of Channel Type Induction Furnace (Bakee, 2011)

The cross - section view of coreless induction furnace is shown in Figure 2.7.

The charge is melted inside a crucible formed from refractory lining. A cylindrical

water-cooled copper coil surrounds the outside of the refractory and produces a

magnetic field pulsating in the vertical direction when supplied with power. Outside

the coil, there are packets of magnetic laminations which provide return paths for the

flux, improving the power factor and reducing the risk of the flux linking with

surrounding metalwork to cause stray heating. The main use of the coreless furnace

is melting of metals and production of high quality alloys. The coreless induction

furnaces allow a wide range of different alloys to treat in the same foundry. It is

capable of taking a wide variety of charge material and heating it to the correct

temperature with correct composition with little or no contamination. Since the melt

is fluid, there will be hydrodynamic forces set up by the interaction of the fluxes and

currents which produce vigorous stirring; while this stirring takes place in all

induction melting. This allows a wide range of charge materials to be melted


17

economically with minimal loss of expensive alloying additions. The final mix is

homogeneous and the composition can be measured and changed quickly and easily

(Davies, 1990). The stirring action in a coreless induction furnace is directly

proportional to the power and inversely proportional to the square root of the applied

frequency (Rudnev et al., 2003). Coreless induction furnaces work commonly under

frequencies between 50 Hz to 5000 Hz. In some special small power applications,

they are designed to work under several tens of kHz. Today the power of coreless

induction furnaces reaches up to several tens of MWs and the molten metal capacity

exceeds 50 tons by the help of developments in power electronics technology.

Magnetic Yoke

Furnace

Coil

Refractory

Lining

Figure 2.7. Cross - Section View of Coreless Induction Furnace (Otto-Junker GmhB)

Vacuum induction furnaces are used for producing special metal alloys. The

melting process in vacuum induction furnace is carried out in a vacuum in order to

eliminate concerns about oxidation and metal purity during the melting and casting

processes (Rudnev et al., 2003).


18

2.3.4. Welding, Brazing and Soldering

Induction heating is also utilized in welding, brazing and soldering

applications. Induction welding, brazing and soldering provide serious energy

savings by the advantage of advance heat and heating area control. The most

common application of induction welding is tube or pipe making process which

involves the heating of a sheet of metal that has been formed into a tubular shape and

constrained in a such way that eddy currents in the workpiece cause the two open

ends of the sheet to be welded together producing the seam as shown in Figure 2.8.

Induction welding is usually a continuous operation. After welding, the seams are

then subsequently annealed with a seam annealing that follows the welding system in

a continuous line (Rudnev et al., 2003). Induction brazing and soldering which rely

on the local heating capability of induction heating are frequently used in joining of

metal workpieces. Induction welding, brazing and soldering applications operate

under high frequencies in the range of 10 kHz to 1000 kHz and relatively low power

when compared to other induction heating applications.

Figure 2.8. Induction Welding of Tubular Products (Zinn et al.,2002)


19

2.3.5. Other Applications

Besides the metalworking applications, induction heating is used in various

type applications. Some of these applications are listed and described below.

Adhesive Bonding: Induction heating is used in bonding metal to nonmetal

workpieces such as clutch plates and brake shoes, as it is used in joining

metal to metal workpieces. Induction heating of the metal parts to curing

temperatures can be an excellent means of achieving rapid bonding (Zinn et

al.,2002).

Semiconductor Fabrication: The growing of single crystals of germanium

and silicon often relies on induction heating. Zone refining, zone leveling,

doping, and epitaxial deposition of semiconductor materials also make use of

the induction process (Zinn et al.,2002).

Sintering: Induction heating is widely used in sintering of carbide preforms

because it can provide the necessary high temperature (2550 °C, or 4620 °F)

in a graphite retort or susceptor with atmosphere control. Other preforms of

ferrous and nonferrous metals can be sintered in a similar manner with or

without atmosphere protection (Zinn et al.,2002).

Cap Sealing: Cap sealing is a very important application in the food and

pharmaceutical industries. This technology provides a way that consumers

can be sure that the product is coming to them in exactly the same form and

purity in which it was packages at the factory. With this process, a small layer

of aluminum foil is placed on the top of a container that has been filled and

inspected. The container with the foil is passed under an induction coil, which

heats the foil to a sufficient temperature to bond it to top of container. The

contents are thus seal and virtually guaranteed safe at the point of final use

(Rudnev et al., 2003).

Food Industry: A variety of applications of induction heating is found in the

food industry. Many systems utilize susceptor that are heated by the induction

which in turn heat the food by conduction. Many induction food warmers and

induction stoves work by this principle. Induction extruders are used to


20

produce many types of grain transformation and confectionery products.

Large cauldrons are used for cooking caramel and other similar products and

there are also fluid heating systems used in the production of milk. One of the

final application in the food industry is the heating of rollers used to make

thin products such as pizza dough and cookies (Rudnev et al., 2003).

Papermaking: In the papermaking industry the primary application is in the

heating of calendar rolls to accurately control the thickness and quality of the

paper produced. A variety of individual coils spaced along the length of the

calendar roll. The roll temperature and paper quality are continually

monitored and the power levels are adjusted accordingly to provide the

desired temperature at each point along the length of the roll (Rudnev et al.,

2003).

Wool and Wood Processing: It is possible to utilize induction heating in

industries that require the drying of materials as they pass along a production

line or batches offline. The induction coil is used to heat a metal plate, which

in turn may contact the material and heat it by conduction and/or convection


Chemical Industry: In the chemical industry, induction heating is used to

heat various types of reactors and distillation equipment, which is used in the

production of the pharmaceutical products. In most industrial systems, the

heat that must be transferred to water in an induction system is an undesirable

byproduct of the heating process. In the chemical and food industry it may

itself be the desired end product. Some of the benefits of using induction as

opposed to open flame heating are ease of control, safety and efficiency


2.4. Power Supplies of Induction Furnaces

As mentioned in previous section, induction furnaces have very wide

application area. The effective power and frequency control of IFs in heating process


21

plays important role in finding such a wide application area. The advanced control

capability of IFs is achieved by the help of solid state based power supplies.

The induction furnaces can be classified into three groups according to

operating frequencies; low frequency furnaces, medium frequency furnaces and high

frequency furnaces. Low frequency IFs generally operate under constant line

frequencies. Medium frequency IFs operate frequency range from 100 Hz to several

tens of kHz and high frequency IFs operate frequency range between several tens of

kHz and several MHz. The power - frequency diagram of induction heating

applications is shown in Figure 2.9. Low frequency IFs are generally used in high

power applications such as melting operations. The power supply of low frequency

induction furnace is formed from a transformer, a furnace coil and compensation

capacitors. The power control of these type furnaces is achieved by the changing the

secondary voltage of transformer by tap changers. The medium and high frequency

IFs use power electronics based power supplies. These power electronic based power

supplies are frequency changers that convert the available utility line frequency

power to the desired single phase power at the frequency required by the induction

heating process (Loveless, 1995). These power supplies are commonly formed from

AC to DC converters which are named as rectifiers and DC to AC converters which

are named as inverters. Many different power supply topologies formed from the

different type rectifier and inverter circuits are used to meet the heating requirements

of a nearly endless variety of induction heating applications (Loveless, 1995).

The rectifier section of induction heating power supply is specified according

to the fixed or variable voltage requirements of inverter section. Generally,

uncontrolled rectifier, fully controlled rectifier and uncontrolled rectifier with

switched mode regulator topologies are used to supply DC power to inverters in

induction heating power supplies available on the market.

The three phase uncontrolled rectifier shown in Figure 2.10 is formed from

diodes. It produces fixed output voltage relative to the input voltage, so the power

control of induction heating must be achieved by the inverter circuit.


22

1000 MW

100 MW

10 MW

1 MW

100 KW

10 KW

Po

we

r

100 Hz 1 KHz 10 KHz 100 KHz 1 MHz 10 MHz

Frequency

MELTING

HEATING

HEAT TREATMENT

WELDING,BRAZING,SOLDERING

SPECIAL APPLICATIONS

SUPPLY

FREQUENCY

SYSTEM

SOLID STATE

POWER

SUPPLIES

VACUUM TUBE

BASED

SYSTEMS

Figure 2.9. Power - Frequency Diagram of Induction Heating Applications

3 Phase

Supply

Voltage

-

+

Constant

DC Output

Voltage

Figure 2.10. Three Phase Uncontrolled Rectifier

When the variable DC output voltage is necessary, the fully controlled

rectifier or the uncontrolled rectifier with switched mode regulator is used. The three

phase fully controlled rectifier is formed from silicon controlled rectifiers (SCRs) as

shown in Figure 2.11. It produces a variable DC output voltage by firing SCRs at

appropriate angles so the power delivered by inverter can be adjusted.


23

-

+

Variable

DC Output

Voltage

3 Phase

Supply

Voltage

Figure 2.11. Three Phase Fully Controlled Rectifier

In the uncontrolled rectifier with switched mode regulator topology, switched

mode regulator is connected to the output of uncontrolled rectifier as shown in Figure

2.11. Thanks to the switched mode regulator (SMR), the fixed output DC voltage

produced by uncontrolled rectifier can be converted to the variable DC voltage and

the power supplied to the inverter can be controlled as fully controlled rectifier. This

topology is commonly suitable for low power applications.

-

+

Variable

DC Output

Voltage

Switched

Mode

Regulator

3 Phase

Supply

Voltage

Figure 2.12. Three Phase Uncontrolled Rectifier with SMR

In high power applications above the megawatts, the rectifier section of

induction heating power supply becomes a thread for the utility because of harmonic

phenomenon. Due to the high harmonic content of six pulse rectifiers, the twelve

pulse or higher pulse rectifiers are preferred in high power induction heating power

supplies.


24

Inverters used in induction heating power supplies on the market can be

generally divided into VSI and CSI according to behavior of the load of inverter.

Basically, the load of an induction heating generator is an inductor in which there is a

workpiece to be heated. A direct feed of the heating coil would result in apparent to

real power ratio too high; therefore compensation of the heating coil is required.

Compensation of the power factor is carried out by a capacitor designed so that this

factor will be close to the unity at the working frequency. The compensation

capacitor can be placed in series or in parallel with the inductor. In the series

connection of coil and capacitor, the load acts like a current source and therefore it

has to be fed by voltage source (VSI). In the parallel connection of coil and

capacitor, when the load is a parallel resonant circuit, it will react like a voltage

source and therefore it has to be fed by a current source (CSI) (Dede et al., 1991). In

the both of VSI and CSI topologies, the single phase half bridge inverters and single

phase full bridge (H - Bridge) inverters can be used but full bridge inverters are more

commonly used than half bridge inverters in induction heating power supplies. Half

bridge inverters can be preferred to decrease the costs in low power applications.

Although the frequency and power are the significant parameters in induction

heating applications, these parameters are also very important in the design of

induction heating power supplies because the power components must be rated to

operate at the specified power and frequency. Especially, solid state power devices

used in VSIs and CSIs show variety according to operating frequency and power.

The operating power – frequency of solid state devices used in inverters of induction

heating power supplies are shown in Figure 2.13. In high power and up to 10 kHz

frequency induction heating applications, thyristor group of solid state devices can be

used in the inverters. Generally, SCRs are preferred in most of induction heating

power supplies but in some of the studies in literature gate turn - off thyristors

(GTOs) and newly integrated gate commutated thyristors (IGCTs) are used (Mertens

et al., 1991; Weber et al., 2002). Insulated gate bipolar transistors (IGBTs) are

preferred in medium power and both low and medium frequency induction heating

applications. In low power and high frequency induction applications, Metal Oxide


25

Semiconductor Field Effect Transistors (MOSFETs) are used. In the overlapped

areas shown in Figure 2.13, either type of solid state devices can be used effectively.

100 MW

10 MW

1 MW

100 KW

10 KW

Po

we

r

100 Hz 1 KHz 10 KHz 100 KHz 1 MHz 10 MHz

Frequency

IGCT

GTO

IGBT

MOSFET

SCR

Figure 2.13. Power-Frequency Diagram of Solid State Devices Used in Inverters of

Induction Heating Power Supplies

Single phase full bridge VSIs are distinguished by the use of a filter capacitor

at the input of the inverter and a series connected output circuit as shown in

Figure 2.14 (Rudnev et al., 2003). When driving a series resonant load by inverter

there will always be a phase shift between output voltage and current except

resonance frequency ideally. In the case of a series load it results that in some time

intervals, the current will flow from the load to the power source. This implies that in

an inverter with series resonant load, the switches must be bidirectional in current

and unidirectional in voltage (Dede et al., 1991). Because of this, anti-parallel diodes

are connected to the terminals of switches as shown in Figure 2.14. There are two

methods for the heating power control of VSI with series connected resonant load.

The first method is the changing of switching frequency of VSI below or above the

resonance frequency of load. When the inverter switching frequency is set to the

resonant frequency of load, the impedance of resonant load get lower value and


26

seems like resistance so the output power factor becomes unity and maximum power

is transferred to the load. By changing the switching frequency of inverter below or

above the resonance frequency of load, the impedance of series resonant load can be

increased as shown in Figure 2.14 so the power delivered to induction heating coil

can be decreased. In this method power control is achieved by inverter so it is

sufficient to use rectifier which produce fixed DC voltage. In the second method, the

switching frequency of inverter is set to resonant frequency of load. By using

variable DC voltage output rectifier, the power supplied to the VSI can be changed

because the VSI is continuously operating under resonant frequency and the

maximum power is transmitted to the load. When SCRs are used as solid state switch

at the VSI in high power applications, inverter must operate below the resonant

frequency of the load because of the commutation of inverter. The first method is

commonly used in induction heating power supplies available on the market because

only controlling the inverter heating power control can be achieved. In VSI, solid

state switches conduct the whole load current so if any short circuit condition is

occurred in the heating coil, solid state devices can be damaged.

+

-

DC

Furnace

Coil

Z

frf

Resonant Tank

Circuit

Figure 2.14. VSI with Series Resonance Load

Another load type used in VSI is series resonant circuit connected to parallel

resonant tank circuit as shown in Figure 2.15. The values of the series inductor and

capacitor are selected to be resonant above the operating or firing frequency of the

inverter with an impedance at this firing frequency that will allow sufficient current

to flow from the bridge to permit full power operation. A very important feature of


27

this style of the inverter is that the internal series circuit isolates the bridge from the

load. This protects the inverter from load faults caused by shorting or arcing and

from badly tuned loads. A second feature of this load configuration is that VSI is

capable of developing full power into the parallel resonant tank circuit tuned to either

the fundamental resonant frequency or third harmonic frequency by tuning the series

resonant circuit to the third harmonic of the resonant frequency of parallel resonant

tank circuit (Rudnev et al., 2003).

+

-

DC

Furnace

Coil

Resonant Tank

Circuit

Figure 2.15. VSI with Series Resonant Circuit Connected to Parallel Resonant Tank

The CSIs are distinguished by the use of a variable voltage DC source

followed by a large inductor at the input of the inverter bridge and a parallel resonant

load circuit at the output as shown in Figure 2.16 (Rudnev et al., 2003). In CSI with

parallel resonant load, there will be some intervals in which the output voltage is

opposite to the output current. In this case, solid state switches must be bidirectional

in voltage and unidirectional in current (Dede et al., 1991). Thyristor group solid

state devices have either high forward or reverse voltage blocking capabilities but

IGBTs and MOSFET have high forward and low reverse voltage blocking

capabilities. Thus, when the IGBTs or MOSFETs are used in CSI with parallel

resonant circuit, series connected diode is used with IGBTs or MOSFETs to increase

the reverse blocking capabilities. When SCRs are used as solid state switch at the

CSI in high power applications, the starting circuit is necessary for starting CSI and

CSI must operate above the resonant frequency of the parallel resonant load because

of the commutation of inverter (Rudnev et al., 2003; Zinn et al.,2002). The heating


28

power control is managed by variable DC voltage output rectifiers in CSI based

induction heating power supplies. The only mission of CSI is applying AC power at

resonant frequency of parallel resonant load to keep the parallel resonant load at

unity power factor and transmit the maximum power to the load. The CSI is exposed

to the full furnace voltage. However, it only sees about 10% of resonant current

because the reactive component of the coil current bypasses the inverter via the

parallel tuning capacitor (Rudnev et al., 1999).

+

-

DC

Furnace

Coil

Resonant Tank

Circuit Z

frf

Figure 2.16. VSI with Parallel Resonance Load

3. HARMONICS & INTERHARMONICS Adnan TAN

29

3. HARMONICS AND INTERHARMONICS

Advances in solid state power devices allow effective utilizing of electrical

energy using power converters in many of industrial, commercial and domestic

applications. On the other hand, developments in power electronics technology

increase the usage of power converters which have non-linear characteristics in the

power systems designed for linear loads. This contradiction causes power quality

problems in power systems. Power quality problems can be defined as any power

problem manifested in voltage, current, or frequency deviations that results of in

failure or misoperation in equipment (Dugan et al., 1996).The most suffering power

quality problems of power converters and other nonlinear loads are harmonics and

interharmonics. These nonlinear loads change the sinusoidal nature of AC current

and voltage because of the voltage drop in system impedances. These distorted

current and voltage cause the flow of harmonic and interharmonic currents in the ac

power system that can cause interference with other equipment (IEEE Std. 519,

1992).

3.1. Harmonics

Harmonics are more encountered power quality problem than interharmonics

and can be defined as the spectral components at frequencies that are integer

multiples of the ac system fundamental frequency (Testa et al., 2007). The most

common harmonic current drawn nonlinear loads are all single and three phase

power converters which contains rectifiers such as DC motor drives, adjustable speed

drives (ASD), uninterruptable power supplies (UPS), switched mode power supplies

(SMPS) etc.; cycloconverters, fluorescent lighting, electrical heating furnaces,

welding machines, arc furnaces. Besides these nonlinear loads, AC generators, AC

motors and transformers also produce harmonic currents. However; besides poor

design or fault conditions of these devices, harmonic currents of them are negligible

when compared to their fundamental currents.


30

In order to understand the amount of harmonic distortion in non-sinusoidal

voltage or current waveforms and set the limits related with harmonic distortion,

some indices are defined in IEEE and IEC standards. These most used indices in

harmonic limits defined IEEE and IEC standards are individual harmonic distortion

(IHD), total harmonic distortion (THD) and total demand distortion (TDD).

IHD is used either voltage or current harmonics and defined as the ratio of

root mean square (rms) value of each harmonic to the rms value of the fundamental

component as given in Eq. 3.1.

hV

f

hI

f

VIHD

V

IIHD

I

(3.1)

THD is also used either voltage or current harmonics and defined as the ratio

of the root-mean-square (rms) value of the harmonic components to the rms value of

the fundamental component and expressed in percent as given in Eq. 3.2. This index

is used to measure the deviation of a periodic waveform containing harmonics from a

perfect sine wave. For a perfect sine wave at fundamental frequency, the THD is zero

(Chang et al., 1998).

2

2

2

2

.100%

.100%

h

h

V

f

h

h

I

f

V

THDV

I

THDI

(3.2)

The THD of voltage is used to define the effect of harmonics on power

system voltage (IEEE Std. 519, 1992). The voltage distortion limits in low voltage,

medium voltage and high voltage power systems are defined in IEEE Std. 519, IEC

61000-2-2, IEC 61000-2-12 and IEC 61000-3-6. When the IEEE and IEC standards


31

related with voltage harmonic limits are compared, it is seen that both standards

provide voltage harmonic limits but the IEEE voltage harmonic limits are constant

across all frequencies whereas the permissible voltage harmonic magnitudes decrease

with frequency in the IEC (Halpin, 2005).

Current distortion levels can be characterized by a THD value but this can

often be misleading. A small current may have a high THD but not be a significant

threat to the system. For example, many adjustable-speed drives will exhibit high

THD values for the input current when they are operating at very light loads. This is

not necessarily a significant concern because the magnitude of harmonic current is

low, even though its relative current distortion is high (Dugan et al., 1996). Because

of these, TDD is used for characterizing current distortion limits. TDD is defined as

the ratio of the rms value of the current harmonic components to the rms value of the

maximum load (demand) current fundamental component drawn from point of

common coupling (PCC) and expressed in percent as given in Eq. 3.3. PCC in the

definition of TDD is a point of metering, or any point as long as both the utility and

the consumer can either access the point for direct measurement of the harmonic

indices (IEEE Std. 519, 1992).

2

2.100%

h

h

L

I

TDDI

(3.3)

The current distortion limits are defined both in IEEE and IEC standards.

IEEE Std. 519-1992 specify current harmonic limits for low voltage (LV), medium

voltage (MV), high voltage (HV) and extra high voltage (EHV) power systems with

respect to proportion between the demand of the load and capacity of the power

system. The capacity of the utility is defined by the short circuit current at PCC, and

the size of the load is the maximum demand load current calculated by the average of

the last twelve monthly peak demand. Therefore, according to IEEE Std. 519-1992,

the load can inject harmonic current to the utility at higher percentages as the size of

the load decreases with respect to the capacity of the system (Uçak, 2010). IEC


32

Standards for controlling harmonic distortion levels on the power system fall into

two categories. IEC 61000-3-2 and IEC 61000-3-12 outline limits for harmonic

emissions from individual equipment. These standards apply for equipment up to 75

amps on LV systems. IEC 61000-3-6 avoids giving current harmonic limits in a

general sense, preferring that these limits be more rigorously derived based on

voltage limits and system impedance characteristics for MV, HV and EHV systems

(McGranaghan et al., 2006).

Power system infrastructure and linear loads in the power systems can be

represented and modeled with resistors, inductors and capacitors. Basically; motors,

transformers, short transmission lines and system impedances can be represented as

series connection of resistors and inductors; and pure reactive power factor

compensation systems can be represented as capacitors. When a nonlinear load is

connected to power system formed from linear loads and large compensation systems

there will be created some interaction between nonlinear load and power system in

certain harmonic frequencies produced by nonlinear loads. This interaction is called

as resonance. There will occur two type resonances of in power systems; parallel

resonance and series resonance.

When the nonlinear load and compensation capacitors are connected parallel

in the same busbar as shown in Figure 3.1, the equivalent circuit of power system is

formed basically as in Figure 3.2 for harmonic frequencies. In this condition, parallel

resonance can occur so parallel connected system impedance and compensation

capacitors show high impedance as shown in Figure 3.3. Thus, the harmonic voltages

are magnified in the common busbar and high harmonic currents flows from the

compensation capacitors to system.


33

Substation

Transfomer

3 Phase

Supply

Voltage

Compensation

Capacitors

Nonlinear

Load

Common

Busbar

Figure 3.1. Single Line Diagram of Power System with Potential Parallel Resonance

Problems

Nonlinear

Load

sourceX TRX

CX

hI

Parallel Resonant

Circuit

Magnified

Current Harmonics

Caused By

Parallel Resonance

,h pI

,h pV

Magnified

Voltage Harmonics

Caused By

Parallel Resonance

EqZ

Figure 3.2. Equivalent Circuit of Power System with Potential Parallel Resonance

Problems

EqZ

fpf

Capacitor +Network Impedance

Only Network Impedance

Figure 3.3. Power System Impedance in Parallel Resonance Problems


34

When the compensation capacitors are placed under the transformer and the

transformer is connected in parallel with nonlinear harmonic producing load as

shown in Figure 3.4, the equivalent circuit of power system in shown in Figure 3.4 is

formed basically as in Figure 3.5 for harmonic frequencies. In this condition, series

connection of transformer and compensation system impedances create series

resonance in addition to parallel resonance caused by parallel connection of system

impedances and series connected transformer impedance with compensation system

impedance. Because of the series resonance, low impedance path is created to

harmonics and harmonics flow from this path. In Figure 3.6, frequency impedance

curve of power system shown in Figure 3.4 is given. High impedance point of curve

occurs in the parallel resonance frequency of the system and low impedance point of

curve occurs in the series resonance frequency of the system. Not only series

connection transformer, but also compensation system impedances and series

connection of any reactance of equipment such as power cables with capacitors can

create series resonance.

Substation

Transfomer

3 Phase

Supply

Voltage

Compensation

Capacitors

Nonlinear

Load

Common

Busbar

Transformer

Figure 3.4. Single Line Diagram of Power System with Potential Parallel and Series

Resonance Problem


35

Nonlinear

Load

sourceX TRX

CX

TRX

hI

Parallel Resonant

Circuit

Series Resonant

Circuit

Magnified

Current Harmonics

Caused By

Parallel Resonance

,h pI

,h pV

Magnified

Voltage Harmonics

Caused By

Parallel Resonance

,h sI

Current

Harmonics

Caused By

Series

Resonance

,h sVHigh Voltage

Distortion

Caused By

Series

ResonanceEqZ

Figure 3.5. Equivalent Circuit of Power System with Potential Parallel and Series

Resonance Problems

EqZ

fpf

sf

Figure 3.6. Power System Impedance in Parallel and Series Resonance Problems

Besides nonlinear loads and compensation systems, linear loads are also

connected to the power systems as shown in Figure 3.7. These loads can be modeled

as series connected resistor and inductor. These linear loads create damping affects to

parallel resonances occurring in the power system. The impedance frequency curve

of power system in Figure 3.8 is shown in Figure 3.9. With increasing ratio of

resistive load, the damping effect is increased. Not only linear loads but also power

lines, transformers and reactors have resistance. These resistances of equipment also

show damping effect on parallel and series resonance conditions.


36

Substation

Transfomer

3 Phase

Supply

Voltage

Compensation

Capacitors

Nonlinear

Load

Common

Busbar

R-L Loads

R-L Loads

Figure 3.7. Single Line Diagram of Power System with Linear, Nonlinear and

Compensation Loads

Common

Busbar

R-L Loads

R-L Loads

Nonlinear

Load

sourceX TRX

CX

TRX

hI

EqZ

Figure 3.8. Equivalent Circuit of Power System with Linear, Nonlinear and

Compensation Loads

EqZ

f

100% Resistive Load

50% Resistive Load

20% Resistive Load

Figure 3.9. Effects of Resistive Loads on Parallel Resonance


37

The effects of harmonics on equipment can be classified into short term

effects and long term effects. Short term effects are instantaneous effects that causes

generally failure, malfunctions or downgraded performance of power converters and

computer based devices through displacement of zero crossing of the voltage wave

because of the harmonics in voltage wave. Long term effects are related with

thermal losses in cables, transformers, motors, capacitors, etc. and mechanical stress

and losses in motors and generators. These problems cause excessive aging or even

damage in equipment (IEC 61000-2-2, 2002; IEEE Std. 519, 1992).

3.2. Interharmonics

Interharmonics are spectral components at frequencies that are not integer

multiples of the system fundamental frequency. Interharmonics can be observed in

an increasing number of loads in addition to harmonics. The main sources of

interharmonics are static frequency converters, cycloconverters, high voltage direct

current (HVDC) transmission systems, subsynchronous converter cascades,

adjustable speed drives, induction motors, welding machines, arc furnaces, and all

loads not pulsating synchronously with the fundamental power system frequency

(Tayjasanant et al., 2005; Yacamini, 1996)

The power converters which convert the AC input power to another AC

output power are the most suffering interharmonic sources. In these type converters,

two AC systems running at different frequencies are joined together through some

form of DC link. The problem arises if the systems are not perfectly decoupled

through the DC link. The DC voltage or current is modulated by the output frequency

of the converter and as a result interharmonic currents appear in the input current,

causing interharmonic voltages to be generated in the mains voltage. The

interharmonics are not only found in the input system also present in output AC

system. The harmonics and interharmonics produced by power converters can be

identified by the following equation; (Yacamini, 1996; IEC 61000-2-1, 1990)


38

1 1 21i of p m f p n f (3.4)

where

if : is the produces harmonics and interharmonics

1f : is the fundamental supply frequency

of : is the output frequency of the inverter

1p : is the pulse number of the rectifier

2p : is the pulse number of the inverter

m : are 0, 1, 2, 3 … (integers)

n : are 0, 1, 2, 3 … (integers)

m and n are not simultaneously equal to 0

The part formed from 1p , m and 1f in Eq. 3.4 gives the harmonic components. These

harmonic components in combination with the part formed from 2p , n and if in Eq.

3.4 give the interharmonic components. The interharmonic sources can generate

interharmonics below the fundamental frequency of power system in addition to high

frequency components. These interharmonics are named as sub – harmonics in many

of studies in literature however there is not any official definition in related

standards.

Interharmonics can create the all effects of harmonics on power systems and

equipment. In addition to these problems, interharmonics cause other power quality

problems such as subsynchronous oscillations, voltage fluctuations, and flickers.

Flickers and interharmonics have an inherent relationship. When a voltage waveform

contains interharmonics, the rms and peak magnitudes of the waveform will fluctuate

because the periods of the interharmonic components are not synchronous with the

fundamental frequency cycle. This fluctuating magnitude is essentially a form of

voltage flicker. If the magnitude is sufficiently large and the fluctuation frequency is

in a range perceptible by human eyes, a light flicker will occur. As a result, devices

that produce interharmonics have been considered as a major source of light flicker

(Testa et al., 2007; Tayjasanant et al., 2005). The only limitations about


39

interharmonics are recommended by IEC 61000-3-6 standard. IEC 61000-3-6

addresses interharmonic voltage limits by recommending a frequency-independent

limit of 0.2% so as to avoid problems with lamp flicker and ripple control, signaling,

and communications equipment (Halpin, 2005).

3.3. Harmonic and Interharmonic Mitigation Techniques

In order to mitigate harmonics and interharmonics, there are two main

approaches. The first approach is to make modifications in the power circuits of the

nonlinear loads and the second approach is to design compensation systems for

nonlinear loads.

In order to solve the harmonic problems of power converters, the most

preferred method is using higher pulse rectifiers in the rectifier section of power

converters with phase multiplication method. Another method is using pulse with

modulation (PWM) rectifiers which are able to provide variable DC link voltage with

generating low harmonic current content. The main drawback of both of these

methods is high costs. Using high pulse rectifiers with phase multiplication method is

preferred in most of the high power applications. However, although the harmonic

content of currents drawn from power systems is decreased seriously, the harmonics

of converters will be still over the limits. In PWM rectifiers, IGBTs or MOSFETs are

used instead of diodes or thyristors in classical rectifiers. Thanks to this difference,

the harmonic content of power converter can be decreased under limits by

controlling the current generated by power converter. There is a limited number of

PWM rectifier based products in market and these products are produced for low

voltage applications. The PWM rectifier based power converters are not preferred

and they are unpractical in many of applications because of the extremely high costs

and complex power circuit and control systems configurations.

Harmonic compensations systems are the most preferred approach in practical

harmonic mitigation applications. There are two main methods; passive filters and

active filters to compensate the harmonics and interharmonic. In order to control the

harmonic and interharmonic distortion generated by nonlinear loads, passive filters


40

have been used over many years. Passive filters are very popular because of the low

installation cost and meeting the reactive power requirements of nonlinear loads

while filtering the harmonics. However, the increasing use of nonlinear loads cause

complicated load characteristics shown in power systems and passive filters become

insufficient because of the drawbacks in the system impedance dependent filtering

performance and the resonance problems. This limited performance of passive filters

causes the development of active filters. The active filters are power electronics

based harmonic compensation systems and they are superior in harmonic filtering

performance, smaller in physical size, and more flexible in application compared to

traditional passive harmonic filters. The main purpose of the usage of APF is the

harmonic filtering but APF can be developed as a power conditioning device that can

solve simultaneously serious power quality problems such as unbalance loading,

reactive power and flicker additional to harmonics (Akagi, 2005).

3.3.1. Passive Filters

Passive filters which are formed from the various connections of capacitors,

inductors, and/or resistors have been used to overcome harmonic currents for a long

time. The passive filters can be classified into series passive filters, shunt passive

filters and broadband passive filters. The series passive filters are connected in series

with nonlinear loads and they create series high impedance to block the harmonic

currents. The shunt passive filters are connected in parallel with nonlinear loads and

they provide low-impedance paths for harmonic frequencies, thus resulting in

absorbing the dominant harmonic currents flowing out of the load current. The

broadband filters are formed from the series and parallel connection of passive

devices and they are used to block multiple or widespread harmonic frequencies

(Akagi, 2005; Dugan et al., 1996; Uçak, 2010).

Series passive filters are formed from parallel connection of inductor capacitor

as shown in Figure 3.10. The series passive filters are tuned to provide high

impedance at a selected harmonic frequency. The use of the series filters is limited in

blocking multiple harmonic currents. Each harmonic current requires a series filter


41

tuned to that harmonic. As all the series components, series filters are subjected to

full line current while shunt connected passive filters carry only a fraction of line

current. Moreover, the reactive power compensation capability of shunt connected

passive filters and the lower installation cost of shunt filters make series passive

filters non preferable (Dugan et al., 1996; Uçak, 2010).

L

C

3 Phase

Supply

Voltage

Nonlinear

Load

Figure 3.10. Series Passive Filter

Shunt passive filters can be classified into tuned filters, high pass filters and

C-type filters. The power circuit topologies and impedance frequency curves of the

most common type filters are shown in Figure 3.11. Single tuned filters are the most

commonly used type of the shunt passive filters. Single tuned filters are also named

as band-pass filters or notch filter in literature. This type filters tuned to a specific

frequency to present low impedance to a particular harmonic current. Thus, harmonic

currents are diverted from their normal flow path on the line through the filter. This

filter type is the most economical type and is frequently sufficient for the application

(Dugan et al., 1996). Single tuned filters are generally used for power factor

compensation in addition to harmonic compensation. In fact in many cases, single

tuned passive filters are used primarily for power factor correction instead of

compensation capacitors to avoid the resonance problems. High pass filters are also

named as damped filters. These filters provide low impedance for a wide spectrum of

harmonics without the need for subdivision of parallel branches with increased

switching and maintenance problems. The main drawback of high pass filters is

higher losses in the resistors and reactors. Another drawback is to achieve a similar

level of filtering performance as single tuned filters, high pass filters must be


42

designed for higher fundamental volt-ampere (VA) ratings. In many of cases it is

impossible to get sufficient filtering performance as single tuned filters with not

exceeding the reactive power compensation requirements. Because of these, high

pass filters are generally designed in low power ratings to damp the high order

harmonics. The three types of high pass filters and their characteristic as shown in

Figure 3.12. First order high pass filter is not preferred in many of applications,

because it has not sufficient filtering performance and it has excessive loss at the

fundamental frequency. The second order filter high pass filter has better

performance on filtering but higher losses with compared to third order high pass

filter (Arrillaga, 1985). The C-type filters are also type of damped filters, the filtering

performance of this type filter is between the second order and third order high pass

filters as shown in Figure 3.12. In C-type filter, C2 and L are tuned to fundament

frequency so in the fundamental frequency series connected C2 - L branch is short

circuited and resistor R is bypassed. Because of these, C-type filters get significant

advantage in fundamental frequency losses. C-type filters are generally tuned

frequencies between 2nd

and 6th

order harmonics and used in arc furnace and HVDC

compensation systems (Nassif et al., 2009).

R

L

C

R

CR

C

L

R

2C

L

1C

R

2C

L

1C

(a) (b) (c) (d) (e)

Figure 3.11. Shunt Passive Filter Topologies (a) Single Tuned (b) First Order (c)

Second Order (d)Third Order (e) C-Type


43

Figure 3.12. Shunt Passive filters Impedance - Frequency Curves (Nassif et al., 2009)

The basic circuit topology of broadband passive filter is shown in Figure

3.13. In many of practical applications, multiple stage shunt or series passive filters

must be used for reducing the different order harmonics under the limits. Unlike the

shunt and series filters that have a narrow band of harmonic suppression, broadband

filters have a wider range of harmonics suppression property. Broadband filters

employ a combination of the two passive techniques, with a high series impedance to

block the undesired current harmonics (from flowing through the grid) and a low

shunt impedance path to divert their flow through the shunt filter (Dugan et al., 1996;

Win, 2008).

L

C

3 Phase

Supply

Voltage

Nonlinear

Load

Figure 3.13. Broad-Band Passive Filter


44

Besides the advantages of passive filters in wide application area, passive

filters have significant drawbacks such as (Das, 2004; Fujita et al., 1991);

The power system impedance has considerable effect on filtering

performance of passive filter. It is difficult to anticipate the changes in the

power system impedance in the design stage.

Parallel resonance may be formed between a source and a passive filter and

causes amplification of harmonic currents on the source side at specific

frequencies.

Passive filters may fall into series resonance with a source so that voltage

distortion produces excessive harmonic currents flowing into the passive

filter.

Passive filters are not suitable for changing system conditions. Once installed,

these are rigidly in place. Neither the tuned frequency nor the size of the filter

can be changed so easily.

The aging, overloading and temperature can change the values of components

in passive filter so these effects can change the tuning frequency of passive

filter.

3.3.2. Active Power Filters

The limited performance of passive filters enforces to find the new solutions

for harmonic mitigation and provide the invention of APFs. In the beginning of

1970s, the basic principle of shunt APFs was originally presented by H. Sasaki and

T. Machida by using linear amplifiers. In 1976, L. Gyugyi and E. C. Strycula

presented a family of shunt and series active filters, and established the concept of

the active filters consisting of PWM inverters using power transistors (Peng et al.,

1990). For over 40 years, the scientists and engineers have been working on APFs

and by the help of developments in power electronics and microprocessor

technology, various type APFs have been developed to solve many of power quality

problems. Today APFs not only achieve harmonic filtering but also solve the various


45

type power quality problems such as reactive power, unbalanced loading, voltage

fluctuation and voltage flicker.

The block diagram of basic power circuit configuration of active power filter

is shown in Figure 3.14. The power circuit of APFs is mainly formed from three part;

output filter, inverter and DC link. Output filter is used for linking the inverter to the

power system and eliminating the switching ripples created by the inverter. The

inverter is main part of the power circuit and various type inverter topologies are

used for generating compensation currents or voltages. The switching devices used in

inverters are selected according to the power ratings of APF. The most common

switching devices used in inverters are MOSFETs and IGBTs. MOSFETs are low

power devices and they have high switching frequency capabilities. Adversely

IGBTs can work under higher powers but their switching frequency is lower than

MOSFETs. DC link of active power filter is formed from inductor or capacitor

depending on the type of inverter. DC link is used as energy storage device. APF

eliminates harmonics and/or other power quality problems by supplying reactive

energy to DC link and/or consuming reactive energy from DC link and additional DC

supply is not necessary. The only active power requirement of active power filter is

for losses of its power circuit.

INVERTEROUTPUT FILTER DC LINK

Figure 3.14. Block Diagram of Power Circuit of APF

The APFs can be classified based on topology, converter type and supply

system. In the topology based on classification, APFs can be classified into shunt

(parallel) APFs, series APFs, combination of both series and shunt APFs and HAPFs.

The shunt APFs is the most preferred type active filter in practical applications. The

shunt APF is connected in parallel with nonlinear load as shown in Figure 3.15. The


46

basic operating principle of shunt APF is to detect the harmonics from the distorted

current drawn by nonlinear load and then inject the reverse of harmonic currents to

the power system to eliminate the harmonic currents from the source current. Shunt

APFs can also achieve reactive-power compensation, balancing of three-phase

currents and voltage fluctuation and flicker reduction caused by nonlinear load

current.

Isource Iload

IAPF

APF

NonlinearLoad

Utility

Figure 3.15. Block Diagram of Shunt APF

The series APF is connected in series with the power system using a

transformer as shown in Figure 3.16. The main operating principle of series APF is

to inject a harmonic voltage in series with the line to render the supply current

sinusoidal (Bhattacharya et al., 2010).The approach is based on a principle of

harmonic isolation by controlling output voltage of the series APF. In other words,

the series APF is to present high impedance to harmonic current, therefore blocking

harmonic current flow from the load to the ac source and from the ac source to the

load side (Peng ,1998). The main drawback of series active power filter is to handle


47

the high load currents which increase their current ratings and losses (Habrouk et al.,

2000).

VAPF

APF

NonlinearLoad

Utility

+ -Isource Iload

Figure 3.16. Block Diagram of Series APF

The combination of shunt and series active filter which is called unified

power quality conditioner (UPQC) is shown in Figure 3.17. UPQC is formed by

connecting the series APF and the shunt APF using the same DC link. UPQC is

advanced power quality compensation device and it can solve voltage sag and swell

in addition to the power quality problems solved by both of series and parallel active

power filters. Its main drawbacks are its large cost and control complexity because of

the large number of solid-state devices involved (Singh, 1999).


48

VAPF

Utility

+ -

IAPF

NonlinearLoad

DC Link

Series APF Shunt APF

Isource Iload

Figure 3.17. Block Diagram of UPQC

HAPFs are formed using active power filters and passive filters are together.

The most popular hybrid power topologies are shown in Figure 3.18. In Figure

3.18(a), the series active power filter and shunt passive filter combination is shown.

In this configuration, while series active power filter compensated voltage

harmonics, it also constitutes a high impedance to increase the harmonic filtering

performance of the parallel passive filter. The drawbacks of series active power filter

constraint the use of this topology. In Figure 3.18(b), the shunt active power filter

and shunt passive filter combination is shown. In this topology shunt passive filter is

used to achieve reactive power compensation and eliminate the part of the harmonics

for reducing the ratings of active power filter. Another popular hybrid active power

filter topology is shown in Figure 3.18(c). In this topology active power filter is

connected to shunt passive filter in series with transformer. In low and medium

voltage applications, transformer can be removed and active filter can be connected

to passive filters directly. This combination of active power filter and passive filter is

significantly important for medium and high voltage applications because series

connection with passive filter reduces the voltage stress applied to the switches in the

active power filter. This specification reduces the rating of the active filter up to tenth

of the ratings of conventional active power filters and provides the ability of

compensation in high voltage applications. filters The operation principle of this type


49

hybrid is slightly different from conventional shunt connected passive filters. The

task of these hybrid active power filter are not to compensate for harmonic currents

produced nonlinear loads, but to achieve “harmonic isolation” between the supply

and the load. As a result, no harmonic resonance occurs, and no harmonic current

flows in the supply (Habrouk et al., 2000; Singh, 1999; Akagi, 2005).

VAPF

Utility

+ -

Nonlinear

Load

Series APF Passive Filter

Utility

+

Nonlinear

Load

Passive Filter

IAPF

Shunt APF

IPF IPF

Utility

Nonlinear

Load

Passive Filter

IPF

VAPF

+

-

Series APF

(a) (b)

(c)

Isource Iload Isource Iload

Isource Iload

Figure 3.18. Hybrid APF Topologies

(a) Series APF and Shunt Passive Filter Topology

(b) Shunt APF and Shunt Passive Filter Topology

(b) APF Series with Shunt Passive Filter Topology

In the converter based on classification, active power filters can be classified

into voltage source inverter based active power filter (VSI-APF) and current source

inverter based active power filter (CSI-APF). In CSI-APF, inductors are used as DC

link energy storage device and series connected diodes are used with switching

devices to maintain the reverse voltage blocking capability as shown in Figure

3.19(a). In VSI-APF, capacitors are used as DC link energy storage device and anti-


50

parallel diodes is connected to switching devices for allowing reverse current flow as

shown in Figure 3.19(b). Both topologies can achieve the sufficient harmonic

filtering performance. Although CSI-APF has faster dynamic response than VSI-

APF in current harmonic compensation, VSI-APF is more preferred in practical

applications because of the disadvantages of CSI-APF in high losses caused by DC

link inductor and necessity of large place for bulky DC link inductor and also the

advantages of VSI-APF in high efficiency, smaller size and the low initial costs

(Benchaita et al., 1999; Routimo et al., 2007; Akagi, 2005).

IAPF

NonlinearLoad

Utility

IAPF

NonlinearLoad

CDCLDC

(a) (b)

Isource Iload Isource IloadUtility

Figure 3.19. Converter Based on Classification of APFs (a) CSI-APF (b) VSI-APF

In the supply system based on classification, APFs can be classified into

single phase active power filters, three phase three wire active power filters and three

phase four wire active power filters. Most of commercial and domestic single phase

loads show nonlinear load characteristics and produce harmonics including third

order harmonics. In small commercial and domestic buildings, single phase active

power filters can be used for eliminating harmonics. The large domestic or

commercial plants such as hospitals, shopping centers etc. are generally supplied

from three phase four wire systems. Because of third harmonic currents of the single

phase nonlinear loads and unbalanced loading in three phase four wire systems, there


51

will occur overcurrent in the neutral conductor. Theoretically, third harmonics cannot

be eliminated by three phase three wire active power filters so, in order to solve these

problems in three phases four wire systems, four wire active power filters are widely

used. In industrial applications, loads are generally formed from balanced three phase

loads and third harmonic problems are rarely encountered. Thus, in industrial

applications three phase three wire active power filters are commonly used.

3.4. Harmonic and Interharmonic Extraction Methods

Harmonic extraction methods have significant importance in active power

filter performance. In order to achieve effective harmonic compensation, the

reference harmonic currents must be calculated as possible as accurate and fast.

Moreover the calculation complexity of harmonic extraction methods must be low

for the efficient use of resources of active power filter controllers. In literature,

various type harmonic extraction methods are proposed. These methods can be

mainly classified into time domain methods and frequency domain methods. In Table

3.1, the classification of most popular harmonic extraction methods are presented.

Table 3.1. Harmonic Extraction Methods

Time Domain

Methods

PQ Theory

Synchronous Reference Frame (SRF)

Filtering Methods; Band pass filters

Adaptive Filters

Neural Networks

Frequency Domain

Methods

Fast Fourier Transform (FFT)

Recursive Discrete Fourier Transform (RDFT)

Wavelet Transform

Kalman Filters


52

The time domain methods offer increased speed and fewer calculations

compared to the frequency-domain methods. PQ Theory and SRF methods are the

most popular time domain harmonic extraction methods. Both of these methods need

voltage and current measurements and they suffer from non-sinusoidal supply

voltage. Non-sinusoidal voltage supply directly affects the performances of these

methods. In literature, some improvements are proposed to improve PQ theory and

SRF methods. In SRF method, selective harmonic extraction can be achieved by the

help of PLL used in method. The most common frequency based methods are

Fourier based methods and wavelet method. The limitations of the frequency domain

approaches are that the designer has to consider the effect of aliasing. The

antialiasing filter used for this purpose is required to be very accurate; otherwise, the

whole calculation will be erroneous. Sampling instant and zero crossing of

fundamental are required to be synchronized. Otherwise, phase estimation by this

process will be erroneous. If number of samples is not a power of two, then zero

padding is required. Moreover, this method of analysis is very susceptible to noise

and transients (Asiminoaei et al., 2007; Bhattacharya et al., 2009).

4. MODELING AND ANALYSIS OF CSI-IF Adnan TAN

53

4. MODELING AND ANALYSIS OF CSI-IF

In the modeling studies of CSI-IF, the coreless CSI-IF in the steel mill

mentioned in the introduction section is taken as a base model and CSI-IF is modeled

by using the power circuit and operating parameters of the coreless CSI-IF in the

steel mill. The modeling purpose of this CSI-IF is to show and analyze the power

quality problems of furnace. Thus, the furnace coil analysis and the effect of furnace

coil in the melting process do not take place in this thesis content. The power circuit

topology of the coreless CSI-IF in the steel mill is shown in Figure 4.1. The furnace

is formed from 12 MVA three phase double secondary transformer which is

connected to 31.5 kV busbar, twelve pulse fully controlled rectifier, DC link

inductors, H-bridge inverter and parallel resonant tank circuit. This furnace has 10

MVA power converter and the current source inverter works between 150 and 250

Hz frequencies to perform melting process at rated power. In this section of thesis the

operating principles and power quality problems of CSI-IF power supply and

modeling studies of CSI-IF is presented.

3 PHASE

DOUBLE SECONDARY

TRANSFORMER

12 MVA

31.5 kV/1.2 kV -1.2kV

Δ/ Δ-Y

FULLY CONTROLLED

12 – PULSE RECTIFIER

DC LINK

CURRENT

SOURCE

H – BRIDGE

INVERTER

PARALLEL

TUNING

CAPACITOR FURNACE COIL

PARALLEL

RESONANT TANK

CIRCUIT

31.5kV BUSBAR

Ssc = 335MVA

Figure 4.1. Power Circuit Topology of the Coreless CSI-IF in The Steel Mill


54

4.1. Operating Principles of CSI – IF Power Supply

The power circuit of CSI-IF shown in Figure 4.1 is mainly formed from three

parts; parallel resonant tank circuit, current source H bridge inverter, and 12-pulse

fully controlled rectifier. The general operating principle of CSI-IF is described in

Section 2.4. The heating power control in CSI-IF is performed by the variable DC

voltage output rectifier and CSI keep the parallel resonant tank circuit in the

resonance in order to maintain the flow of resonance current to the furnace coil.

These operating principle of CSI-IF is analyzed by investigating the main parts of

power supply of CSI-IF individually.

4.1.1. Parallel Resonant Tank Circuit

The parallel resonance circuit tank circuit of coreless CSI-IF is formed from

furnace coil and parallel connected capacitors. Before proposing the characteristic of

parallel resonant circuit, the electrical characteristics of furnace coil must be known.

The coreless induction furnaces are electrically analogous to the transformers. The

equivalent circuit approximation of transformer is used for furnace coil of coreless

CSI-IF (Tremayne, 1983). The equivalent circuit of a transformer is shown in Figure

4.2. PrR and PrX are the primary winding resistance and leakage reactance of

transformer. CR is the core resistance and MX is the magnetizing reactance of

transformer. PN is the primary winding turn number and SN is the secondary

winding turn number. SR and SX are the secondary winding resistance and leakage

reactance of transformer. LR and LX are the load resistance and load reactance.

When this transformer model is applied to coreless induction furnace, PR and PX

are the coil resistance fcR and reactance

fcX of furnace. CR and MX can be

neglected in the furnace model. Because of the coreless structure, their effects are

very less. PN is equal to coil turn number coilN and SN is equal to 1. SR and SX are

not used in furnace model because the secondary winding is absent in the coreless


55

furnace. gapX is the reactance of insulation material and gap between coil and inside

of crucible. The load resistance LoadR and reactance LoadX are equal to the metal

charge resistance chR and reactance chX of coreless induction furnace. The

resistance chR and inductance chX of the charge changes depending on the type of

metal charge, temperature of charge and amount of charge in the furnace. The

approximate equivalent circuit of coreless induction furnace is shown in Figure 4.3

and the equivalent circuit of furnace referred to primary side is shown at the Figure

4.4. fR is the equivalent furnace coil resistance which is equal to the sum of coil

resistance fcR and charge resistance referred to primary side

,ch refR . fX is the

equivalent furnace coil reactance which is equal to the sum of furnace coil reactance

fcX , gap reactance referred to primary side ,gap refX and load reactance referred to

primary side,ch refX . Also,

fR and fX are variable depending on the chR and chX

(Tremayne, 1983; Tokunç, 2010).

+ +

PrR

PrX

PrV CR MX

:P S

N NSR

SX

SVLoadR

LoadX--

Figure 4.2. Equivalent Circuit of Transformer

:1coil

N

+

PV

-

+

SV

-

fcR fcX

chR

chX

gapX

Figure 4.3. Equivalent Circuit of Coreless Induction Furnace


56

+

PV

-

, , f fc gap ref ch refX X X X,f fc ch ref

R R R

Figure 4.4. Coreless Induction Furnace Equivalent Circuit Referred to Primary Side

The parallel connected furnace coil and capacitors form a parallel resonance

circuit. If the resonance occurs in the electric circuits, the energy absorbed by one

reactive element is the same as that released by another reactive element within the

system. In other words, energy pulsates from one reactive element to the other.

Therefore, once an ideal system has reached a state of resonance, it requires no

further reactive power and they only deliver active power (Boylestad, 2007). Ideal

parallel resonance circuit is formed parallel connected resistance, inductance and

capacitance in parallel with a current source as shown in Figure 4.5.

PR PL PCI

,P PZ Y

PV

-

+

Figure 4.5. Ideal Parallel Resonant Circuit

The impedance-frequency curve of parallel RLC circuit is shown in Figure

4.6. In the resonance conditions of ideal parallel RLC circuit, the impedance of RLC

circuit reaches its maximum value and the reactance of PL is equal to the reactance

of PC so the resonance frequency of parallel RLC circuit is found as;

P PL CX X (4.1)


57

12

2P P

p P

f Lf C

(4.2)

1

2P

P P

fL C

(4.3)

PZ

mPZ

Pf

CapacitiveInductive

Figure 4.6. Impedance-Frequency Curve of Ideal Parallel Resonant Circuit

However, in practical applications as in the parallel resonant tank circuit

which is formed from parallel connected capacitor group with the furnace coil,

inductor has initial resistance and connected series with the inductance of inductor as

shown in Figure 4.7. If this circuit is converted to the parallel network equivalent

shown in Figure 4.8, the PR and PLX is equal to

2 2

LL L

P

L

R XR

R

(4.4)

2 2

L

P

L

L L

L

L

R XX

X

(4.5)


58

LL

PCI

LR

Figure 4.7. Practical Parallel Resonant Circuit

2 2

L

L

L L

P

L

R XR

R

2 2

L

P

L

L L

L

L

R XX

X

PCXI PV

-

+

Figure 4.8. Practical Parallel Resonant Circuit Converted To Ideal Parallel Resonant

Circuit Form

In the ideal parallel resonance circuit, the resonant frequency is the frequency

at which the impedance was a maximum, the current a minimum, and the input

impedance purely resistive, and the network have a unity power factor. However, for

the practical parallel circuits, since the resistance PR in our equivalent model is

frequency dependent, the frequency at which maximum PV is obtained is not the

same as required for the unity-power-factor characteristic (Boylestad, 2007).

For unity power factor, the power must be zero so; the inductive reactance

must be equal to capacitive reactance as in Eq 4.6.

P PL CX X (4.6)

With using Eq. 4.5, Eq.4.6 can be written as;

2 2

L

P

L

L L

C

L

R XX

X

(4.7)


59

From Eq. 4.7 unity power factor frequency uf is;

211

2

L Pu

LL P

R Cf

LL C (4.8)

The maximum impedance frequency of practical parallel resonant circuit can

be calculated as differentiating the impedance of parallel resonant circuit and then

determining the frequency at which the resulting equation is equal to zero. From

using this way, the maximum impedance frequency can be expressed as (Boylestad,

2007);

21 11

42

L Pm

LL P

R Cf

LL C

(4.9)

When Eq. 4.8 is compared to Eq. 4.9., it is seen that;

m uf f (4.10)

The quality factor of resonant circuit is equal to the ratio of reactive power to

the active power at resonance frequency. For practical parallel resonant circuit

quality factor is calculated as;

2 22

2 2 2

L

P

L

L

L LP

L LP

P L L

PL

R XVX R

QFV R X

R X

(4.11)

so that


60

LL

P

L

XQF

R (4.12)

The quality factor of parallel resonant tank circuit of coreless induction

furnace is considerably high (Chudnovsky et al., 1997; Rudnev et al., 2003). If the

quality factor of practical parallel resonant load PQF is larger than 10, the following

approximation can be done. If Eq. 4.4 is reorganized as (Boylestad, 2007).,

21P P LR QF R (4.13)

then, for 10PQF , 2 21 P PQF QF so

2

P P LR QF R (4.14)

If Eq. 4.5 is reorganized as,

2P

LL L

P

XX X

QF (4.15)

then, for 10PQF , 2 0L PX QF so

PL LX X (4.16)

If Eq 4.8 is reorganized with using Eq. 4.14 (Boylestad, 2007),

2

1 11

2u

PL P

fQFL C

(4.17)

then, for 10PQF , 21 0PQF so


61

1

2u

L P

fL C

(4.18)

,If Eq 4.9 is reorganized with using Eq. 4.14 (Boylestad, 2007),

2

1 1 11

42m

PL P

fQFL C

(4.19)

then, for 10PQF , 21 0PQF so

1

2m

L P

fL C

(4.20)

When the quality factor of practical parallel resonant load PQF is larger than

10, the capacitor current and inductor current is investigated by the following

equations. When the practical parallel resonant circuit in resonance, by using Eq. 4.6

and Eq. 4.16 the capacitive reactance of circuit is defined as;

P P LC L LX X X (4.21)

so the total impedance of practical parallel resonant circuit is calculated as,

1 1 1

P P

p

P L C

ZR jX jX

(4.22)

By using Eq. 4.14 and Eq. 4.21, At the resonance frequency maximum value of PZ is

found from Eq. 4.22 (Boylestad, 2007),


62

2

P P LZ QF R (4.23)

For 10PQF , from Ohm’s Law the voltage between the terminals of practical

parallel resonant circuit at the resonance frequency is calculated as;

2

P P LV IQF R (4.24)

and the capacitor and inductor current is calculated as;

2

P

P P

P P LC

C C

V IQF RI

X X (4.25)

2

P

P P

P P LL

L L

V IQF RI

X X (4.26)

For 10PQ , by using Eq. 4.12 and Eq. 4.21, Eq. 4.25 and 4. 26 can be organized as;

PC PI QF I (4.27)

PL PI QF I (4.28)

These approximations show that if the quality factor of practical parallel

resonant circuit is equal or larger than 10, this parallel resonant circuit is approved as

ideal parallel resonant circuit. Thus, the maximum impedance frequency and unity

power factor frequency of practical parallel resonance circuit can be accepted as

equal. Also from Eq. 4.27 and Eq. 4.28, it is shown that at the resonant frequency,

approximately while current source produce constant value of current, the quality

factor times of current of source flows from inductor and capacitor.


63

The parallel resonant tank circuit of coreless melting furnace is accepted as

ideal parallel resonance circuit, because of the high quality factor of parallel resonant

tank circuits. Hence these approximations are valid for the resonant tank circuit of

coreless melting CSI-IF. The advantage of parallel resonant tank circuit is shown in

Eq. 4.28. By using parallel resonant tank circuit with high quality factor excessive

high current can be drawn by the furnace coil while the switching devices used in the

CSI see only the constant value source current.

4.1.2. CSI

In order to keep the parallel resonant tank circuit in resonance, CSI must be

used. In low and medium power applications, IGBTs or MOSFETs can be used as

switching devices in inverters. However in high power applications, the capabilities

of IGBTs and MOSFETs are inadequate so SCRs are preferred in high power

applications as the high power melting CSI-IFs. When SCRs are used as power

switched in the inverters, load commutation must be maintained. In the load

commutation, the proper operation of these switches requires that a reverse voltage

be maintained across the switch, in order to turn off the SCRs. The reverse voltage

across the power switches can be provided with leading power factor load (Dawson

et al., 1991; Bose, 1986). The parallel resonant tank circuit of coreless CSI-IF can act

as both inductive load and capacitive load according to the operating frequency as

mentioned in previous section. The operating principles of CSI with SCRs and why

the capacitive load is necessary for the load commutation are proposed by giving the

capacitive mode operation waveforms of parallel resonant tank circuit.

The CSI with the parallel resonant tank circuit of IF is shown in Figure 4.9.

As mentioned in previous section; if the ideal or high quality factor parallel resonant

load operates below the resonant frequency of the parallel resonant load, it acts as

inductive load and if it operates in above the resonant frequency of the parallel

resonant load, it acts as capacitive load. In Figure 4.10, the waveforms of parallel

resonant operating above the resonant frequency are shown. If the current source is

accepted as an ideal source, the current waveform is formed from square wave and


64

voltage across the parallel resonant load is formed from sinusoidal wave in the

parallel resonant circuit. And also current leads the voltage because parallel load acts

as capacitive load above the resonant frequency.

LL

PC

LR

+

-

DCV

I

PV -+1Q

3Q

4Q

2Q

Figure 4.9. CSI of IF with Parallel Resonant Tank Circuit

I

PV

st

rf f Capacitive

Figure 4.10. Voltage and Current Waveforms of Parallel Resonant Tank Circuit

Operating Above the Resonance Frequency

The SCRs in the CSI are switched alternately for 180° to form a square

current wave. During the positive cycle of current 1Q and 2Q are switched on as

shown in Figure 4.11. When at the st time 3Q and 4Q are triggered and the voltages


65

are formed instantaneously for the upper switches as shown in Figure 4.12(a) and

lower switches as shown in Figure 4.12(b).

LL

PC

LR

PV -+

I

+

-

DCV

1Q

3Q

4Q

2Q

Figure 4.11. Operation of Load Commutated CSI

LL

PC

LR

+

DCVPV -+1

Q3

QLLLR

-DCV 4

Q2

Q

PC

PV -+

(a) (b)

Figure 4.12. Load Commutation of CSI

(a) Top Switches

(b) Bottom Switches

From the Kirchhoff’s Mesh Law, the voltage shown between the anode and

the cathode terminals of 1Q and 2Q is expressed as;

1Q PV V (4.29)


66

2Q PV V (4.30)

As shown in Figure 4.12 at time st , the magnitude of

PV is positive and

negative voltage is formed across the between the anode and the cathode terminals of

1Q and 2Q so that the 1Q and 2Q are closed and 3Q and 4Q is getting on. If the load is

inductive at the switching time of 3Q and 4Q , the voltage across the terminals of

load is negative and positive voltage is seen across the anode - cathode voltage of 1Q

and 2Q so that the 1Q and 2Q are not closed and the CSI cannot operate. Because of

these, in order to operate the CSI with using SCRs, the load must be capacitive.

Besides requirement of capacitive load for load commutation of CSI, CSI

requires an initial pre-charge on the parallel capacitor in order to initiate oscillations

in the parallel resonant circuit and built up sufficient voltage across the terminals of

the parallel resonance load circuit. Because of these problems, various type starting

circuit topologies are developed (Dawson et al., 1991; Bonert et al., 1994). Starting

circuits are used only in several cycles of inverter and then detached from the

inverter.

In the CSI-IF, the furnace coil is varied dependent to the temperature and the

amount of the charge material and fixed value capacitor is used to form parallel

resonant tank circuit. CSI of coreless melting IF provides the high current flow from

the furnace coil and load commutation for switching of SCRs by operating

continuously slightly above the resonance frequency of varying load.

4.1.3. 12-Pulse Fully Controlled Rectifier

The heating power in CSI-IF is related with the value of current flow from the

furnace coil. The value of furnace coil current is dependent on DC link current while

the CSI operates in resonance frequency as represented in previous sections. The DC

link current of CSI-IF is provided by the output DC voltage rectifier so the DC link

current is changed by varying rectifier output DC voltage. Therefore the control of

heating power can be performed by the rectifier of CSI-IF.


67

In high power CSI-IF, 12-pulse or higher pulse rectifiers are preferred

because of the lower demand for current harmonic content as in the base model CSI-

IF. The CSI-IF in the steel mill used as a base model in this thesis study has a series

connected 12-pulse fully controlled rectifier. The series connected 12-pulse fully

controlled rectifier is formed from two series connected 6-pulse rectifier which are

supplied from one of double secondary transformer as shown in Figure 4.13. The star

and delta connected secondaries have an inherent 6-phase which have 60° phase

difference.

6-Pulse Fully Controlled Rectifier

Double SecondaryTransformer

6-Pulse Fully Controlled Rectifier

Figure 4.13. Circuit Diagram of 12-Pulse Rectifier

The variable output of 12 pulse fully controlled rectifier is performed by

firing thyristor with delay/firing angle. When the thyristors are triggering at the zero

crossing point of each phase voltage signals, the firing angles of thyristors are equal

to zero and maximum DC voltage can be obtained from the output of the rectifier. If

the firing angle of rectifier is increased the voltage across the output of rectifier is

decreased. The mean value of the output DC voltage of rectifier is calculated as;


68

712

512

12sinmean mV V d

(4.31)

where

: firing angle

mV : input line to line voltage peak value of rectifier

From this equation meanV is equal to

1.977 cosmean mV V (4.32)

4.2. Power Quality Problems of CSI-IF

The CSI-IF in the steel mill used as a base model in this thesis study is a type

of AC to AC converter and formed from basically rectifier, DC link and inverter. As

introduced in Section 3.2, the harmonics and interharmonics created by the static

power converters can be derived from the equation;

1 1 21i of p m f p n f (4.33)

where

if : is the generated harmonics and interharmonics

1f : is the fundamental supply frequency

of : is the output frequency of the inverter

1p : is the pulse number of the rectifier

2p : is the pulse number of the inverter

m : are 0, 1, 2, 3 … (integers)

n : are 0, 1, 2, 3 … (integers)

The first part of this equation which contains 1p , m and 1f represents the

harmonics produced by the rectifiers. The second part of equation contains 2p , n


69

and of represents the effects of inverters and combination with first part gives the

interharmonics produced by power converter. If the rectifier and the inverter of

power converter can be decoupled perfectly with DC link, the second part of the

equation can be cancelled and only rectifier harmonics is seen in the current of power

converters. However if good decoupling cannot be provided between the rectifier and

the inverter of converter the interharmonics components are seen in the current of

power converter.

Generally when DC link of power converter is formed from only inductors,

the good isolation cannot be provided and inverter operating frequency affects the

input currents of the power converter. The coreless CSI-IF in the steel mill is an

example of such systems because the inductor is used in the DC link of power

converter of furnace. The CSI-IF in steel mill is connected to 50 Hz system voltage

and it has a 12 pulse rectifier. The inverter of this CSI-IF is a single phase H-bridge

CSI and operates between 150 Hz and 250 Hz. When the parameters are placed in

Eq. 4.33 if is obtained as;

12 1 50 2i of m nf (4.34)

When the values of variables m and n are increased in the Eq. 4.34, the amplitudes

of the harmonics and interharmonics decrease. Also, the amplitudes of the harmonics

and interharmonics decrease in the higher frequencies of of . By giving values of m

and n in Eq. 4.34 up to 2, the most dominant harmonics related with the operating

frequency of CSI-IF are shown in Figure 4.14. In this figure, the lines of most

dominant harmonics are colored with darker colors. In Figure 4.14, it is shown that

besides 11th

, 13th

, 23rd

and 25th

harmonics created by rectifier, the harmonic and

interharmonic frequencies at 1 2 of f and 1 4 of f are shown as dominant.


70

150

250

350

450

550

650

750

850

950

1050

1150

1250

50

1350

15

0

16

0

17

0

18

0

19

0

20

0

21

0

22

0

23

0

24

0

25

0

f1+2fo

-f1+2fo

f1+4fo

-f1+4fo

13f111f1

25f123f1

13f1-2fo

11f1-2fo

13f1+2fo11f1+2fo

25f1-2fo

23f1-2fo

CSI-IF Operating Frequency

fo

Ge

ne

rate

d H

arm

on

ics

by

CS

I-IF

f i

Figure 4.14. Generated Harmonics Related with Operating Frequency of CSI-IF

It is shown clearly in the Figure 4.14 that harmonic and interharmonic content

of furnace current is varying with the operating frequency of furnace and cover wide

harmonic and interharmonic spectrum. Moreover the high harmonic and

interharmonic content of CSI-IF current can cause serious voltage harmonics and

interharmonics especially in weak power systems. The voltage interharmonics cause

voltage fluctuations and flicker problems because of the interaction of non-

synchronous interharmonic components with the fundamental frequency (Dugan et

al., 1999).


71

4.3. Modeling of Current Source Induction Furnace

4.3.1. Power Circuit Parameters

The power circuit model of CSI-IF in PSCAD/EMTDC. shown in Figure 4.15

DCI

DCV

DCL

DCL

res-tankI

res-tankV

fur coilI

stray

L

fL

fC

fR

sysL

sysL

sysL

aI

bI

cI

aV

bV

cV

3 PHASE

DOUBLE SECONDARY

TRANSFORMER

12 MVA

31.5 kV/1.2 kV -1.2kV

Δ/ Δ-Y

3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

Figure 4.15. Power Circuit Model of CSI-IF

The CSI-IF is connected to 31.5kV busbar which has 335 MVA short circuit

power. The system impedance is calculated from Eq. 4.35 and found as 9.4 mH.

2

,

100

L L rms

sys

sc

VL

S

(4.35)

The power circuit parameters of CSI-IF is determined by using the coreless

melting CSI-IF in the steel mill. Except the furnace coil parameters, all other

parameters are taken from the CSI-IF in steel mill. The furnace coil inductance

values are calculated using the fixed capacitor values of resonant tank circuit and

operating frequency of furnace by using Eq. 4.18. The furnace coil resistance values

are determined by quality factor of parallel resonant tank circuit and calculated

furnace inductance values. According to Eq. 4.28, quality factor of parallel resonant

tank circuit is determined by using current measurements of DC link current and


72

furnace coil current. When the measurements of DC-link current and furnace coil

current are investigated, the quality factor of parallel resonant tank circuit has lower

values during charging period of furnace. When the furnace crucible is full, quality

factor of parallel resonant tank circuit reaches higher values. By using this

information about quality factor and furnace coil inductance values, the furnace

resistance values are determined by using Eq. 4.12. The power circuit parameters of

CSI-IF modeled in PSCAD/EMTDC are given in Table 4.1.

Table 4.1. Power Circuit Parameters of CSI-IF Model

Transformer 12 MVA, 31.5 kV/1.2 kV -1.2kV, Δ/ Δ-Y

VSC,1-2, VSC,1-3=12%, VSC,2-3=30%,

LDC 1 mH

LStray 12.5 µH

Cf 5.28 mF

Rf 0.011 – 2 mΩ

Lf 0.076 – 0.213 mH

In proposed model of CSI-IF, starting circuit is not used for CSI. Initial

voltage of furnace capacitor is given by an external voltage source at start-up of

furnace.

4.3.2. Controller of CSI-IF

The controller of CSI-IF applied in simulation study is formed from two parts

as DC link current controller and CSI controller. DC link current controller controls

the firing angles of thyristor in order to maintain constant DC link current value. The

DC link controller is formed from basic PI controller and firing pulse generator as

shown in Figure 4.16. The PI controller produces the firing angle according to error

signal between the set value and measurement value of DC link current. Firing pulse

generator generates appropriate firing angles for SCRs of 12-pulse rectifier by using

PLL. PLL gives the phase information of voltage of phase A. Firing angles of each


73

thyristor produced by comparing firing angle and the phase value of voltage by firing

pulse generator.

DCI

aV

PI

CONTROLLER

PLL

FIRING

PULSE

GENERATOR

12

DCI

aV

,DC setI

a

12-Pulse

Rectifier

-

+

Figure 4.16. Block Diagram of DC Link Controller of CSI-IF

By changing set value of DC link current, DC link current is changed by 12-

pulse rectifier so, the heating power control can be adjusted. Also at the starting of

CSI-IF, CSI cannot determine the resonant frequency immediately. Because of these

at the starting of furnace, the impedance of parallel resonant circuit is low. Until the

CSI of furnace catches the operation frequency of parallel resonant tank circuit, the

rectifier must decrease DC link voltage to prevent excessive current flow from the

power switches of CSI.

In order to control the CSI of IF, many controllers are proposed in literature

and also the most of these controllers are PLL based controllers. The main idea of

these PLL controllers is to keep the parallel resonant load in resonance with reducing

the phase shift between current and voltage across the terminals of parallel resonant

circuit to approximately zero by increasing or decreasing the CSI operating

frequency. The proposed controller of CSI-IF shown in Figure 4.17 is developed by

using the CSI controllers proposed by Khan et al. (2000), Peng et al., (1989) and

Ponwiangkum et al. (2007). These PLL based controller is mainly formed from


74

phase detector (PD), PI controller, voltage controlled oscillator (VCO) and switching

signal generator.

ZERO

CROSSING

DETECTOR

ZERO

CROSSING

DETECTOR

XORLOW

PASS

FILTERx -

+

PI

CONTROLLER

-

+ VOLTAGE

CONTROLLED

OSCILLATOR

COMPARATOR

res-tankV

res-tankI

res-tankI

res-tankV

set

,CSI i

CSI CSI

1Q

2Q

3Q

4Q

1Q

2Q

3Q

4Q

Figure 4.17. Block Diagram of CSI of CSI-IF

In the phase detector part, both current and voltage signals are formed as

square wave with the same zero crossing points as original signals by zero crossing

detectors. Then with using exclusive OR gate (XOR) and low pass filter, the phase

difference between the current and voltage waveforms of parallel resonant circuit

is determined. In order to convert the phase difference to radian the output of low

pass filter is multiplied with π. This phase difference is subtracted from phase

difference set value set to form error signal. In ideal conditions, set is taken 0 but in


75

order to provide the load commutation of CSI set must higher from 0. error goes

into the PI controller. The output of PI controller is summing with the initial CSI

angular frequency ,CSI i and CSI inverter operating angular frequency CSI is

calculated. ,CSI i is selected higher than operating values of CSI in order to enforce

the parallel resonant load in capacitive region at start-up CSI-IF. The angular

frequency CSI goes into VCO. VCO is formed from simple integrator and it is used

for generating the phase information CSI related to CSI . CSI is used to generate

switching signals of CSI by the help of the comparator. If CSI is higher than , 1Q

and 2Q are triggered and if CSI is lower than , 3Q and 4Q are triggered.

4.3.3. Simulation Results of CSI-IF

By using the operating frequency range and process conditions of CSI-IF in

steel mill, a simulation scenario is designed operate the furnace from no load to full

load by setting the furnace coil inductance and resistance values. Firstly furnace has

no charge and with starting of simulation furnace is started to charge up to full load

capacity. Then furnace runs for melting the charge completely and heating the charge

up to necessary process temperature. According to this scenario, furnace starts

operating at 150 Hz and operation frequency of furnace reaches up to 250 Hz related

to filling rate of furnace as shown in Figure 4.18.


76

50

100

200

250

300

150

f (Hz)

t

Charging and Melting Heating Melted Metal

Figure 4.18. Frequency Trend of Modeled Induction Furnace According To

Operating Scenario

Firstly the controller signals of CSI-IF are presented. In the Figure 4.19 the

trend of phase difference signal is given and in Figure 4.20 the trend of CSI

angular operating frequency CSI is given during the start-up of CSI-IF. It is shown

that in Figure 4.19, the phase difference between parallel resonant tank circuit

current and voltage across the terminals of parallel resonant tank circuit is high at the

beginning. After CSI gets down to the resonance frequency of tank circuit, the phase

difference reaches the set value 15 .


77


Load

Figure 4.20. Trend of CSI Angular Operating Frequency

The trend of phase difference and the trend of CSI angular operating

frequency CSI values taken during the entire simulation time are presented in Figure

4.21 and Figure 4.22. It is clearly shown that while the operating frequency of

furnace is changing, the CSI controller keeps the phase difference constant. In

other words, the CSI controller keeps the resonant tank circuit in resonant although

the furnace coil inductance and resistance are changing continuously. Moreover CSI

CSI_Controller : Graphs

sec 0.000 0.100 0.200 0.300 0.400 0.500 0.600 0.700 0.800 0.900 1.000 ...

...

...

0.00

0.20

0.40

0.60

0.80

1.00

1.20

1.40

1.60

1.80

Phase

Diffe

rence

(R

adia

n)

PhaseDifference


sec 0.000 0.100 0.200 0.300 0.400 0.500 0.600 0.700 0.800 0.900 1.000 ...

...

...

0.8k0.9k

1.0k1.1k1.2k1.3k

1.4k1.5k1.6k1.7k

1.8k1.9k2.0k2.1k

2.2k2.3k2.4k

CS

I O

pera

ting F

req (

Radia

n)

PI_ControlSignal


78

in the Figure 4.22 shows the same variation with the determined modeling operating

scenario. This shows that the controller of CSI reacts correctly to change of the

furnace coil parameter.


Load during Entire Simulation

Figure 4.22. Trend of CSI Angular Operating Frequency during Entire Simulation

In Figure 4.23 the phase of the operating frequency of CSI CSI and in Figure

4.24 gate pulses of SCRs in CSI during the start-up of CSI-IF is shown. It is


sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

0.00

0.20

0.40

0.60

0.80

1.00

1.20

1.40

1.60

1.80

Phase

Diffe

rence

(R

adia

n)

PhaseDifference


sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

0.8k0.9k

1.0k1.1k1.2k1.3k

1.4k1.5k1.6k1.7k

1.8k1.9k2.0k2.1k

2.2k2.3k2.4k

CS

I O

pera

ting F

req (

Radia

n)

PI_ControlSignal


79

obviously seen that the frequency of CSI signal is higher and the firing pulses of

SCRs are more frequently at the start-up of CSI because of the high value of CSI at

the start-up of CSI-IF.

Figure 4.23. Phase of the Operating Frequency of CSI

Figure 4.24. Thyristor Gate Pulses of CSI


sec 0.0000 0.0100 0.0200 0.0300 0.0400 0.0500 0.0600 0.0700 0.0800 0.0900 0.1000 ...

...

...

0.00

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

5.00

5.50

6.00

6.50

7.00

Ref

Phase

of

CS

I (R

adia

n)

Phase

CSI-IF Controller : Graphs

sec 0.0000 0.0100 0.0200 0.0300 0.0400 0.0500 0.0600 0.0700 0.0800 0.0900 0.1000 ...

...

...

0.0

1.20

Q1-Q

2 G

ate

Puls

es

Q12

0.0

1.20

Q3-Q

4 G

ate

Puls

es

Q34


80

The current waveform of parallel resonant tank circuit and the voltage

waveform between the terminals of parallel resonant tank circuit are shown in Figure

4.25. It is seen that while the current res-tankI has square wave shape, the voltage

res-tankV has sinusoidal waveform and alsores-tankI leads res-tankV .with angle .

Figure 4.25. Current and Voltage Waveform between Terminals of Parallel Resonant

Tank Circuit

In Figure 4.26 the current waveform of furnace coil fur-coilI is given. It is seen

that, fur-coilI is higher than the times of res-tankI because of the resonance.

Figure 4.26. Current Waveform of Furnace Coil

CSI-IF : Graphs

sec 3.0000 3.0020 3.0040 3.0060 3.0080 3.0100 3.0120 3.0140 3.0160 3.0180 3.0200 ...

...

...

-6.0

-5.0

-4.0

-3.0

-2.0

-1.0

0.0

1.0

2.0

3.0

4.0

5.0

6.0

Reso

nant

Tank

Circu

it I

&V

(kA

,kV

)

Ires-tank Vres-tank

CSI-IF : Graphs

sec 3.0000 3.0020 3.0040 3.0060 3.0080 3.0100 3.0120 3.0140 3.0160 3.0180 3.0200 ...

...

...

-35.0

-30.0

-25.0

-20.0

-15.0

-10.0

-5.0

0.0

5.0

10.0

15.0

20.0

25.0

30.0

35.0

Furn

ace

Coil

Curr

ent

(kA

)

Ifur-coil


81

The trend of firing angle of rectifier during the start-up of CSI-IF is shown

in Figure 4.27. It is seen that at the start-up the firing angle has high delay angle

because CSI cannot reach directly to the resonant frequency of parallel resonant load

at start up. When the parallel resonant load is not operating at the resonant

frequency, it shows low impedance. Because of these, the controller of rectifier

increases the firing angle of SCRs and decreases the DC link voltage DCV in order to

limit the DC link current DCI and prevent the excessive current flow through the CSI

as shown in Figure 4.28 and 4.29.

Figure 4.27. Trend of Firing Angle of 12-Pulse Rectifier

Main : Graphs

sec 0.000 0.100 0.200 0.300 0.400 0.500 0.600 0.700 0.800 0.900 1.000 ...

...

...

0

10

20

30

40

50

60

70

80

90

100

12 P

uls

e R

ec

Firin

g

Ang (

Degre

e)

Alpha


82

Figure 4.28. Trend of DC Link Voltage of CSI-IF

Figure 4.29. Trend of DC Link Current of CSI-IF

The trend of DCV and DCI values taken during the entire simulation time are

presented in Figure 4.30 and Figure 4.31.

CSI-IF : Graphs

sec 0.000 0.100 0.200 0.300 0.400 0.500 0.600 0.700 0.800 0.900 1.000 ...

...

...

0.00

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

CS

I-IF

DC

Lin

k V

oltage (

kV)

Vdc

CSI-IF : Graphs

sec 0.000 0.100 0.200 0.300 0.400 0.500 0.600 0.700 0.800 0.900 1.000 ...

...

...

0.00

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

5.00

CS

I-IF

DC

Lin

k C

urr

ent

(kA

)

Idc


83

Figure 4.30. Trend of DC Link Voltage of CSI-IF during Entire Simulation

Figure 4.31. Trend of DC Link Current of CSI-IF during Entire Simulation

In Figure 4.32, the ripple on the DCI is given. It is seen that there is high

ripple content on the DCI which resulted from CSI and causes interharmonic

distortion in the supply current of CSI-IF. The ripple frequency on the DCI is equal to

the two times of the CSI operating frequency.

CSI-IF : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

0.00

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

CS

I-IF

DC

Lin

k V

oltage (

kV)

Vdc

CSI-IF : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

-1.00

-0.50

0.00

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

5.00

CS

I-IF

DC

Lin

k C

urr

ent

(kA

)

Idc


84

Figure 4.32. Ripple on DC Link Current of CSI-IF

The waveforms of current drawn by CSI-IF and busbar voltage of CSI-IF is

given in Figure 4.33 and Figure 4.34. It is obviously seen that current waveform is

highly distorted and voltage form is also distorted because of the voltage drop on the

system impedance.

Figure 4.33. Supply Current Waveform of CSI-IF

CSI-IF : Graphs

sec 3.0000 3.0020 3.0040 3.0060 3.0080 3.0100 3.0120 3.0140 3.0160 3.0180 3.0200 ...

...

...

2.00

2.10

2.20

2.30

2.40

2.50

2.60

2.70

2.80

2.90

3.00

3.10

CS

I-IF

DC

Lin

k C

urr

ent

(kA

)

Idc

CSI-IF : Graphs

sec 3.000 3.010 3.020 3.030 3.040 3.050 3.060 3.070 3.080 3.090 3.100 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Supply

Curr

ents

(kA

)

Ia Ib Ic


85

Figure 4.34. Supply Voltage Waveform of CSI-IF

The trend of active and reactive power of CSI-IF is shown in Figure 4.35. The

high reactive power is drawn by CSI-IF at the start-up of CSI-IF because of the high

firing angles of 12-pulse rectifier. However, it is shown that CSI-IF has low and slow

varying reactive power demand.

Figure 4.35. Trend of Active and Reactive Power of CSI-IF

In Figure 4.36, the angular operating frequency CSI at the 1st second of the

simulation is shown. At this time the furnace approximately operates at 175 Hz. In

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-40

-30

-20

-10

0

10

20

30

40

CS

I-IF

Supply

Voltages

(kV

)

Va Vb Vc

CSI-IF : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

-1.0

0.0

1.0

2.0

3.0

4.0

5.0

6.0

7.0

8.0

9.0

10.0

CS

I-IF

P&

Q (

MW

,MV

Ar)

P_CSI-IF Q_CSI-IF


86

Figure 4.37, the most dominant harmonics and interharmonics of CSI-IF calculated

for 175 Hz with Eq. 4.33 is presented and in Figure 4.38(a) and 4.38(b) the harmonic

spectrum of current drawn by CSI-IF is shown. The harmonic spectrum shows the

harmonics with steps of 10 Hz in order to indicate the interharmonics. In Figure

4.38(a) lower order harmonics and interharmonics are presented with larger scale and

in Figure 4.38(b) higher order harmonics and interharmonics are presented with

lower scale. When the Figure 4.37 is compared to harmonic spectrum of furnace

current, it is obviously seen that CSI-IF model produce the all calculated harmonics

and interharmonics.

Average ωo ≈ 1100 rad

fo ≈ 175 Hz

Figure 4.36. Angular Operating Frequency at the 1

st Second of the Simulation


87

150

250

350

450

550

650

750

850

950

1050

1150

1250

50

1350

15

0

16

0

17

0

18

0

19

0

20

0

21

0

22

0

23

0

24

0

25

0

f1+2fo

-f1+2fo

f1+4fo

-f1+4fo

13f111f1

25f123f1

13f1-2fo

11f1-2fo

13f1+2fo11f1+2fo

25f1-2fo

23f1-2fo


fo

Ge

ne

rati

ng

Ha

rmo

nic

s b

y C

SI-

IF

f i

f1+6fo -f1+6fo

Figure 4.37. Generated Harmonics When CSI-IF Operates at 175 Hz


88

Fig

ure

4.3

8. H

arm

onic

Spec

trum

of

CS

I-IF

Curr

ent

when

CS

I-IF

oper

ates

at

175 H

z

(a)

Low

er O

rder

Har

mon

ics

wit

h L

arge

Sca

le

(b)

Hig

her

Ord

er H

arm

onic

s w

ith L

ow

er S

cale


89

Another example is given at the 5th

second of the simulation. It is shown in

Figure 4.39, CSI-IF operates at 235 Hz at this time. In Figure 4.40, the calculated

harmonics and interharmonics for 235 Hz is given and in Figure 4.41 the harmonic

spectrum of current of CSI-IF is presented. Again when the figure 4.40 is compared

to harmonic spectrum of furnace current, it is obviously seen that CSI-IF model

produce the all calculated harmonics and interharmonics.

Average ωo ≈ 1490 rad

fo ≈ 235 Hz

Figure 4.39. Angular Operating Frequency at the 5

th Second of the Simulation


90

150

250

350

450

550

650

750

850

950

1050

1150

1250

50

1350

15

0

16

0

17

0

18

0

19

0

20

0

21

0

22

0

23

0

24

0

25

0

f1+2fo

-f1+2fo

f1+4fo

-f1+4fo

13f111f1

25f123f1

13f1-2fo

11f1-2fo

13f1+2fo11f1+2fo

25f1-2fo

23f1-2fo


fo

Ge

ne

rati

ng

Ha

rmo

nic

s b

y C

SI-

IF

f i

Figure 4.40. Generated Harmonics When CSI-IF Operates at 235 Hz


91

Fig

ure

4.4

1. H

arm

onic

Spec

trum

of

CS

I-IF

Curr

ent

when

CS

I-IF

oper

ates

at

235 H

z

(a)

Low

er O

rder

Har

mon

ics

wit

h L

arge

Sca

le

(b)

Hig

her

Ord

er H

arm

onic

s w

ith L

ow

er S

cale


92

4.3.4. Comparison of Simulation Results with Real Power Quality

Measurements of Induction Furnace

In this section, harmonics and interharmonics simulation results of CSI-IF

shown in previous section is compared to the real power quality measurements of

CSI-IF in the steel mill. These measurements are taken by IEC 61000-4-30 Class A

compatible measurement device. The harmonics and interharmonic measurements

are recorded with averaging measurements in every 3 seconds.

In Figure 4.42 the harmonic spectrum of CSI-IF during one casting time and

in Figure 4.43 the interharmonic spectrum during one casting time are given. The

interharmonics are calculated with steps of 5 Hz according to IEC 61000-4-7. The nth

interharmonic indicates that the interharmonic frequencies between nth

and (n+1)th

harmonics. In the simulation results, it is presented that the CSI-IF creates

dominantly the harmonics and interharmonics in the range of 1 2 of f and 1 4 of f

in addition to 11th

, 13th

, 23rd

and 25th

harmonics. The CSI-IF in the steel mill

approximately operates between 150 Hz and 250 Hz so 1 2 of f harmonics and

interharmonics are created between 250 Hz and 550 Hz and, 1 4 of f harmonics and

interharmonics are created between 650 Hz and 1050 Hz. The same characteristic of

CSI-IF shown in simulation results is obviously seen in the real harmonic and

interharmonic measurements of CSI-IF in the steel mill shown in both Figure 4.42

and 4.43.


93

Figure 4.42. Harmonic Spectrum Obtained by the PQ Measurements in the Steel Mill

Figure 4.43. Interharmonic Spectrum Obtained by the PQ Measurements in the Steel

Mill


94

5. MODELING AND ANALYSIS OF PQ COMPENSATION FOR CSI-IF

Adnan TAN

95

5. MODELING AND ANALYSIS OF PQ COMPENSATION SYSTEMS FOR

CSI-IF

In the previous section the power quality problems of high power melting

CSI-IF is presented in detail. It is presented that CSI-IF produces time varying

harmonic and interharmonic currents in wide frequency spectrum range. Moreover

these distorted currents also causes voltage distortion at the busbar which CSI-IF is

connected. Besides harmonic and interharmonic problems, reactive power demand of

CSI-IF must be compensated. Fortunately the reactive power demand of CSI-IF has

slow varying characteristic and can be easily handled with traditional methods. In

this section passive and active filtering systems are investigated and the most

effective compensation system is determined for the power quality problems of high

power CSI-IF.

5.1. Passive Filters

In order to solve the power quality problems of high power CSI-IF, single

tuned filter, C-type filter and broad-band filter are designed and the performances of

these filters are presented.

5.1.1. Single Tuned Passive Filters

Single tuned passive filters are most commonly applied type shunt passive

filters. As shown in Figure 5.1 single tuned passive filter can be modeled as series

connected inductor, resistor and capacitor. The capacitor and inductor are tuned near

a harmonic frequency in order to bypass that harmonic providing low impedance.

The capacitor of filter also provides reactive power compensation and the capacitor

value is selected according to the reactive power demand of system. The resistor

represents the internal resistance of inductor or an external resistor which can be used

additionally. The resistance value of the filter affects the sharpness and bandwidth of

filter. Because single tuned filters are filtering only harmonics in the tuned


Adnan TAN

96

frequency of filter, one single tuned filter may not be adequate to filter effectively all

the troublesome harmonics (Das, 2004; Nassif et al., 2009).

STFL

,STF YC

STFR

Figure 5.1. Single Tuned Passive Filter

Although single tuned filter acts as a very low impedance at the tuned

frequency, it always creates parallel resonance with system impedance and shows

quite high impedance at a frequency lower than the tuned frequency of filter as

shown in Figure 5.2 (Gonzalez et al., 1987).

Filter +Network

Impedance

Only Network

Impedance

Tuning Frequency

of Filter

Figure 5.2. Impedance Frequency Curve of Single Tuned Filter


Adnan TAN

97

The single tuned passive filter design can be performed by using Eq. 5.1, Eq.

5.2 and Eq. 5.3. In order to design passive filters, firstly the harmonic analysis of

system is performed and harmonics which are exceeding the limits and cause

disturbance in the system is determined. Secondly the reactive power requirement of

system is determined. After harmonic distortion and reactive power requirements of

system are determined, the filter tuning frequencies and reactive power are

determined. The single tuned filters are commonly not tuned exactly to the frequency

it is intended to suppress. The filter capacitors and inductors have tolerances. Besides

these, aging and temperature effects alter the filter inductor and capacitor values

(Das, 2004). These changes in the capacitor and the inductor values of filter can

cause to shift the resonance frequency above the harmonics in the system. This

causes the amplification of harmonics in the power system. Because of these single

tuned filters are tuned slightly below the harmonic frequencies. The reactive power

requirements of system are divided to filters according to amplitudes and frequencies

of harmonics. The lower frequency harmonics are generally higher amplitudes. In

order to filter these harmonics effectively, the filters tuned to lower frequencies

shows lower impedance to these harmonics. It can be achieved by increasing

capacitor value or in other words, the reactive power of filter. The capacitor and

inductor values of single tuned filter are calculated by using Eq. 5.1 and Eq. 5.2

according to tuning frequency and reactive power of filter. The QF of filter

determines the sharpness of filter and resistance value of filter is determined by Eq.

5.3. The quality factors of single tuned filters used in power systems are generally in

the range of 20-100.

,

1

2STF

STF STF Y

fL C

(5.1)

,

2

STF STF Y

LSTF

L C

VQ

X X

(5.2)


Adnan TAN

98

,

STF

STF Y

STF

STF

LC

QFR

(5.3)

If a harmonic compensation system formed from single tuned passive filters

is used for CSI-IF, tuning frequencies of passive filters must be selected correctly.

The modeled CSI-IF has non varying reactive power demand so 2 MVAr reactive

power compensation system is adequate for compensating reactive power demand of

CSI-IF. If a passive filtering system formed from 500 MVAr single tuned filters

which are tuned at 5th

, 7th

, 11th

and 13th

harmonics is designed for compensating the

harmonics and reactive power of CSI-IF, although this compensation system has

good filtering performance in tuning frequencies, this compensation system does not

work well and cause serious resonance problems. As shown in Figure 5.3, the single

tuned filters cause parallel resonance below the tuning frequencies of each branch.

The modeled CSI-IF generates the most dominant time varying harmonics and

interharmonics between 250 Hz and 650 Hz. Because of these this filtering system

cause parallel resonance problem which creates voltage distortion at PCC and causes

excessive harmonic current flow from both single tuned filters and power system.

Therefore such a passive filtering system is not suitable for CSI-IF.

Filters +Network

Impedance

Only Network

Impedance

Figure 5.3. Impedance-Frequency Curve of 5

th, 7

th, 11

th and 13

th Harmonic Single

Tuned Filters


Adnan TAN

99

The solution of avoiding the parallel resonance is to tune the passive filters

below frequencies of harmonics and interharmonics produced by the modeled CSI-

IF. When the simulation results of the modeled CSI-IF in Section 4 is investigated, it

is shown that CSI-IF produces harmonics and interharmonics in very wide spectrum

range. However below the 150 Hz, the magnitudes of harmonics and interharmonics

are inconsiderable, so a single passive filter tuned below the 150 Hz does not cause

resonant problems with power system. Because of these, 2 MVAr single tuned

passive filter tuned at 135 Hz is designed for CSI-IF. The design parameters of single

tuned filter are given in Table 5.1.

Table 5.1. Power Circuit Parameters of Single Tuned Passive Filter Model

Single Tuned Passive Filter

STFQ = 2MVAr , STFf =135 Hz, STFQF =50

STFL 251mH

,STFC 1.845µF

STFR 4.26Ω

The single line diagram of CSI-IF with passive filters and equivalent circuit

diagram for harmonic frequencies is shown in Figure 5.4. In harmonic frequencies

the 50 Hz voltage source acts as short circuited and CSI-IF acts as current source.

sysL

STFR

,STF YC

STFL

CSI-IF

sysL

CSI-IF

,CSI IF harI

Filter SystemZ(a) (b)

STFR

,STF YC

STFL

Figure 5.4. Single Line Diagram and Equivalent Circuit of CSI-IF with Single Tuned

Passive Filter


Adnan TAN

100

The impedance-frequency diagram of 135 Hz single tuned filter with power

system is shown in Figure 5.5. It is shown that the 135 Hz single tuned filter does not

create parallel resonance with system in the dominant harmonic frequencies of CSI-

IF. However, the impedance of this passive filter is slightly below the impedance of

power system in frequencies of dominant harmonics and interharmonics of CSI-IF so

this passive filters has very little filtering effect on harmonics and interharmonics of

CSI-IF and this filtering system is only suitable for reactive power compensation.

Filter +Network

Impedance

Only Network

Impedance

Figure 5.5. Impedance-Frequency Curve of Single Tuned Filter Designed for CSI-IF

This filtering characteristic of 2 MVAr passive filter tuned at 135 Hz is

shown in the PSCAD/EMTDC simulation program. The 2MVAr passive filter is

connected to the input of CSI-IF as shown in Figure 5.6.


Adnan TAN

101

sysL

sysL

sysL

,a sourceI

,b sourceI

,c sourceI

aV

bV

cV

3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

,a CSI IFI

,b CSI IFI

,c CSI IFI

,a STFI

,b STFI

,c STFI

CSI-IF

STFR

,STFC

STFL

STFR

STFL

STFR

STFL

,STFC

,STFC

Figure 5.6. Power Circuit Model of Single Tuned Filter

The waveforms of CSI-IF currents, source currents and single tuned filter

currents are shown in Figure 5.7, Figure 5.8 and Figure 5.9 respectively. It is shown

that there is nearly no change between the waveforms of CSI-IF currents and source

currents. Only the phase of source current is shifted by the effect of reactive power

compensation effect of single tuned filters.


Adnan TAN

102

Figure 5.7. CSI-IF Current Waveform in Single Tuned Filter Simulation

Figure 5.8. Source Current Waveform in Single Tuned Filter Simulation

CSI-IF : Graphs

sec 1.000 1.010 1.020 1.030 1.040 1.050 1.060 1.070 1.080 1.090 1.100 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Curr

ents

(kA

)

Ia_CSI-IF Ib_CSI-IF Ic_CSI-IF

Source : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

Supply

Curr

ent

(kA

)

Ia_source Ib_source Ic_source


Adnan TAN

103

Figure 5.9. Single Tuned Filter Current Waveform in Single Tuned Filter Simulation

The reactive powers drawn from source, CSI-IF and single tuned filters are

shown in Figure 5.10. It is shown that 2MVAr reactive power is supplied by single

tuned filters and reactive power drawn from the source decreases so by the help of

single tuned filters, reactive power compensation requirements of CSI-IF is provided

and reactive power demand of system is kept under the limits.

Figure 5.10. Reactive Powers Drawn From source, Drawn By CSI-IF and Drawn by

Single Tuned Filter in Single Tuned Filter Simulation

STF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.080

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

0.080

ST

F C

urr

ents

(kA

)

Ia_STF Ib_STF Ic_STF

SYS,CSI-IF,STF : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

-3.0

-2.0

-1.0

0.0

1.0

2.0

3.0

4.0

5.0

6.0

7.0

8.0

9.0

10.0

React

ive P

ow

ers

(M

VA

r)

Q_sys Q_CSI-IF Q_STF


Adnan TAN

104

The harmonic spectrums of CSI-IF currents and source currents is given in

Figure 5.11 and Figure 5.12 while CSI-IF is operating at 175 Hz. The harmonic

spectrums in Figure 5.11 and Figure 5.12 show the harmonics with steps of 10 Hz in

order to indicate the interharmonics. In Figure 5.11 lower order harmonics and

interharmonics are presented with larger scale and in Figure 5.12 higher order

harmonics and interharmonics are presented with lower scale. When the harmonic

spectrums of CSI-IF currents and source currents are investigated, it shown that

single tuned filters has very little effect on filtering harmonics and interharmonics.


Adnan TAN

105

Fig

ure

5.1

1.

Har

monic

Spec

trum

of

Low

er O

rder

Har

monic

s of

CS

I-IF

Curr

ent

and

Sourc

e C

urr

ent

in S

ingle

Tuned

Fil

ter

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

106

Fig

ure

5.1

2.

Har

monic

Spec

trum

of

Hig

her

Ord

er H

arm

onic

s of

CS

I-IF

Curr

ent

and S

ourc

e C

urr

ent

in S

ingle

Tuned

Fil

ter

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

107

5.1.2. C - Type Passive Filters

The C-type filters are generally used for attenuation of low order harmonics

and interharmonics created by HVDC transmission systems, EAFs and

cycloconverters (Bodger et al., 1990; Gerçek et al., 2011; Aravena et al., 2009). The

power circuit topology of C-type filter is shown in Figure 5.13. The C-type filters

show impedance and filtering characteristics between the 2nd

and 3rd

order high pass

filters. C-type filter is preferred because of the advantage in low losses in

fundamental frequency of power system. The series ,1CtypeL and

,1CtypeC in parallel with

the CtypeR is tuned to the fundamental frequency of power system.

CtypeR is, therefore,

bypassed by the zero impedance branch formed by the tuned ,1CtypeL and

,1CtypeC at

fundamental frequency. The filter thus behaves as a capacitor at the fundamental

frequency. There is little current flowing through the resistor and the loss is

minimized. As frequency increases, ,1CtypeL becomes resonating with

,1 ,2Ctype CtypeC C ,

what makes the filter behave as a single-tuned filter with a damping resistor. At high

frequencies, ,1CtypeL becomes large, and the current will flow through the resistive

branch, resulting in a performance similar to that of the first-order filter (Nassif et al.,

2009; Xiao et al., 2004).

,1CtypeL

,2CtypeC

CtypeR

,1CtypeC

Figure 5.13. Power Circuit Topology of C-Type Filter


Adnan TAN

108

The values of elements in the C-type filter can be designed by the following

equations. The series ,1CtypeL and

,1CtypeC are tuned to the system fundamental

frequency 1f so that;

1

,1 ,1

1

2 Ctype Ctype

fL C

(5.4)

The damping frequency of filter is commonly selected between the 2nd

and 6th

harmonics and it is determined as;

,1 ,2

,1

,1 ,2

1

.2

Ctype

Ctype Ctype

Ctype

Ctype Ctype

fC C

LC C

(5.5)

By using Eq. 5.4 and Eq. 5.5, the following relation between,1CtypeC and

,2CtypeC is

extracted;

2

,1 ,2

1

1Ctype

Ctype Ctype

fC C

f

(5.6)

The value ,2CtypeC is selected for the desired reactive power rating of the filter and it

must be large enough for the filter to be effective. ,1CtypeC is determined by using Eq.

5.6 and ,1CtypeL is determined by using Eq. 5.4 with the calculated

,1CtypeC value. The

damping resistance CtypeR is obtained by designating the desired quality factor

CtypeQF

for the filter at designed damping frequencyCtypef . For the filter to be effective, the

value CtypeQF should be selected below 5.

CtypeR is calculated as (Bodger et al., 1990);


Adnan TAN

109

,12 Ctype Ctype

Ctype

Ctype

f LR

QF

(5.7)

Using these equations a C-type filter is designed in order to compensate

power quality problems of modeled CSI-IF. The parameters of C-type filter designed

for CSI-IF is given in Table 5.2.

Table 5.2. Power Circuit Parameters of C-Type Filter Model

C-Type Passive Filter

CtypeQ = 2MVAr , Ctypef =100 Hz,

CtypeQF =1

,1CtypeL 526 mH

,1CtypeC 19.24 µF

,2CtypeC 6.415 µF

CtypeR 330 Ω

The single line diagram of C-type filters with CSI-IF and equivalent circuit



sysL

CSI-IF

sysL

CSI-IF

,CSI IF harI

Filter SystemZ

(a) (b)

,1CtypeL

,2CtypeC

CtypeR

,1CtypeC

,1CtypeL

,2CtypeC

CtypeR

,1CtypeC

Figure 5.14. Single Line Diagram and Equivalent Circuit of CSI-IF with C-Type

Filter


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110

The impedance-frequency diagram of C-type filter with power system is

shown in Figure 5.15. It is shown that the C-type filter does not create any parallel

resonant with system because its quality factor is selected very low in order to

increase the damping effect but, unfortunately the impedance of C-type passive filter

with power system shows the same characteristics with power system impedance

below 900 Hz and above 900 Hz, the impedance of C-type passive filter with power

system gets slightly lower than the system impedance. Due to this impedance-

frequency curve of designed C-type filter, C-type filter has no filtering effect on

dominant harmonics and interharmonics of CSI-IF between 250 Hz and 650 Hz. This

filtering system only performes the reactive power requirements of CSI-IF.

2MVAr C-Type +Network

Impedance

Network Impedance

Figure 5.15. Impedance-Frequency Curve of C-Type Filter Designed for CSI-IF

If the damping of C-type filter is chosen at lower levels by selecting higher

CtypeQF value, the C-type filter designed for the modeled CSI-IF shows high

impedance with system impedance and causes amplification below 900 Hz as shown

in Figure 5.16. In this condition, all dominant harmonics and interharmonics of CSI-

IF is amplified and creates serious voltage distortion in PCC.


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111

As all the damped passive filters, C-type filters need high fundamental MVAr

ratings in order to achieve effective harmonic filtering. In Figure 5.17, the

impedance-frequency curves of C-type filters with power system are shown. In this

figure, C-type filters have different MVAr ratings and their quality factors are equal

to 1. It is clearly shown that the filtering performance is increased with increasing

reactive power rating of filter.


Impedance

Q.F.=1


Impedance

Q.F.=3


Impedance

Q.F.=5

Network Impedance

Figure 5.16. Impedance-Frequency Curve of C-Type Filter with Different Quality

Factor


Adnan TAN

112


Impedance

Network Impedance


Impedance


Impedance


Impedance


Impedance

Figure 5.17. Impedance-Frequency Curve of C-Type Filter with Different Reactive

Power Ratings

The filtering characteristic of 2 MVAr C-type filter tuned at 100 Hz is shown

in the PSCAD/EMTDC simulation program. The 2MVAr C type filter is connected

to the input of CSI-IF as shown in Figure 5.18.


Adnan TAN

113

sysL

sysL

sysL

,a sourceI

,b sourceI

,c sourceI

aV

bV

cV

3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

,a CSI IFI

,b CSI IFI

,c CSI IFI

,a C typeI

,b C typeI

,c C typeI

CSI-IF

,1CtypeL

,2CtypeC

CtypeR

,1CtypeC

,1CtypeL

,2CtypeC

CtypeR

,1CtypeC

,1CtypeL

,2CtypeC

CtypeR

,1CtypeC

Figure 5.18. Power Circuit Model of C-Type Filter

The waveforms of CSI-IF currents, source currents and C-type filter currents

are shown in Figure 5.19, Figure 5.20 and Figure 5.21 respectively. It is shown that

there is nearly no change in between the waveforms of CSI-IF currents and source

currents. Only the phase of source current is shifted by the effect of reactive power

compensation effect of C-type filters.


Adnan TAN

114

Figure 5.19. CSI-IF Current Waveform in C-Type Filter Simulation

Figure 5.20. Source Current Waveform in C-Type Filter Simulation

When C-type filter current waveforms are investigated in Figure 5.21, the

filter has distorted current waveform. It shows that the C-type filter filters a part of

high frequency harmonics and interharmonics currents because of the lower

impedance of C-type filter than the system impedance above 900 Hz as shown in

impedance-frequency curve of designed filter.

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Curr

ents

(kA

)


Source : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

Supply

Curr

ent

(kA

)



Adnan TAN

115

Figure 5.21. C-Type Filter Current Waveform in C-Type Filter Simulation

The reactive powers drawn from source, CSI-IF and C-type filter are shown

in Figure 5.22. It is shown that 2MVAr reactive power is supplied by C-type filters

and reactive power drawn from the source is decreased. By the help of C-type filter,

reactive power compensation requirements of CSI-IF is provided and reactive power

demand of system is kept under the limits.

Figure 5.22.Reactive Powers Drawn From source, Drawn By CSI-IF and Drawn by

C-Type Filter in C-Type Filter Simulation

C-type : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.080

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

0.080

ST

F C

urr

ents

(kA

)

Ia_C-type Ib_C-type Ic_C-type

SYS,CSI-IF,C-type : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

-3.0

-2.0

-1.0

0.0

1.0

2.0

3.0

4.0

5.0

6.0

7.0

8.0

9.0

10.0

11.0

12.0

React

ive P

ow

ers

(M

VA

r)

Q_sys Q_CSI-IF Q_C-type


Adnan TAN

116

The harmonic spectrums of CSI-IF currents and source currents is given in

Figure 5.23 and 5.24 while CSI-IF is operating at 175 Hz. The harmonic spectrums

show the harmonics with steps of 10 Hz in order to indicate the interharmonics. In

Figure 5.23 lower order harmonics and interharmonics are presented with larger

scale and in Figure 5.24 higher order harmonics and interharmonics are presented

with lower scale. When the harmonic spectrums of CSI-IF currents and source

currents are investigated, it is shown that C-type filters has no effect on filtering

harmonics and interharmonics below 900 Hz. C-type filter shows very little filtering

performance to harmonics and interharmonics above 900 Hz.


Adnan TAN

117

Fig

ure

5.2

3. H

arm

onic

Spec

trum

of

Low

er O

rder

Har

monic

s of

CS

I-IF

Cu

rren

t an

d S

ourc

e C

urr

ent

in C

-Type

Fil

ter

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

118

Fig

ure

5.2

4. H

arm

onic

Spec

trum

of

Hig

her

Ord

er H

arm

onic

s of

CS

I-IF

Cu

rren

t an

d S

ourc

e C

urr

ent

in C

-Type

Fil

ter

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

119

5.1.3. Broad - Band Passive Filters

Unlike the shunt and series filters that have a narrow band of harmonic

suppression, broadband filters have a wider range of harmonics suppression property.

Broadband filters employ a combination of the two passive techniques, with a high

series impedance to block the undesired current harmonics (from flowing through the

grid) and a low shunt impedance path to divert their flow through the shunt filter.

They are tuned to a low cut off frequency such that only fundamental component will

pass from the input to the output (Win, 2008). In Figure 5.25, the most known broad-

band filter topologies; LC and LLCL are shown. These types of broad-band filters

are available in market especially for filtering harmonics caused by low voltage

ASDs. However, any high voltage application is not encountered in literature.

,BBF sL

,BBF Y sC

,BBF sR

,i BBFL

,o BBFL

BBFL

BBFC

(a) (b)

Figure 5.25. Broad Band Passive Filter Topologies

(a) LC Type Broad Band Passive Filter

(b) LLCL Type Broad Band Passive Filter

The LC type broadband filter is shown in Figure 5.25(a) and design of this

type filter is proposed by Swamy (1995). This type broad-band filter involves a

simple structure with relatively large components. Also, in particular at no-load the

filter output voltage becomes excessive. In order to prevent high voltage a buck

transformer is used instead of inductor in the filter. Moreover, the power factor is

largely leading at all load conditions to achieve effective harmonic filtering. Thus,


Adnan TAN

120

the filter does not yield overall a satisfactory performance (Swamy, 2005; Zubi et al.,

2010).

In distribution system applications, the effect of broadband filters can be

obtained by installing a capacitor bank on the low-voltage side of a substation

transformer. The size of the capacitor bank would have to be so selected to provide

the desired cutoff frequency when combined with the transformer leakage inductance

and the system impedance. It is then capable of preventing harmonics above the

cutoff frequency from penetrating the high-voltage side of the transformer. Since the

cutoff frequency can be sometimes quite low, the size of the capacitor bank may be

fairly large. This will result in a significant voltage rise. To keep the voltage of

system in rated values, load tap changer of transformer can be used (Dugan et al.,

1996). Again, because of the large capacitor values, over compensation problem

occurs in the system. Other restriction of such a broad-band passive filter occurs in

multi-feeder systems placed at the low voltage side of substation transformer. In

multi-feeder systems, using of large value capacitors under the substation

transformers can cause not estimated resonance problems with the nonlinear loads

connected under distribution transformers which are placed in feeders under the low

voltage side of substation transformer.

In the LLCL type broad-band filters, the tradeoff between the over voltage,

over reactive power compensation and filtering performance can be solved more

easily. As shown in Fig. 5.25(b), the LLCL broad-band filter is created by splitting

the L filter of the simple LC broadband filter into two elements as ,i BBFL and ,BBF sL ,

the filter input-to-output behavior is altered. Further, a 3-5% smoothing reactor

,o BBFL is inserted between the rectifier terminals and the filter output terminals.

Filter capacitors ,BBF Y sC are usually delta connected. ,BBF sR is represents the

internal resistance of inductor or an external damping resistor which can be used

additionally. ,BBF sR changes the damping characteristics of filter (Zubi et al., 2010).

There is limited information about the design and determining optimal

parameters of LLCL type broad-band passive filters in literature. In this thesis work,

by the help of the iteration based design methodology presented by Havva (2005)


Adnan TAN

121

and PSCAD/EMTDC simulation program optimal values of elements in the LLCL

type broad-band filter is determined for solving the power quality problems of

modeled CSI-IF.

The initial values of LLCL filter can be determined by using following

equations. The parallel resonance frequency of filter is determined as;

,

, , ,

1

2p BBF

i BBF BBF s BBF Y s

fL L C

(5.8)

The series resonance frequency of filter is determined as;

,

, ,

1

2s BBF

BBF s BBF Y s

fL C

(5.9)

The value ,BBF Y sC is selected for the desired reactive power rating of the filter. ,i BBFL

and ,BBF sL are determined using Eq. 5.8 and Eq. 5.9 with using calculated ,BBF Y sC

value. The parallel resonance frequency ,p BBFf of filter in Eq. 5.8 is selected near 2

nd

or 3rd

harmonic frequencies and the series resonance frequency,s BBFf of filter in Eq.

5.9 is selected slightly below the most dominant harmonic frequency of nonlinear

load. For CSI-IF the series resonance frequency of filter is selected below the

dominant frequencies of CSI-IF in order to prevent resonances. The damping

resistance ,BBF sR is obtained by designating the desired quality factor BBFQF for the

filter at designed parallel resonance frequency,p BBFf . ,BBF sR is calculated as;

, , ,

,

2 p BBF i BBF BBF s

BBF s

BBF

f L LR

QF

(5.10)

The smoothing reactor ,o BBFL is selected 3-5 % of load impedance value and

calculated as;


Adnan TAN

122

2

,

,

1

3 5 %2

L L rms

o BBF

Load

VL

f S

(5.11)

Using these equations initial values of LLCL type broad band filter is

determined in order to compensate power quality problems of modeled CSI-IF. The

specifications and initial values of LLCL type broad-band filter designed for CSI-IF

is given in Table 5.3.

Table 5.3.Initial Values for Power Circuit Parameters of LLCL Type Broad-Band

Filter Model

LLCL Type Broad-Band Filter

Initial Parameters

BBFQ = 2MVAr , ,p BBFf =150 Hz,

,s BBFf =250 Hz, 1.5BBFQF

,i BBFL 120 mH

,o BBFL 12 mH

,BBF sL 64 mH

,BBF sC 2.05µF

,BBF sR

34 Ω

The power quality design constraints of LLCL type broad-band filter are

input current THD, input power factor and filter output voltage regulation (Havva et

al., 2005). The LLCL type broad-band filter designed for CSI-IF should provide

input current THD below 5%, supply 2MVAr reactive power and keep the filter

output regulation below 5%. With initial values of LLCL filter, these desired

specifications of LLCL type filter for CSI-IF are not maintained. In order to maintain

these specifications the optimum values of filter is determined by the help of

simulation program. The specifications and final values of LLCL type filter designed

for modeled CSI-IF is shown in Table 5.4.


Adnan TAN

123

Table 5.4. Final Values for Power Circuit Parameters of LLCL Type Broad-Band

Filter Model

LLCL Type Broad-Band Filter

Final Parameters

,i BBFL 42 mH

,o BBFL 12 mH

,BBF sL 36 mH

,BBF sC 3.7µF

,BBF sR

38 Ω

The single line diagram of LLCL filters with CSI-IF and equivalent circuit



sysL

CSI-IF

CSI-IF

,CSI IF harI

Filter SystemZ(a) (b)

,BBF sL

,BBF Y sC

,BBF sR

,i BBFL

,o BBFL sys

L

,BBF sL

,BBF Y sC

,BBF sR

,i BBFL

,o BBFL

Figure 5.26. Single Line Diagram and Equivalent Circuit of CSI-IF with LLCL Type

Broad Band Filter

(a) Single Line Diagram of CSI-IF with LLCL Type Broad Band Filter

(b) Equivalent Circuit of CSI-IF with LLCL Type Broad Band Filter at

harmonic frequencies

The impedance-frequency diagram of designed LLCL filter with power

system is shown in Figure 5.27. By investigating the impedance-frequency curve of

LLCL type filter, the filtering characteristic of LLCL type filter cannot be

determined because of the series inductors of filters.


Adnan TAN

124

Figure 5.27. Impedance-Frequency Curve of LLCL Type Broad-Band Filter

Designed for CSI-IF

In order to show the filtering characteristic of LLCL filter, the AC analysis

technique is applied to the circuit shown in Figure 5.26(b). By applying 1A reference

from current source which represents the CSI-IF to the system in Figure 5.26(b), how

much current flow to system and shunt branch of LLCL filter is determined. As

shown in Figure 5.28, above the 250 Hz, approximately only 40 percentage of

reference current flow to the power system and 60 percentage of current flow to the

shunt branch of LLCL filter.


Adnan TAN

125

Reference Current Current Drawn by

Shunt Brach of

BBF FilterCurrent Drawn by

Power System

Figure 5.28. Filtering Characteristic of LLCL Type Broad-Band Filter Designed for

CSI-IF

This effective filtering performance of designed LLCL type broad-band filter

is shown in the PSCAD/EMTDC simulation program. The LLCL type filter is placed

to the input of CSI-IF as shown in Figure 5.29.

sysL

sysL

sysL

,a sourceI

,b sourceI

,c sourceI

3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

,a CSI IFI

,b CSI IFI

,c CSI IFI

,a BBFI

,b BBFI

,c BBFI

CSI-IF

,BBF sR

,BBF sC

,BBF sL

,BBF sR

,BBF sC

,BBF sL

,BBF sC

,BBF sR

,BBF sL

,o BBFL

,i BBFL

,o BBFL

,i BBFL

,o BBFL

,i BBFL

,a sourceV

,b sourceV

,c sourceV

, a CSI IFV

, b CSI IFV

, c CSI IFV

Figure 5.29. Power Circuit Model of LLCL Type Broad-Band Filter


Adnan TAN

126

The waveforms of CSI-IF currents, source currents and shunt branch of

LLCL filter currents are shown in Figure 5.30, Figure 5.31and Figure 5.32,

respectively. It is shown that LLCL type filter has effective filtering performance.

During the first second of simulation which the current waveforms are taken, THD of

CSI-IF current is 14.35 % and source current THD decreases to 4.85 %. During the

overall simulation time which CSI-IF operates from 150 Hz to 250 Hz, THD of CSI-

IF current is varying from 15.7 % to 7 % and source current THD is varying from 5

% to 2.5 %.

Figure 5.30. CSI-IF Current Waveform in LLCL-Type BBF Filter Simulation

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Curr

ents

(kA

)



Adnan TAN

127

Figure 5.31. Source Current Waveform in LLCL-Type BBF Filter Simulation

Figure 5.32. Current Waveform of Shunt Branch of LLCL Type Filter in LLCL-Type

BBF Filter Simulation

The waveforms of CSI-IF voltage and source voltage are shown in Figure

5.33 and Figure 5.34. respectively. The main drawback of LLCL type filter is shown

in Figure 5.34. Because of the high values of series inductors of LLCL filter, the

current harmonics and interharmonics cause significant voltage drop on the inductors

and create distorted voltage waveforms. During the first second of simulation which

the voltage waveform of CSI-IF is taken, the THD of CSI-IF voltage is 9.1%. During

Source : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

Supply

Curr

ent

(kA

)


BBF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.200

-0.150

-0.100

-0.050

0.000

0.050

0.100

0.150

0.200

BB

F C

urr

ents

(kA

)

Ia_BBF Ib_BBF Ic_BBF


Adnan TAN

128

the overall simulation time which CSI-IF operates from 150 Hz to 250 Hz, THD of

CSI-IF voltage is varying from 6.5 % to 12 %.

Figure 5.33. Source Voltage Waveform in LLCL-Type BBF Filter Simulation

Figure 5.34. CSI-IF Voltage Waveform in LLCL-Type BBF Filter Simulation

The reactive power drawn from source and CSI-IF are shown in Figure 5.35.

It is shown that LLCL type filter provides irregular reactive power compensation. At

the beginning of simulation CSI-IF operates at low power and after 6th

second of the

simulation CSI-IF reaches the maximum power. The reactive power value of the

SYS : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-40

-30

-20

-10

0

10

20

30

40

Supply

Voltages

(kV

)

Va_source Vb_source Vc_source

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-40

-30

-20

-10

0

10

20

30

40

CS

I-IF

Voltages

(kV

)

Va_CSI-IF Vb_CSI-IF Vc_CSI-IF


Adnan TAN

129

LLCL filter is designed to rated power furnace and when the CSI-IF operates at the

rated power, LLCL filter supplies 2 MVAr reactive power as shown in Figure 5.35.

When the CSI-IF operates at low powers, the current magnitude of CSI-IF decreases

as expected. When the current of CSI-IF decreases, the voltage drop on ,i BBFL

decreases and the voltage at the connection point of shunt branch of filter increased.

Thus, increasing of voltage at the connection point of shunt branch of filter increases

the reactive power supplied by the capacitor so when the CSI-IF operates at lower

powers the reactive power supplied from filter is increased. By the help of designing

the voltage regulation of filter 4.96 %, the reactive power capacity of LLCL filter is

kept so as to avoid over compensation of CSI-IF as shown in Figure 3.35.

Figure 5.35. Reactive Powers Drawn From source and Drawn By CSI-IF in LLCL

Type Filter Simulation

The harmonic spectrums of CSI-IF currents and source currents are given in






spectrums of CSI-IF currents and source currents are investigated, it is shown that

SYS,CSI-IF : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

-1.00

-0.50

0.00

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

5.00

React

ive P

ow

ers

(M

VA

r)

Q_sys Q_CSI-IF


Adnan TAN

130

LLCL type filter eliminates approximately 60 % of harmonic and interharmonic

currents.

The harmonic spectrum of CSI-IF voltage harmonics is given in Figure 5.38

while CSI-IF is operating at 175 Hz. It is shown that the same harmonics and

interharmonics in the load current occur at the voltage harmonic spectrum.


Adnan TAN

131

Fig

ure

5.3

6.

Har

monic

Spec

trum

of

Low

er O

rder

Har

monic

s of

CS

I-IF

Curr

ent

and S

ourc

e C

urr

ent

in L

LC

L T

ype

Bro

ad-B

and

Pas

sive

Fil

ter

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

132

Fig

ure

5.3

7.

Har

monic

Spec

trum

of

Hig

her

Ord

er H

arm

onic

s of

CS

I-IF

Curr

ent

and S

ourc

e C

urr

ent

in L

LC

L T

ype

Bro

ad-B

and

Pas

sive

Fil

ter

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

133

Fig

ure

5.3

8. H

arm

onic

Spec

trum

of

CS

I-IF

Volt

age

LL

CL

Type

Bro

ad-B

and P

assi

ve

Fil

ter

Sim

ula

tion

(a)

Low

er O

rder

Har

mon

ics

wit

h L

arge

Sca

le

(b)

Hig

her

Ord

er H

arm

onic

s w

ith L

ow

er S

cale


Adnan TAN

134

5.2. Hybrid Active Power Filters

In order to form an effective compensation system and decrease loading of

APFs in high power applications, APF and passive filter combinations which are

named as HAPF are commonly used. The most common HAPF types which are

introduced in Section 3.3.2 are shunt APF - shunt passive filter combination, series

APF - shunt passive filter combination and APF series with shunt passive filter

combination which is also named as shunt hybrid active power filter (SHAPF).

The PQ problems of modeled CSI-IF is presented in Section 4 and the

inadequate PQ compensation performances of passive filters are proposed in section

5.1. To solve PQ problems and reactive power compensation of CSI-IF with only

pure active power filters is theoretically possible with increasing the number of APFs

or using multilevel topologies, but in practice because of the cost and design

complexities, such a system formed from pure APFs is not feasible. Because of

these, HAPFs has an important role in high power applications. In this section of

thesis, shunt APF - shunt passive filter type HAPF and SHAPF systems are designed

for power quality and reactive power compensation of the modeled CSI-IF and, their

performances and ratings are investigated.

5.2.1. Shunt Active Power Filter and Parallel Passive Filter Combination

5.2.1.1. Power Circuit Configuration of Proposed HAPF System

The proposed HAPF formed from shunt APF and shunt passive filter for

compensating the power quality problems of modeled CSI-IF is shown in Figure

5.39. The shunt APF is formed from two identical modules which are connected to

31.5 kV busbar via transformer. The proposed APF is divided into two module which

are formed from 2 level bridge VSI based APFs because of the large current ratings

of compensating harmonics and interharmonics of CSI-IF. In Section 4, the dominant

harmonics and interharmonics of CSI-IF are presented. The harmonics and

interharmonics content of CSI-IF are at higher values when the furnace operates at


Adnan TAN

135

lower frequencies. The maximum rms value of the harmonics and interharmonics of

CSI-IF formed at the same time are calculated 35A-rms at 31.5k V level. The shunt

APF must be connected to 31.5kV level via transformer because of the limits of

power semiconductor devices in VSI. In proposed design 31.5/1 kV Y-Y connected

transformer is used to connect APF to the 31.5kV busbar. APF must produce 1100A-

rms at 1kV level in order to compensate the harmonic and interharmonic currents of

CSI-IF. This is possible with very little input reactor value and extremely high DC

link voltage value. DC link voltage value of APF is limited with the ratings of power

semiconductors in the VSI. Instead of 2 level VSI, multilevel inverters can be

preferred but, multilevel inverters have disadvantages in control complexity and high

costs. Because of these reasons, shunt APF is divided into two identical modules.

CSI-IF3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

3MVA

Y/Y

31.5/1kV

PASSIVE

FILTER

Q=2MVAR

SHUNT APF SYSTEM

SHUNT APF

MODULESHUNT APF

MODULE

Figure 5.39. Proposed HAPF System for CSI-IF


Adnan TAN

136

The power circuit configuration of APF module is formed from interface

inductor, VSI and DC-Link Capacitor as shown in Figure 5.40. The interface

inductors establish a link between VSI and power system. Also the interface

inductors convert the VSI voltage to the current and allow APF to act as a current

source. The value of interface inductor is very crucial in the performance of the

APFs. If a small value of interface inductor is selected, then large switching ripples

are injected into the supply currents, and a large value of interface inductor does not

allow proper tracking of the compensating currents close to the desired values (Singh

et al., 1999). VSI which generates compensating currents with using capacitor

voltage is the main part of APF. Three phase 2 level bridge inverter is the most used

topology in APF applications. DC link voltage must be higher than the peak value of

the utility voltage, otherwise the generated compensation currents cannot be injected

to the power system (Uçak, 2010). DC link capacitors are used as energy storage

elements. APF eliminates harmonics and/or other power quality problems by

supplying energy to DC link capacitors and/or consuming energy from DC link

capacitors. DC link capacitor value of the APF is another important parameter. With

a small value of, large ripples in the steady state and wide fluctuations in the DC link

voltage under transient conditions are observed. A higher value of reduces ripples

and fluctuations in the dc-bus voltage, but increases the cost and size of the system

(Singh et al., 1999).

VSI

INTERFACEINDUCTORS

DC LINK

Figure 5.40. Power Circuit Configuration of APF Modules


Adnan TAN

137

In proposed HAPF system, the reactive power compensation of CSI-IF is

performed by 2 MVAr single tuned passive filters tuned at 135 Hz. The design and

performance of this filter are investigated in Section 5.1.1, it is shown that this 2

MVAr single tuned filter tuned at 135 Hz does not cause any resonance problem and

performs the reactive power requirements of CSI-IF successfully.

5.2.1.2. Control Method of APF Modules in Proposed HAPF System

In the proposed HAPF System, the controllers of each APF modules are

different but the same control method is used at both of the controllers of APF

modules. The block diagram of control method of APF modules is shown in Figure

5.41. It is shown that the control method of APF modules is formed from the

harmonic current extraction, DC link voltage controller and current controller. The

harmonic current extraction method uses the source currents which are the controlled

variables of this control method so the proposed controller method has a feedback

control configuration and APFK is the feedback gain of the controller.

Harmonic

Extraction

DC Link

Controller

Current

Controller

VSI

&

Interface

Reactor

+hI

X

APFK

-

,cap APFV

sourceV

,capref APFI

,ref APFI

+

-

,error APFI

+

-Gate

Signals APFI

sourceI

CSI IFI

sourceI

Figure 5.41. Control Method of Shunt APF Modules in Proposed HAPF System

(1) Harmonic Extraction Method:The block diagram of harmonic current

extraction method used in control method of APF modules is shown in Figure

5.43. In this thesis study, in order to calculate the reference harmonic currents,

an adaptive filtering method named enhanced phase locked loop (EPLL) method

is used. The reason of preferring EPLL is that it has simple structure than most


Adnan TAN

138

preferred time based and frequency based methods and it has fast and accurate

harmonic extraction capability with changing load conditions.

,a sourceI EPLL

,a hI

,b sourceI EPLL

,b hI

,c sourceI EPLL

,c hI

Figure 5.42. Harmonic Extraction Method in Shunt APF Modules

EPLL is proposed by Karimi-Ghartemani and Irvani (2001). The EPLL is

formed from three main parts as conventional phase lock loop (PLL) as shown in

Figure 5.43. These are phase detector (PD), low pass filter (LPF) and voltage

controlled oscillator (VCO). The structure and functions of LPF and VCO in EPLL is

same as LPF and VCO of PLL. The innovation is performed in PD of EPLL. The

new PD adds new features like amplitude estimation, in phase output signal and

more robust and stable loop than conventional PLL. Moreover EPLL has higher

convergence time than conventional PLL. These new features are performed by

changing the structure of PD. The amplitude estimation feature is achieved by adding

a peak detector mechanism into PD. There is used very basic peak detector

mechanism but this structure has many successful effects on the stability, robustness

and convergence time of loop. The speed, accuracy and robustness of EPLL is

proved by Karimi-Ghartemani et al. (2002; 2005).

The EPLL receives the input signal iI and provides an on-line estimate of the

following signals (Karimi-Ghartemani et al., 2004):


Adnan TAN

139

The synchronized fundamental component, ( )fI t

The amplitude, ( )fA t , of ( )fI t .

The phase angle, ( )f t , of ( )fI t .

Time-derivatives of the amplitude, phase and frequency.

The error signal, ( ) ( ) ( )i fe t I t I t , is the total distortion signal of the

input.

x + +

PHASE DETECTOR

PD

LOW-PASS FILTER

LPF

VOLTAGE-CONTROLLED

OSCILLATOR

VCO

+x

-

x

iI

AK

PK

iK

s1

s1

fA

s1

sin

sin

2

fI

o

f

e

Figure 5.43. Structure of EPLL

The gains and time constants of integrals can affect the lock time of loop,

amplitude estimation time and phase accuracy of input signal. Increasing the value of

Ka decreases the estimation time of amplitude and loop lock time but if Ka is

increased so much, it will start to increase the oscillations in the amplitude signal and

disturb the shape of fundamental signal. Decreasing Ka and Kp yields an estimation

of the peak which is insensitive/robust to the undesirable variations and noise in the

input signal (Karimi-Ghartemani et al., 2005).

When the distorted current or voltage signal is applied to EPLL, the

harmonics and interharmonics of distorted signal can be obtained from the ( )e t signal

of EPLL. The performance of EPLL is shown in PSCAD/EMTDC program. The


Adnan TAN

140

current signal of CSI-IF is applied to the EPLL, EPLL extracts the fundamental

components and harmonics and interharmonics of CSI-IF current successfully as

shown in Figure 5.44.

Figure 5.44. Distorted Input signal, Extracted Fundamnetal Signal and Extracted

Harmonics of EPLL

EPLL : Graphs

sec 1.0000 1.0050 1.0100 1.0150 1.0200 1.0250 1.0300 1.0350 1.0400 1.0450 1.0500 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Curr

ent

(kA

)

Ia_CSI-IF

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

Fun.C

om

p.o

fCS

I-IF

Cur.

(kA

)

Ia_CSI-IF_fun

-0.100

-0.080

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

0.080

0.100

Harm

.ofC

SI-

IFC

ur.

(kA

)

Ia_CSI-IF_har


Adnan TAN

141

(2) DC Link Voltage Controller:The active filter does not require any external dc

power supply, because it can build up and regulate the dc voltage across the

capacitor by itself (Tangtheerajaroonwong et al., 2007). The DC link control of

Shunt APF is achieved by control of active power. In ideal conditions APF does

not consume any active power but due to switching and conduction losses of

switching devices, and resistance of inverter output filter, APF must consume

active power to control the DC link voltage. The block diagram of DC link

controller is shown in Figure 5.45. DC link voltage control is achieved by using

PI controller. In order to keep DC link voltage at a constant level, APF must

draw active power by drawnig current in phase with line voltage. To draw a

current in the same phase with system voltage, phase information of system

voltage must be known. This can be achieved by using EPLL. When the phase

voltage of system is applied to EPLL, EPLL gives the phase information of

system voltage. With using phase of system voltage, DC link control reference

current signal is created by multiplying the PI controller output and sinewave

created by phase information of system voltage as shown in Figure 5.45.


Adnan TAN

142

,capset APFV

+

-

PI

CONTROL

,cap APFV

,cappi APFV

,a sourceV EPLL

_ ,a f sourceV

+

-

+

+

sin

sin

sin

X

X

X

_ ,a capref APFI

_ ,b capref APFI

_ ,c capref APFI

2 3

2 3

,cappi APFV

,cappi APFV

,cappi APFV

Figure 5.45. DC Link Voltage Control of Shunt APF Modules

(3) Current Controller:In the current controller of APF modules, PWM switching

method is used as shown in Figure 5.46. The switching pulses are generated by

comparing the difference of reference current and APF current with triangular

wave. PWM switching method obtains a constant switching frequency to the

power electronics devices in VSI. This is important because in high voltage

applications, power electronics devices have limited switching frequency

capabilities because of the turn-on and turn-off times and switching losses.

Because of these, the switching frequency of power electronics devices in APF

modules is selected as 3000 Hz.


Adnan TAN

143

+

-Triangular

Wave

_ ,a ref APFI

+

-

,a APFI

+

-Triangular

Wave

_ ,b ref APFI

+

-

,b APFI

+

-Triangular

Wave

_ ,c ref APFI

+

-

,c APFI

G

A

T

E

S

I

G

N

A

L

S

Figure 5.46. Current Control of Shunt APF Modules

5.2.1.3. Simulation Results of Proposed HAPF System

The performance of the proposed HAPF system in compensating the PQ

problems of CSI-IF is investigated in PSCAD/EMTDC simulation program. In this

section the simulation results of system is presented. The proposed HAPF system is

connected to the input of CSI-IF as shown in Figure 5.47. The power circuit

parameters of proposed HAPF used in simulation program are given in Table 5.5.


Adnan TAN

144

sysL

sysL

sysL

,a sourceI

,b sourceI

,c sourceI

,a sourceV

,b sourceV

,c sourceV

3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

,a CSI IFI

,b CSI IFI

,c CSI IFI

,a STFI

,b STFI

,c STFI

CSI-IF

STFR

STFC

STFL

STFR

STFC

STFL

STFC

STFR

STFL

,a APFsI

,b APFsI

,c APFsI

1APFI 2APF

I

, a CSI IFV

, b CSI IFV

, c CSI IFV

, 1cap APFV, 2cap APFV-+ -+

APFL APF

L

,dc APFC

,dc APFC

3MVA

Y/Y

31.5/1kV

Figure 5.47. Power Circuit Model of Proposed HAPF System


Adnan TAN

145

Table 5.5. Power Circuit Parameters of HAPF System

The Proposed HAPF System

Coupling Transformer

3MVA

31.5/1kV Y/Y

Vsc%=%2

Shunt APF Module APFL = 100µH

,dc APFC = 25mF

Single Tuned

Passive Filter

STFQ = 2MVAr

STFf =135 Hz

STFQF =50

STFL =251mH

,STFC =1.845µF

STFR =4.26Ω

The waveforms of CSI-IF currents, current drawn from the connection point

of CSI-IF and single tuned filter, source currents are shown in Figure 5.48, 5.49 and

5.50 respectively. It is shown that proposed HAPF system cannot filter harmonics

and interharmonic currents of CSI-IF completely but it has sufficient filtering

performance on keeping the harmonic currents under the limits. During the first

second of the simulation which the current waveforms are taken, THD of CSI-IF

current is 14.35 % and source current THD decreases to 4.1 %. During the overall

simulation time which CSI-IF operates from 150 Hz to 250 Hz, THD of CSI-IF

current is varying from 15.7 % to 7 % and source current THD is varying from 4.5 %

to 2 %.


Adnan TAN

146

Figure 5.48. CSI-IF Current Waveform in Proposed HAPF Simulation

Figure 5.49. CSI-IF and STF Current Waveform in Proposed HAPF Simulation

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Curr

ents

(kA

)


CSI-IF+STF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

+S

TF

Curr

ents

(kA

)

Ia_CSI-IF+STF Ib_CSI-IF+STF Ic_CSI-IF+STF


Adnan TAN

147

Figure 5.50. Source Current Waveform in Proposed HAPF Simulation

In Figure 5.51 and Figure 5.52, compensating currents of passive filters and

APFs are shown. APFs in proposed HAPF system generates harmonic and

interharmonic currents whereas the passive filter provides 2MVAr reactive power

requirement of CSI-IF in order to compensate the PQ problems of CSI-IF.

Figure 5.51. STF Currents in Proposed HAPF Simulation

SYS : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

Sourc

e C

urr

ents

(kA

)


STF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.080

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

0.080

ST

F C

urr

ents

(kA

)

Ia_STF Ib_STF Ic_STF


Adnan TAN

148

Figure 5.52. High Voltage Side APFs Currents in Proposed HAPF Simulation







spectrums of CSI-IF currents and source currents are investigated, it is shown that

proposed HAPF system eliminates approximately 60 % of harmonic and

interharmonic currents.

APFs : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

AP

FsC

ur.

A (

kA)

Ia_APFs

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

AP

FsC

ur.

B (

kA)

Ib_APFs

-0.060

-0.040

-0.020

0.000

0.020

0.040

0.060

AP

FsC

ur.

C (

kA)

Ic_APFs


Adnan TAN

149

Fig

ure

5.5

3.

Har

monic

S

pec

trum

of

Low

er

Ord

er

Har

monic

s of

CS

I-IF

C

urr

ent

and

Sourc

e C

urr

ent

in

Pro

pose

d

HA

PF

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

150

Fig

ure

5.5

4. H

arm

on

ic

Spec

tru

m

of

Hig

her

O

rder

H

arm

on

ics

of

CS

I-IF

C

urr

ent

and

Sourc

e C

urr

ent

in

Pro

pose

d

HA

PF

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

151

In Figure 5.55, the voltage waveforms measured from the input of CSI-IF is

given. It is shown that the distortion caused by the low order harmonics and

interharmonics of CSI-IF are not seen at the voltage waveform. However, switching

noise of APFs in the proposed system causes high frequency distortion in the voltage

waveform because of the weak power system.

Figure 5.55. CSI-IF Voltage Waveform in Proposed HAPF Simulation

The reactive powers drawn from source, CSI-IF and single tuned filters of

proposed HAPF are shown in Figure 5.56. It is shown that 2MVAr reactive power is

supplied by single tuned filters of proposed HAPF system and reactive power drawn

from the source is decreased. By the help of the proposed system, reactive power

compensation requirements of CSI-IF is provided and reactive power demand of

system is kept under the limits.

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-30

-20

-10

0

10

20

30

CS

I-IF

Voltages

(kV

)



Adnan TAN

152

Figure 5.56. Reactive Powers Drawn From source, Drawn By CSI-IF and Drawn By

Single Tuned Filter in HAPF Simulation

In Figure 5.57 and Figure 5.58 the injected currents of each APF modules in

the HAPF system are presented. It is seen that the identical APF modules are

generating the same injected currents in same amplitudes. The peak points of injected

currents of each APF module reach to 800A.

SYS,STF,CSI-IF : Graphs

sec 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 ...

...

...

-3.00 -2.50

-2.00 -1.50 -1.00 -0.50

0.00 0.50 1.00 1.50

2.00 2.50 3.00 3.50

4.00 4.50 5.00

React

ive P

ow

ers

(M

VA

r)

Q_source Q_STF Q_CSI-IF


Adnan TAN

153

Figure 5.57. Injected Current Waveform of APF1 in Proposed HAPF Simulation

APF1 : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.80

-0.40

0.00

0.40

0.80

AP

F1C

urA

(kA

)

Ia_APF1

-0.80

-0.40

0.00

0.40

0.80

AP

F1C

urB

(kA

)

Ib_APF1

-0.80

-0.40

0.00

0.40

0.80

AP

F1C

urC

(kA

)

Ic_APF1


Adnan TAN

154

Figure 5.58. Injected Current Waveform of APF2 in Proposed HAPF Simulation

In Figure 5.59 and Figure 5.60, the DC link voltage trends of each APF

module are presented. It is shown that the DC link voltage controllers of APF

modules operate successfully and keep the DC link voltage at constant 2000V level.

Also the ripples of DC link voltages are in sufficient levels. Besides the switching

ripples in the DC link voltage there is a non-regular ripple is seen on the DC link

voltage. This non-regular ripple is caused by the time varying harmonics and

interharmonics of CSI-IF.

APF2 : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.80

-0.40

0.00

0.40

0.80

AP

F2C

urA

(kA

)

Ia_APF2

-0.80

-0.40

0.00

0.40

0.80

AP

F2C

urB

(kA

)

Ib_APF2

-0.80

-0.40

0.00

0.40

0.80

AP

F2C

urC

(kA

)

Ic_APF2


Adnan TAN

155

Figure 5.59. DC Link Voltage Waveform of APF1 in Proposed HAPF Simulation

Figure 5.60. DC Link Voltage Waveform of APF2 in Proposed HAPF Simulation

The reference currents which are calculated in the controllers and injected

currents of each APF module in HAPF system are shown in Figure 5.61 and Figure

5.62. It is shown that in both of APF modules, the injected currents cannot catch the

reference currents so the reference currents cannot be completely generated by APF

modules of HAPF. At this point, the DC link voltage of APF modules must be

increased in order to generate calculated reference currents completely. However, it

APF1 : Graphs

sec 1.000 1.020 1.040 1.060 1.080 1.100 1.120 1.140 1.160 1.180 1.200 ...

...

...

1.900

1.920

1.940

1.960

1.980

2.000

2.020

2.040

2.060

2.080

2.100

AP

F1 D

C-L

INK

Voltage (

kV)

Vcap_APF1

APF2 : Graphs

sec 1.000 1.020 1.040 1.060 1.080 1.100 1.120 1.140 1.160 1.180 1.200 ...

...

...

1.900

1.920

1.940

1.960

1.980

2.000

2.020

2.040

2.060

2.080

2.100

AP

F2 D

C-L

INK

Voltage (

kV)

Vcap_APF2


Adnan TAN

156

is not feasible to realize such a 3 phase 2 level VSI with DC link voltage over 2000V

in practical applications, because of the limited ratings of power electronics devices.


Proposed HAPF Simulation


Proposed HAPF Simulation

According to these simulation results, each VSI power rating in APFs of

HAPF system is calculated as ;

APF1_Control,APF1 : Graphs

sec 1.0000 1.0050 1.0100 1.0150 1.0200 1.0250 1.0300 1.0350 1.0400 ...

...

...

-1.50

-1.00

-0.50

0.00

0.50

1.00

1.50

AP

F1 R

ef&

Inje

cted C

urr

ents

(kA

)

Ia_ref_APF1 Ia_APF1

APF2_Control,APF2 : Graphs

sec 1.0000 1.0050 1.0100 1.0150 1.0200 1.0250 1.0300 1.0350 1.0400 ...

...

...

-1.60

-1.20

-0.80

-0.40

0.00

0.40

0.80

1.20

1.60

AP

F1 R

ef&

Inje

cted C

urr

ents

(kA

)

Ia_ref_APF2 Ia_APF2


Adnan TAN

157

, ,3

2 2

cap APF APF peak

HAPF VSI

V IS (5.12)

From Eq. 5.12, the each converter power rating is find as,

2000 8003 1385

2 2HAPF VSI

V AS kVA (5.13)

And total power ratings of converters of APFs in proposed HAPF is approximately

equal to 2 1385 2770kVA kVA .

5.2.2. Active Power Filter Series with Passive Filter Combination: SHAPF

5.2.2.1. Power Circuit Configuration of Proposed SHAPF System

The proposed SHAPF for PQ problems of CSI-IF is formed from two parallel

connected SHAPF modules which are connected to 31.5kV busbar via 31.5/1 kV Y-

Y connected transformer as shown in Figure 5.62. The proposed SHAPF is divided

into two identical module because the passive filter ratings cause high APF currents

in order to obtain the reactive power requirements of CSI-IF. From the simulation

results of CSI-IF in Section 4, it is determined that 2 MVAr compensation system is

necessary for keeping the reactive power demand of system under the limits. This

causes approximately 1150 A-rms to flow into the APF converter. Also as introduced

in the Section 5.2.1.1, in order to compensate the harmonic and interharmonic

current of CSI-IF additionally 1100 A-rms must be drawn by the SHAPF. In total,

approximately 1600 A-rms must be drawn by SHAPF and also by APF converter. In

SHAPF topology this high value of harmonic and interharmonic currents do not

cause extremely high DC link voltage in converter of SHAPF as in pure shunt APF,

because due to the presence of the LC filter, the fundamental voltage is decoupled

from the power system, thus reducing the voltage rating of the active filter (Inzunza

et al.,2005). If a single branch SHAPF is designed for PQ problems of CSI-IF,


Adnan TAN

158

1600A-rms current must be drawn by SHAPF and APF converter. This problem may

be solved by using high power rating power electronic devices but using of high

power devices in high switching frequencies cause serious switching losses. Because

of these problems, the proposed SHAPF is designed with two SHAPF by placing the

passive filters to the low voltage side of the transformers.

CSI-IF3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

3MVA

Y/Y

31.5/1kV

SHAPF SYSTEM

SHAPF

MODULE

SHAPF

MODULE

Figure 5.63. Proposed SHAPF System for CSI-IF

The compensation characteristic of SHAPFs is different from pure shunt

APFs. In pure shunt APF, APF acts as current source and filtering the harmonic

currents of nonlinear load by injecting the reverse of the harmonic currents to the

power system. In SHAPF, the passive filter suppresses harmonic currents produced

by the load, whereas the active filter improves the filtering characteristics of the

passive filter by acting as a harmonic isolator between the source and the load (Fujita


Adnan TAN

159

et al., 1991). This compensation characteristic of SHAPF is explained with single

branch SHAPF. The single line equivalent circuits of SHAPF compensation system

is shown in Figure 5.64.

(c)

sysZ

+-

PFZ

sourceI

,load hIPFI

sysZ

+-

PFZ

,source hI

,load hI,PF hI

,SHAPF SHAPF source hV K I

sysZ

PFZ

,source hI

,load hI,PF hI

SHAPFK

(a) (b)

Figure 5.64. Single Line Equivalent Circuit of SHAPF

In Figure 5.64(a), the 50 Hz equivalent circuit of system is represented.

Assuming that the active power filter is an ideal controllable voltage source APFV ,

and that the load is a current source loadI . sV is the source voltage , sysZ is the source

impedance, PFZ is the impedance of single tuned filter, sourceI is the source current

and PFI represents the passive filter current. In Figure 5.64(b) the harmonic

equivalent of circuit in Figure 5.64(a) is given. In harmonic frequencies, sV is equal

to zero, and represented as short circuited. The controllable voltage source APFV

depends on the source harmonic currents as,

,APF SHAPF source hV K I (5.14)


Adnan TAN

160

where SHAPFK is the feedback gain of SHAPF controller. If the circuit in Figure

5.64(b) is analyzed by applying Kirchhoff’s laws with Superposition Theorem, shI is

found as;

,PFh

source h

PFh sysh SHAPF

ZI

Z Z K

(5.15)

Eq. 5.15 shows that APF acts as a resistor in circuit shown in Figure 5.64(c)

to damp parallel resonance between sysZ and PFZ , and increase the filtering

performance of passive filter (Fujita et al., 1991; Srianthumrong et al., 2003). From

the Eq. 5.15, it is shown that theoretically the harmonic content of the source current

goes to zero when SHAPFK approaches to infinity. However due to the stability

problems in the control loop, the gain SHAPFK should be limited to certain values

(Uçak, 2010).

The power circuit configuration of one SHAPF module of proposed system is

formed form single tuned filter, VSI and DC-Link capacitor shown in Figure 5.65.

VSI is the main part of SHAPF and generates voltage reference signal in order to

maintain the harmonic isolation between passive filter and system. Three phase 2

level bridge inverter topology can be easily used through the low DC link voltage of

SHAPF. As in the pure shunt APF, DC link capacitors are used as energy storage

elements. The DC link capacitor value has enough value to keep the DC link voltage

ripple in acceptable limits.


Adnan TAN

161

VSI

PASSIVEFILTERS

DC LINK

Figure 5.65. Power Circuit Configuration of SHAPF Modules

In proposed SHAPF system, the design of single tuned filter is realized

according to reactive power demand and the harmonic spectrum of CSI-IF currents.

The power ratings of single tuned filters in each SHAPF module are determined as

1MVAr in order to meet the reactive power demand of CSI-IF. The tuned frequency

of passive filter in SHAPF is generally selected the most dominant harmonic

frequency of load current (Cheng, 1998). The dominant harmonics of CSI-IF are

changed with operating frequency of furnace and they are seen between 250Hz and

650Hz as introduced in Section 4. In order to maintain equal filtering characteristic

between 250 Hz and 650 Hz, tuning frequency of passive filters is selected at 300Hz.

The filter values are determined by using the design equations of single tuned filter

in Section 5.1.1. In the design of single tuned filter of proposed SHAPF, the

transformer leakage reactance must be taken into account when inductor value of

passive filter is calculated. The filtering characteristics of only one module and both

of two modules of proposed SHAPF system at different SHAPFK values are shown in

Figure 5.66 and Figure 5.67 respectively. It is shown that proposed SHAPF system

obtain effective filtering in wide harmonic spectrum range.


Adnan TAN

162

KSHAPF=1

KSHAPF=1.5

KSHAPF=2

Figure 5.66. Filtering Characteristics of a Single SHAPF Module with Different

SHAPFK Values

KSHAPF=1

KSHAPF=1.5

KSHAPF=1

Figure 5.67. Filtering Characteristics of Proposed SHAPF System with Different

SHAPFK Values

5.2.2.2. Control Method of SHAPF Modules in Proposed SHAPF System

In the proposed SHAPF system, identical SHAPF modules have separate

controllers but the controllers of each SHAPF module use same control methods. The

general block diagram of control method of SHAPF modules is shown in Figure

5.68. It is shown that the control method of SHAPF modules is formed from the


Adnan TAN

163

harmonic current extraction, DC link voltage controller and voltage controller. The

harmonic current extraction method uses the source currents which are the controlled

variables of this control method so the proposed controller method has a feedback

control configuration and SHAPFK which is introduced in previous section is the

feedback gain of the controller.

Harmonic

Extraction

DC Link

Controller

Voltage

ControllerVSI

+hI

X

SHAPFK

-

,cap SHAPFV

,a SHAPFI

,capref SHAPFV

,ref SHAPFV

+

-Gate

Signals SHAPFI

sourceI

CSI IFI

sourceI

hV

Passive

Filter

+

-SHAPF

V

sourceV

Figure 5.68. Control Method of Shunt APF Modules in Proposed SHAPF System

In the control method of SHAPF modules, EPLL method is used as harmonic

extraction method as used in the controllers of APF modules in HAPF system.

Because the structure and operating principle of EPLL method are presented in

previous section, it is not presented again in this section.

The block diagram of DC link voltage controller of SHAPF modules is shown

in Figure 5.69. It is shown that DC link voltage control is obtained by PI controller.

As the pure shunt APF, SHAPF can build up and regulate the dc capacitor voltage

without any external power supply. The pure shunt APF obtains the DC link voltage

control by generating current in phase with the system voltage. However, SHAPF

controls the DC link voltage by generating voltage in phase with leading current flow

from passive filter. If SHAPF outputs a fundamental voltage that is in phase with the

fundamental leading current of the passive filter, the active power formed by the

leading current and the fundamental voltage is supplied to the dc capacitor (Fujita et

al., 1991). To generate a current in the same phase with fundamental component of

SHAPF current, fundamental component of SHAPF current must be known. In

controller of proposed SHAPF, this can be achieved by using EPLL as shown in

Figure 5.69. When SHAPF current is applied to EPLL, EPLL gives the phase


Adnan TAN

164

information of fundamental component of SHAPF current. Using phase of

fundamental component of SHAPF current, DC link control reference voltage signal

is created by multiplying the PI controller output and sinewave created by phase

information of fundamental component of SHAPF current as shown in Figure 5.69.

,capset SHAPFV

+

-

PI

CONTROL

,cap SHAPFV

,cappi SHAPFV

,a SHAPFI EPLL

_ ,a f SHAPFI

+

-

+

+

sin

sin

sin

X

X

X

_ ,a capref SHAPFV

_ ,b capref SHAPFV

_ ,c capref SHAPFV

2 3

2 3

,cappi SHAPFV

,cappi SHAPFV

,cappi SHAPFV

Figure 5.69. DC Link Voltage Control of SHAPF Modules

The APFs in SHAPF modules acts as voltage source. Therefore, voltage

references are created in the control method of SHAPF and these voltage references

are generated using voltage controller by APF in SHAPF module. The voltage

control of APF is achieved by PWM method as shown in Figure 5.70. The switching

pulses are generated by comparing the generating reference voltage signal with

triangular wave. In the proposed SHAPF the switching frequencies of power

electronic devices is selected 3000 Hz because the switching losses of power

electronic devices extremely increase with switching frequency of power electronic

devices.


Adnan TAN

165

+

-Triangular

Wave

_ ,a ref SHAPFV

+

-Triangular

Wave

_ ,b ref SHAPFV

+

-Triangular

Wave

_ ,c ref SHAPFV

G

A

T

E

S

I

G

N

A

L

S

Figure 5.70. Voltage Control of SHAPF Modules

5.2.2.3. Simulation Results of Proposed SHAPF System

The performance of the proposed SHAPF system in compensating the PQ

problems of CSI-IF is investigated in PSCAD/EMTDC simulation program. In this

section the simulation results of system is presented. The proposed SHAPF system is

connected to the input of CSI-IF as shown in Figure 5.71. The power circuit

parameters of proposed SHAPF used in simulation program is given in Table 5.6.


Adnan TAN

166

sysL

sysL

sysL

,a sourceI

,b sourceI

,c sourceI

,a sourceV

,b sourceV

,c sourceV

3 PHASE

POWER SUPPLY

31.5kV

Ssc = 335MVA

,a CSI IFI

,b CSI IFI

,c CSI IFI

CSI-IF

SHAPFsI

1SHAPFI

2SHAPFI

, a CSI IFV

, b CSI IFV

, c CSI IFV

, 1cap SHAPFV, 2cap SHAPFV-+ -+

SHAPFL

,dc SHAPFC

,dc SHAPFC

3MVA

Y/Y

31.5/1kV

SHAPFC

SHAPFL

SHAPFC

Figure 5.71. Power Circuit Model of Proposed SHAPF System


Adnan TAN

167

Table 5.6. Power Circuit Parameters of SHAPF System

The Proposed SHAPF System

Coupling Transformer

4MVA

31.5/1kV Y/Y

Vsc% = 3%

SHAPF Module

PFQ = 2MVAr

PFf =300 Hz

PFQF =20

SHAPFL =66µH

SHAPFC =3.1mF

,dc SHAPFC = 90mF

The waveforms of CSI-IF currents, source currents and the proposed SHAPF

currents in 31.5kV level are shown in Figure 5.72, Figure 5.73 and Figure 5.74

respectively. It is shown that proposed SHAPF system compensate the harmonic and

interharmonic currents of CSI-IF effectively and show a better performance than the

proposed HAPF system. During the first second of the simulation which the current

waveforms are taken, THD of CSI-IF current is 14.35% and source current THD

decreases to 2.9%. During the overall simulation time which CSI-IF operates from

150 Hz to 250 Hz, THD of CSI-IF current is varying from 15.7% to 7 % and source

current THD is varying from 3.2 % to 1.8 %. When the proposed SHAPF current

waveform is investigated, it is shown that the proposed SHAPF draws fundamental

leading current with injected harmonic and interharmonic currents.


Adnan TAN

168

Figure 5.72. CSI-IF Current Waveform in Proposed SHAPF Simulation

Figure 5.73. Source Current Waveform in Proposed SHAPF Simulation

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

CS

I-IF

Curr

ents

(kA

)


SYS : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.40

-0.30

-0.20

-0.10

0.00

0.10

0.20

0.30

0.40

Sourc

e C

urr

ents

(kA

)



Adnan TAN

169

Figure 5.74. High Voltage Side APFs Currents in Proposed SHAPF Simulation






harmonics and interharmonics are presented with lower scale. In this simulation,

SHAPFK is selected as 1.5 in the control methods of each SHAPF modules. When the

harmonic spectrums of CSI-IF currents and source currents are investigated, it is

shown that proposed SHAPF system shows an approximate filtering characteristic as

in the presented filtering characteristic curve in Section 5.2.2.1. SHAPF cannot show

the same characteristic in the filtering characteristic curve in Section 5.2.2.1 because

this filtering characteristic curve is for SHAPF which has an ideal voltage source

instead of VSI. As shown in the harmonic spectrums, the proposed SHAPF system

eliminates almost all harmonic currents at 300 Hz which is the tuned frequency of

APFs : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-0.080

-0.040

0.000

0.040

0.080

AP

FsC

ur.A

(kA

)

Ia_APFs

-0.080

-0.040

0.000

0.040

0.080

AP

FsC

ur.B

(kA

)

Ib_APFs

-0.080

-0.040

0.000

0.040

0.080

AP

FsC

ur.C

(kA

)

Ic_APFs


Adnan TAN

170

series connected passive filter, 80 % of harmonic currents at 400 Hz, 70% of

harmonic currents at 550 Hz, 60% of harmonic currents at 650 Hz and 750 Hz and

below than 50 % of harmonic current at above 1000 Hz.


Adnan TAN

171

Fig

ure

5.7

5.

Har

monic

S

pec

trum

of

Low

er O

rder

H

arm

onic

s of

CS

I-IF

C

urr

ent

and S

ourc

e C

urr

ent

in P

ropose

d S

HA

PF

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

172

Fig

ure

5.7

6.

Har

monic

S

pec

trum

of

Low

er O

rder

H

arm

onic

s of

CS

I-IF

C

urr

ent

and S

ourc

e C

urr

ent

in P

ropose

d S

HA

PF

Sim

ula

tion

(a)

Har

monic

Spec

trum

of

CS

I-IF

Curr

ent

(b)

Har

monic

Spec

trum

of

Sourc

e C

urr

ent


Adnan TAN

173

In Figure 5.77, the voltage waveforms measured from the input of CSI-IF is

given. It is shown that high frequency harmonics are shown in the voltage signal.

This high frequency distortion is the effect of the switching noise of SHAPF in weak

power system. When with compared to HAPF simulation results with SHAPF

results, the switching noise distortion in the SHAPF voltage is lower than HAPF

voltage because the series connected passive filters in SHAPF modules suppress the

switching ripples produced by VSIs.

Figure 5.77. CSI-IF Voltage Waveform in Proposed SHAPF Simulation

The reactive powers drawn from source, CSI-IF, each module of proposed

SHAPF system and complete of the proposed SHAPF system are shown in Figure

5.78. It is shown that 1MVAr reactive power is supplied by series connected single

tuned filters of each module of SHAPF and totally 2 MVAr reactive power is

supplied from complete of proposed SHAPF system. By the help of the proposed

SHAPF system, the reactive power compensation requirement of CSI-IF is provided

and the reactive power demand of CSI-IF can be kept under the limits.

CSI-IF : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-30.0

-25.0

-20.0

-15.0

-10.0

-5.0

0.0

5.0

10.0

15.0

20.0

25.0

30.0

CS

I-IF

Voltages

(kV

)



Adnan TAN

174

Figure 5.78. Reactive Powers Drawn From Source, Drawn By CSI-IF and Drawn By

SHAPF Modules in SHAPF Simulation

In Figure 5.79 and Figure 5.80 the injected currents in 1000V level of each

SHAPF modules in the proposed SHAPF system are presented. It is seen that the

identical SHAPF modules draw the same fundamental leading currents with same

injected harmonic and interharmonic currents. The peak points of injected currents of

each SHAPF modules reach to 1600A.

SYS,CSI-IF,APFs,APF1,APF2 : Graphs

0.00 1.00 2.00 3.00 4.00 5.00 6.00 7.00 8.00 9.00 10.00 11.00 12.00 ...

...

...

-3.00 -2.50

-2.00 -1.50 -1.00 -0.50

0.00 0.50 1.00 1.50

2.00 2.50 3.00 3.50

4.00 4.50 5.00

y (M

VA

r)

Q_source Q_CSI-IF Qapfs Qapf1 Qapf2


Adnan TAN

175



APF1 : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-1.60

1.60

AP

F1C

urA

(kA

)Ia_APF1

-1.60

1.60

AP

F1C

urB

(kA

)

Ib_APF1

-1.60

1.60

AP

F2C

urC

(kA

)

Ic_APF1

APF2 : Graphs

sec 1.0000 1.0100 1.0200 1.0300 1.0400 1.0500 1.0600 1.0700 1.0800 1.0900 1.1000 ...

...

...

-1.60

1.60

AP

F2C

urA

(kA

)

Ia_APF2

-1.60

1.60

AP

F2C

urB

(kA

)

Ib_APF2

-1.60

1.60

AP

F2C

urC

(kA

)

Ic_APF2


Adnan TAN

176

In Figure 5.81 and Figure 5.82, the DC link voltage trends of each SHAPF

module are presented. It is shown that the DC link voltage controllers of APF

modules operate successfully and keep the DC link voltage at constant 600V level.

Also the ripples of DC link voltages are in the sufficient levels. Besides the switching

ripples in the DC link voltage there is a non-regular ripple is seen on the DC link

voltage. This non-regular ripple is caused by the time varying harmonics and

interharmonics of CSI-IF.


Simulation


Simulation

APF1 : Graphs

sec 1.000 1.020 1.040 1.060 1.080 1.100 1.120 1.140 1.160 1.180 1.200 ...

...

...

0.5900

0.5920

0.5940

0.5960

0.5980

0.6000

0.6020

0.6040

0.6060

0.6080

0.6100

AP

F1 D

C-L

INK

Voltage (

kA)

Vcap_APF1

APF1 : Graphs

sec 1.000 1.020 1.040 1.060 1.080 1.100 1.120 1.140 1.160 1.180 1.200 ...

...

...

0.5900

0.5920

0.5940

0.5960

0.5980

0.6000

0.6020

0.6040

0.6060

0.6080

0.6100

AP

F1 D

C-L

INK

Voltage (

kA)

Vcap_APF1


Adnan TAN

177

In Figure 5.83 and Figure 5.84, the reference voltages which are calculated in

the controllers and the triangular signals which are generated in the voltage control

blocks are shown for each SHAPF module. It is shown that the generated voltage

reference signals are modulated uniformly and gate signal of VSIs of SHAPF

generated correctly.


Controller in Proposed SHAPF Simulation


Controller in Proposed SHAPF Simulation

APF1_Control : Graphs

sec 1.0000 1.0010 1.0020 1.0030 1.0040 1.0050 1.0060 1.0070 1.0080 1.0090 1.0100 ...

...

...

-0.030

-0.020

-0.010

0.000

0.010

0.020

0.030

AP

F1 R

efV

oltage (

kV)

Va_ref_APF1 PWM_tri_APF1

APF2_Control : Graphs

sec 1.0000 1.0010 1.0020 1.0030 1.0040 1.0050 1.0060 1.0070 1.0080 1.0090 1.0100 ...

...

...

-0.030

-0.020

-0.010

0.000

0.010

0.020

0.030

AP

F1 R

ef&

Inje

cted C

urr

ents

(kV

)

Va_ref_APF2 PWM_tri_APF2


Adnan TAN

178

According to these simulation results, each VSI power rating in SHAPFs of

proposed SHAPF system is calculated as;

_ _3

2 2

cap SHAPF SHAPF peak

SHAPF VSI

V IS (5.16)

From Eq. 5.16, the each converter power rating is found as,

600 16003 831

2 2SHAPF VSI

V AS kVA (5.17)

And total power rating of converters of APFs in proposed SHAPF is approximately

equal to 2 831 1662kVA kVA .

The power rating of each SHAPF module which contains both the passive

filters and APF is calculated as;

_

,32

peak SHAPF

SHAPF L L rms

IS V (5.18)

From Eq. 5.16, the power rating of each SHAPF module is find as,

16003.1000 . 1960

2SHAPF

AS V kVA (5.19)

And total power rating of converters of SHAPFs in proposed SHAPF system is

approximately equal to 2 1960 3920kVA kVA .

6. CONCLUSIONS Adnan TAN

179

6. CONCLUSIONS

Electric power systems suffer from almost all type power converters because

of the nonlinear load characteristics. The major part of these power converters

produce harmonics related with only the rectifier part of power converter. In most

case, the harmonic distortion of these converters can be reduced under the limits with

using only passive filtering systems. However, such kind of power converters whose

rectifier and inverter sections cannot be satisfactorily isolated as used in CSI-IF

generates harmonics and interharmonics related with operating frequency of inverter.

CSI-IFs are especially preferred in high power applications because of the feature of

CSI and parallel resonant tank circuit of CSI-IF in providing high resonant currents

to the furnace coil with lower converter currents. With increasing power of CSI-IFs,

the effects of CSI-IFs on the power systems are also increased. The reasons of poor

effects of high power CSI-IF on power systems are the high values of harmonic and

interharmonic currents and varying of high value harmonic and interharmonic

currents in wide frequency spectrum related with the wide operating frequency range

of CSI of the furnace power converter.

In this thesis, the time varying harmonics and interharmonics of CSI-IF and

the relation of these varying harmonics and interharmonics with the operating

frequency of CSI of furnace is firstly introduced theoretically. Secondly, these

varying harmonics and interharmonics and relation with the operating frequency of

CSI of furnace are demonstrated in the simulation of CSI-IF model created by using

the furnace parameters of a real CSI-IF. Finally, the simulation results and power

quality measurements of CSI-IF in the steel mill is compared and it is shown that the

simulation results of CSI-IF are corresponding to the power quality measurements of

CSI-IF.

In order to find solutions to these power quality problems of CSI-IF, passive

and active filtering methods are investigated. Firstly, passive filtering methods are

examined by modeling single tuned filters, C-type filters and LLCL type broad band

filters. The use of single tuned filters in PQ compensation of CSI-IF is not possible

because the CSI-IF generates time varying harmonics and interharmonics in very


180

wide frequency range. If a single tuned filter which is tuned to a frequency in range

of time harmonics and interharmonics of CSI-IF is connected to the input of CSI-IF,

the parallel resonance will occur between filter and power system when CSI-IF

generates harmonics or interharmonics below the tuning frequency of single tuned

filter. Because of these, the only solution to place a single tuned passive filter to the

input of CSI-IF is to tune the filter below the frequencies of generated harmonics and

interharmonics. However, this time single tuned filter has no effect on filtering

harmonics and interharmonics of CSI-IF because single tuned filters attenuate the

harmonics or interharmonics in the tuning frequency. In above frequencies than

tuning frequency, single tuned filter cannot show any filtering effect. Thus single

tuned filters provide solution to the reactive power requirements of CSI-IF.

The C-type filters are thought as being able to provide a solution to the

harmonics and interharmonics of CSI-IF because there are many applications of C-

type filters in interharmonic generating systems such as HVDC transmission

systems, cycloconverters and EAFs. When the C-type filters are investigated, it is

shown that reactive power rating of C-type filter must be large in order to show

lower impedance than power system and damp the harmonics and interharmonics.

When C-type filter is designed according to the reactive power ratings of CSI-IF, it is

observed that the filter has no effect on compensating harmonics and interharmonics

of CSI-IF. Because of the low reactive power rating of CSI-IF, the reactive power

rating of filter cannot be increased in order to maintain reactive power demand

limits.

The LLCL type broad band filters are another attractive passive filter type.

They can show filtering effect in wide frequency range. When this type filter is

designed for the CSI-IF, it is shown that the filter not only shows very effective

filtering performance and decrease the harmonic and interharmonic contents of CSI-

IF currents under the limits but also maintain the reactive power requirements of

CSI-IF. However, this LLCL filter causes high voltage distortion in the output

voltage of filter because of the series reactors of filter. The voltage distortion can be

decreased by decreasing the values of series reactor but this time the harmonic

filtering performance of filter is decreased. While this designed filter keeps the CSI-


181

IF current harmonic distortion under 5%, it increases the voltage distortion up to

12% level. In this condition, the furnace cannot operate in real operation because

these distorted voltage signals can cause firing problems in rectifier section of

furnace power supply. Moreover, the series reactors with high current ratings are also

a disadvantage of LLCL type broadband filter.

Because of the inadequate performances of passive filtering systems, HAPF

systems are proposed to solve the time varying harmonics and interharmonics

problems and meet the reactive power requirements of CSI-IF. The reason of

preferring HAPF is that it is not feasible to provide solution to both harmonic and

interharmonics problems, and reactive power requirements of CSI-IF with using only

pure APFs because with increasing ratings of power electronics based compensation

systems the cost of system increases incredibly.

In the first proposed HAPF topology, shunt APF and shunt passive filter

combination is used. In this topology, two identical APF modules are used in shunt

APF system because of the high compensating harmonics and interharmonics of CSI-

IF. When the simulation results of the proposed compensation system are

investigated, it is shown that the proposed system performs an adequate performance

on filtering the harmonics and interharmonics of CSI-IF and meets the reactive

power requirements of CSI-IF completely. However, in the control methods of shunt

APF modules in proposed HAPF system, effective current control cannot be

achieved so, the harmonic and interharmonic eliminating performance of APF

modules is decreased. In order to maintain regular current control, the ratings of

shunt APF system must be increased. In proposed system, it can be achieved with

two methods. The first one is the increasing of DC-link voltage of VSI and the

second one is using additional APF modules. Increasing DC-link voltage in APF

modules is not feasible because of the power ratings of power electronics devices.

Because of these, additional APF modules must be used in order to obtain the regular

current control and superior harmonic and interharmonic eliminating performance.

However, this solution increases the cost of proposed HAPF system.

In the second compensation system, SHAPF system is proposed. SHAPF

topology is formed from series connected APF with shunt passive filter. In proposed


182

SHAPF system, two identical SHAPF modules are used because the passive filter

ratings cause high APF currents in order to obtain the reactive power requirements of

CSI-IF. When the simulation results of the proposed compensation system are

investigated, it is shown that SHAPF provides better harmonic and interharmonic

filtering performance than the shunt APF - shunt passive filter combination and

meets the reactive power requirements of CSI-IF completely.

When the converter ratings of APFs in both proposed HAPF systems are

compared, it is seen that the ratings of converters in SHAPF is significantly lower

than ratings of converters in shunt APF modules because of the low DC link voltage

level. In shunt APF modules the DC link voltage must be higher from the peak value

of supply voltage in order to prevent the reactive power flow to the APF and

injecting the harmonics and interharmonics currents to the power system. However

in SHAPF topology, the DC link voltage only provides the harmonic and

interharmonics currents because the fundamental voltage of system drops to

terminals of series connected passive filters. This feature of SHAPF provides major

advantage in decreasing the ratings of converters.

In this thesis, the performance of passive and active filtering systems is

demonstrated by modeling the systems in the PSCAD/EMDTC. According to the

simulation results, the comparison table of compensation system is presented in

Table 6.1 and the following results are extracted from this study. The single tuned

passive filter and C-type passive filter are certainly not appropriate for the

compensation of the harmonic and interharmonic problems of CSI-IF. The LLCL

type broadband passive filter shows effective harmonic and interharmonic

eliminating performance but the voltage distortion at the output of filter can prevent

the proper operation of CSI-IF so, this filter is also not appropriate for the

compensation of CSI-IF. When the proposed HAPF systems are compared, both of

them presents sufficient performance in both compensating the harmonics and

interharmonics of CSI-IF and meeting the reactive power requirements of CSI-IF.

However, when the converter power ratings of filters are compared the converters in

the proposed SHAPF system has lower power ratings than the converters of shunt

APFs. The power rating of converters is directly proportional to the cost of


183

converters so the increasing power ratings of converters in APFs increases the costs

of APFs Moreover the increasing power ratings of converters also cause to increase

the losses of the converters. Therefore, the proposed SHAPF system presents more

effective compensation solution for the harmonics and interharmonics problems and

reactive power requirements of CSI-IF.

Table 6.1. Comparison of Investigated Compensation Systems

Single Tuned

Filter C-Type Filter

LLCL Type

Broad Band

Filter

ProposedHAP

F

System

Proposed

SHAPF

System

Load Current

THD% 7-15% 7-15% 7-15% 7-15% 7-15%

Source Current

THD% 7-15% 6.6-14.6% 2.5-5% 2-4.5% 1.8-3.2%

Source Voltage

THD% 2-3.5% 2-3.5% 2%< 2%< 2%<

Output Voltage

of Filter THD% - - 6.5-12% - -

Switching

Frequency - - - 3kHz 3kHz

DC Link Voltage - - - 2000V 600V

RMS Current [email protected] [email protected] [email protected]

APFs=36A

STF=37A

@31.5kV

[email protected]

Max. Peak

Current [email protected] [email protected] [email protected]

APFs=52A

STF=52.5A

@31.5kV

[email protected]

Harmonic

Compensation

Performance

Poor Poor Good Excellent Excellent

Reactive Power

Compensation

Performance

Good Good

Changes

depending on the

loading of CSI-IF

Good Good

Cost Very Low Low Medium High High

Overall

Performance Poor Poor Poor Good Excellent


184

The following remarks are suggested as future works of this thesis study;

In this thesis the modeling aim of CSI-IF demonstrate the time varying

harmonics and interharmonics of CSI-IF. Because of these, the proposed

controller of CSI-IF and start-up of CSI-IF is not analyzed in detail. The

controller stability and start-up furnace will be investigated as a future work.

The power losses, power ratings and costs of the proposed HAPF and SHAPF

systems will be investigated and compared more detailed as a future work.

185

REFERENCES

AKAGI H., 2005. Active Harmonic Filters. Proceedings of the IEEE, 93(12): 2128 –

2141.

ARAVENA P., VALLEBUONA G., MORAN L., DIXON J. and GODOY O., 2009.

Passive Filters for High Power Cycloconverter Grinding Mill Drives. IEEE

Industry Applications Society Annual Meeting, Page(s):1 – 7.

ARRILLAGA J. and BRADLEY D. A., 1985. Power System Harmonics. John Wiley

& Sons Ltd, Chichester, 335 pages.

ASIMINOAEI L., BLAABJERG F. and HANSEN S., 2007. Detection is Key -

Harmonic Detection Methods for Active Power Filter Applications. IEEE

Industry Applications Magazine, 13(4): 22 – 33.

BAAKE E. Lecture Notes: Technical Basics and Applications of Induction Furnaces.

Leibniz University of Hanover. http://www.uie.org/webfm_send/391 . Last

Access Date: 07/2011.

BENCHAITA L., SAADATE, S. and SALEMNIA, A., 1999. A Comparison of

Voltage Source and Current Source Shunt Active Filter by Simulation and

Experimentation. IEEE Transactions on Power Systems, 14(2):642 - 647.

BHATTACHARYA A., CHAKRABORTY, C. and BHATTACHARYA S., 2009.

Shunt Compensation. IEEE Industrial Electronics Magazine, 3(3): 38 – 49.

BHATTACHARYA S. and MIRZAEE H., 2010. Series Active Filter Control and

Implementation for Utility Interface of Multiple Adjustable Speed Drives

IEEE Industrial Electronics Society 36th Annual Conference, Page(s): 2663 –

2668.

BODGER P.S., IRWIN G.D. and WOODFORD D.A., 1990. Controlling Harmonic

Instability of HVDC Links Connected to Weak AC Systems. IEEE

Transactions on Power Delivery, 5(4): 2039 – 2046.

BONERT R. and LAVERS J.D., 1994. Simple Starting Scheme for A Parallel

Resonance Inverter for Induction Heating. IEEE Transactions on Power

Electronics, 9(3): 281 – 287.

BOSE B.K. 1986. Power Electronics and AC Drives. Prentice Hall, USA, 402 pages.

http://www.uie.org/webfm_send/391

186

BOYLESTAD R.L., 2007. Introductory Circuit Analysis. Prentice Hall, USA, 1156

pages.

CHANG G.W., RIBEIRO P. F., 1998. Harmonics Theory, 1998 IEEE PES Summer

Meeting, Papers of Tutorial on Harmonics Modeling and Simulation.

CHENG P., BHATTACHARYA S. and DIVAN D.M., 1998. Control of Square-

Wave Inverters in High-Power Hybrid Active Filter Systems. IEEE

Transactions on Industry Applications, 34(3): 458 – 472.

CHUDNOVSKY V., AXELROD B. and SHENKMAN A.L.,1997. An Approximate

Analysis of a Starting Process of a Current Source Parallel Inverter with a

High-Q Induction Heating Load. IEEE Transactions on Power Electronics,

12(2): 294 – 301.

DAS, J.C., 2004. Passive Filters-Potentialities and Limitations. IEEE Transactions on

Industry Applications, 40(1): 232 – 241.

DAVIES E.J., 1990. Conduction and Induction Heating, IEE Power Engineering

Series, Peter Peregrinus Ltd, London, 397 pages.

DAWSON F.P. and JAIN P., 1991. A Comparison of Load Commutated Inverter

Systems For Induction Heating and Melting Applications. IEEE Transactions

on Power Electronics, 6 (3): 430 – 441.

DEDE E.J., JORDAN J., GONZALEZ J.V., LINARES J., ESTEVE V. and MASET

E. 1991. Conception and Design of A Parallel Resonant Converter for

Induction Heating, Applied Power Electronics Conference and Exposition,

Page(s): 38 – 44.

DUGAN R.C., MCGRANAGHAN M.F. and BEATY H. W., 1996. Electrical Power

Systems Quality. McGraw Hill, New York, 265 pages.

DUGAN R.C. and CONRAD L.E., 1999. Impact of Induction Furnace

Interharmonics on Distribution Systems, IEEE Transmission and Distribution

Conference, Page(s): 791 – 796.

EPRI, 1993. Induction Heating Technology.

FUJITA H. and AKAGI H., 1991. A Practical Approach to Harmonic Compensation

in Power Systems-Series Connection of Passive and Active Filters. IEEE

Transactions on Industry Applications, 27(6): 1020 – 1025.

187

GERÇEK C. Ö., ERMIS M., ERTAS A., KOSE K. N. K. and ÜNSAR Ö, 2011.

Design, Implementation, and Operation of a New C-Type 2nd Harmonic

Filter for Electric Arc and Ladle Furnaces. IEEE Transactions on Industry

Applications, 47(4): 1545 – 1557.

GONZALEZ D. A. and MCCALL J. C., 1987. Design of Filters to Reduce Harmonic

Distortion in Industrial Power Systems. IEEE Transactions on Industry

Applications, IA-23(3): 504 – 511.

HABROUK E., DARWISH M and MEHTA M.K., 2000. Active Power Filters: A

Review, IEE Proceedings on Electric Power Applications, 147:403-413.

HALPIN S.M., 2005. Comparison of IEEE and IEC Harmonic Standards. IEEE

Power Engineering Society General Meeting 2005, 3: 2214 – 2216.

HAVA A.M. and ZUBI H., 2005. Improved Broadband Harmonic Filter Design for

Adjustable Speed Drives. International Conference on Power Electronics and

Drives Systems, 1: 298 – 303.

IEC 61000-2-1, 1990. Electromagnetic compatibility (EMC) - Part 2: Environment -

Section 1: Description of the environment - Electromagnetic environment for

low-frequency conducted disturbances and signalling in public power supply

systems.

IEC 61000-2-2, 2002. Electromagnetic compatibility (EMC) - Part 2-2: Environment

- Compatibility levels for low-frequency conducted disturbances and

signalling in public low-voltage power supply systems.

IEC 61000-2-12, 2003. Electromagnetic compatibility (EMC) - Part 2-12:

Environment - Compatibility levels for low-frequency conducted

disturbances and signalling in public medium-voltage power supply systems.

IEC 61000-3-2, 2005. Electromagnetic compatibility (EMC) - Part 3-2: Limits -

Limits for harmonic current emissions (equipment input current ≤ 16 A per

phase).


Assessment of emission limits for the connection of distorting installations to

MV, HV and EHV power systems.

188


Limits for harmonic currents produced by equipment connected to public

low-voltage systems with input current > 16 A and ≤ 75 A per phase.

IEEE Std.519, 1992. IEEE Recommended Practices and Requirements for Harmonic

Control in Electrical Power Systems.

INDUCTOHEAT INC., Induction Heating Application Note,

http://www.inductoheat.com/literature/misc_products/IHEAT-AppNote

LargeGear-2.pdf, Access Date: 07.2011.

INZUNZA R. and AKAGI H., 2005. A 6.6-kV Transformerless Shunt Hybrid Active

Filter for Installation on a Power Distribution System Power Electronics,

IEEE Transactions on Volume : 20 , Issue:4 On page(s): 893 – 90.

KARIMI-GHARTEMANI M. and IRAVANI M.R., 2001. A New Phase-Locked

Loop (PLL). Proceedings of the 44th IEEE 2001 Midwest Symposium on

System Circuits and Systems, 2001,1: 421 – 424.

KARIMI-GHARTEMANI M. and IRAVANI M.R., 2002. A Nonlinear Adaptive

Filter for Online Signal Analysis in Power Systems Applications. IEEE

Transactions on Power Delivery, 17: 617-622.

KARIMI-GHARTEMANI M., MOKHTARI H., IRAVANI M.R. and SEDIGHY,

M., 2004. A Signal Processing System for Extraction of Harmonics and

Reactive Current of Single-Phase Systems. IEEE Transactions on Power

Delivery, 19(3): 979 – 986.

KARIMI-GHARTEMANI M., IRAVANI M.R. and KATIRAEI F., 2005. Extraction

of Signals for Harmonics, Reactive Current and Network-Unbalance

Compensation. IEE Proceedings Generation, Transmission and Distribution,

152(1): 137 – 143.

KHAN I., TAPSON, J. and DE VRIES I., 2000. Frequency Control of a Current-Fed

Inverter for Induction Heating. Proceedings of the 2000 IEEE International

Symposium on Industrial Electronics, 1: 343 – 346.

LOVELESS D., 1995. An Overview of Solid State Power Supllies for Induction

Heating. Metal Producing & Processing.

http://www.inductoheat.com/literature/misc_products/IHEAT-AppNote%20LargeGear-2.pdf

http://www.inductoheat.com/literature/misc_products/IHEAT-AppNote%20LargeGear-2.pdf

189

MCGRANAGHAN M. and BEAULIEU G., 2006. Update on IEC 61000-3-6:

Harmonic Emission Limits for Customers Connected to MV, HV, and EHV.

IEEE PES Transmission and Distribution Conference and Exhibition,

Page(s): 1158 – 1161.

MERTENS A. and SKUDELNY H.C., 1991. Switching Losses in a GTO Inverter for

Induction Heating, IEEE Transactions on Power Electronics, 6(1): 93 – 99.

NASSIF A.B., WILSUN XU and FREITAS W., 2009. An Investigation on The

Selection of Filter Topologies for Passive Filter Applications. IEEE

Transactions on Power Delivery, 24(3): 1710 – 1718.

OTTO-JUNKER GmbH, Medium Frequency Induction Furnaces.

http://www.ottojunker.de, Access Date: 07.2011.

PENG F.Z., AKAGI, H. and NABAE A. 1990. A New Approach to Harmonic

Compensation in Power Systems-A Combined System of Shunt Passive and

Series Active Filters, IEEE Transactions on Industry Applications, 26(6): 983

– 990.

PENG F.Z., AKAGI H., NABAE A. and SUGAWARA S., 1989. High-Frequency

Current-Source Inverters Using SI Thyristors for Induction Heating

Applications. IEEE Transactions on Industry Applications, 25(1):172 – 180.

PENG F.Z., 1998. Application Issues of Active Power Filters, IEEE Industry

Applications Magazine, 4(5): 21 – 30.

PONWIANGKUM N. and KITTIRATSATCHA S., 2007. Switching Frequency

Control Based on Phase-Locked Loop for a Current-fed Parallel Resonant

Inverter. Power Conversion Conference, Pages(s): 157 – 161.

ROUTIMO M., SALO M. and TUUSA H., 2007. Comparison of Voltage-Source and

Current-Source Shunt Active Power Filters. IEEE Transactions on Power

Electronics, 22(2): 636 – 643.

RUDNEV V., LOVELESS D., COOK R. and BLACK M., 1999. Power Quality for

Induction Melting in Metal Production, Electric Power Research Institute,

U.S.A.

RUDNEV V., LOVELESS D., and COOK R., 2003, Handbook of Induction

Heating, Marcel Dekker Inc., New York, Basel, 777 pages.

190

RUDNEV V, 2007. Systematic Analysis of Induction Coil Failures Part 11a:

Frequency selection, Heat Treating Progress, Page(s):19-21.

RUDNEV V, 2008. Successful Induction Heating of RCS Billets, Forge Magazine,

Page(s): 15-18.

RUDNEV V, BROWN D. and DOYON G., 2008. Innovative Induction Heating

Technologies. Proceedings of Material Science & Technology, Conference &

Exhibition.

RUDNEV V, 2009. Single-Coil Dual-Frequency Induction Hardening of Gears. Heat

Treating Progress, Page(s): 9-11.

RUDNEV V., 2011. Intricacies of Computer Simulation of Induction Heating

Processes. Proceedings of the 28th Forging Industry Technical Conference,

Schaumburg.

SINGH B., AL-HADDAD K. and CHANDRA A., 1999. A Review of Active Filters

for Power Quality Improvement, IEEE Transaction on Industrial Electronics,

46: 960-971.

SIEVEKING H.A, 1940, Electrically Manufactured Steels. Journal of the Institution

of Electrical Engineers, 86(521): 401 – 423.

SRIANTHUMRONG S. and AKAGI H., 2003. A Medium-Voltage Transformerless

AC/DC Power Conversion System Consisting of a Diode Rectifier and a

Shunt Hybrid Filter. IEEE Transactions on Industry Applications, 39(3): 874

– 882.

SWAMY M. M., BISEL G. R. and ROSSITER S. L., 1995. Passive Harmonic Filter

System for Variable Frequency Drives, United States Patent, Patent Number:

5,444,609.

SWAMY M. M., 2005. Passive Techniques for Reducing Input Current Harmonics.

http://www.yaskawa.com/site/dmdrive.nsf/link2/AHUG-6J5R4A. Last

Access Date: 07/2011.

TOKUNÇ C.O., 2010. Modeling of a Shunt Active Power Filter With EPLL Based

Control Method for Mitigation of Power Quality Problems Caused by

Induction Furnaces. MSc Thesis, Cukurova University, Institute of Natural

and Applied Sciences.

http://www.yaskawa.com/site/dmdrive.nsf/link2/AHUG-6J5R4A

191

TAYJASANANT T., WANG W., LI C. and XU W., 2005. Interharmonic-Flicker

Curves, IEEE Transactions on Power Delivery, 20(2): 1017 – 1024.

TESTA A., AKRAM M.F., BURCH R., CARPINELLI G., CHANG G., DINAVAHI

V., HATZIADONIU C., GRADY W.M., GUNTHER E., HALPIN M.,

LEHN P., LIU Y., LANGELLA R., LOWENSTEI, M., MEDINA A.,

ORTMEYER T., RANADE S., RIBEIRO P., WATSON N., WIKSTON J.

and XU W., 2007. Interharmonics: Theory and Modeling, IEEE Transactions

on Power Delivery, 22(4): 2335 – 2348.

TREMAYNE J. F, 1983. Impedance and Phase Balancing of Mains – Frequency

Induction Furnaces, IEE Proceedings on Electric Power Applications,

130(3):161 – 170.

UÇAK O., 2010, Design and Implementation of A Voltage Source Converter Based

Hybrid Active Power Filter, MSc Thesis, Middle East Technical University,

The Graduate School Of Natural And Applied Sciences.

WEBER A., CARROLL E. and FRECKE M., 2002. IGCTs for Induction Heating,

International PCIM Conference.

WIN P.P, 2008. Harmonic Mitigation Techniques in Industrial Power System,

GMSARN International Conference on Sustainable Development: Issues and

Prospects for the GMS.

TANGTHEERAJAROONWONG W., HATADA T., WADA K. and AKAGI H.,

2007. Design and Performance of a Transformerless Shunt Hybrid Filter

Integrated Into a Three-Phase Diode Rectifier. IEEE Transactions on Power

Electronics. 22(5): 1882 – 1889.

YACAMINI, R., 1996. Power System Harmonics: Part 4 – Interharmonics, Power

Engineering Journal, 10(4):185 – 193.

XIAO Y., ZHAO J. and MAO S., 2004. Theory for the Design of C-type Filter. 11th

International Conference on Harmonics and Quality of Power, Page(s): 11 –

15.

ZINN S. and SEMIATIN S, 2002, Elements of Induction Heating Design Control

and Applications, ASM International, USA, 323 pages.

192

ZUBI H.M., DUNN R.W., ROBINSON F.V.P. and EL-WERFELLI M.H., 2010.

Passive Filter Design Using Genetic Algorthims for Adjustable Speed Drives.

IEEE Power and Energy Society General Meeting, Page(s): 1 – 7.

193

BIOGRAPHY

Adnan Tan was born in Adana, Turkey in 1985. He received his B.Sc. in

Electrical and Electronics Engineering Department from Çukurova University,

Adana, Turkey. He has been working as Research Assistant in Electrical and

Electronics Engineering Department of Çukurova University since 2008. His

research interests include electrical power quality and utility applications of power

electronics.

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