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Towardsafullypackagedhigh-performanceRFsensorfeaturingslottedphotoniccrystalwaveguides

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DOI:10.1117/12.2213557

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Towards a Fully Packaged High-performance RF Sensor Featuring Slotted Photonic Crystal Waveguides

Chi-Jui Chunga, Harish Subbaramanb, Xingyu Zhanga, Hai Yana, Jingdong Luoc, Alex K.-Y. Jenc, Robert L. Nelsond, Charles Y.-C. Leed, Ray T. Chena

aMicroelectronics Research Center, Department of Electrical and Computer Engineering, The University of Texas at Austin, Austin, Texas 78758, USA; bOmega Optics, Inc., 8500 Shoal Creek

Blvd,Building 4, Suite 200, Austin, Texas 78759, USA; cDepartment of Materials Science and Engineering, University of Washington, Seattle, Washington 98195, USA; dAir Force Research

Laboratory at Wright Patterson, Dayton, Ohio 45433, USA.

ABSTRACT

A low loss and high sensitivity X-band RF sensor based on electro-optic (EO) polymer filled silicon slot photonic crystal waveguides (PCW) and bowtie antenna is proposed. By taking advantage of the slow light enhancementt in the PCW(>20X), large EO coefficient of the EO polymer(r33>200pm/V), as well as significant electric field enhancement of bowtie antenna on silicon dioxide substrate(>10000X), we can realize a large in-device EO coefficient over 1000pm/V so as to realize a high performance RF wave sensor. In addition, on-chip Mach-Zender interferometer (MZI) layout working under push-pull configuration is adopted to further increase the sensitivity of the sensor. Furthermore, inverse taper couplers and slotted photonic crystal waveguides are carefully designed and discussed in this paper to reduce the insertion loss of the device so as to increase the device signal-to-noise ratio. The minimum detectable electromagnetic power density is pushed down to 2.05 mW/m2, corresponding to a minimum sensing electric field of 0.61 V/m. This photonic RF sensor has several important advantages over conventional electronics RF sensors based on electrical scheme including high data throughput, compact in size, and great immunity to electromagnetic interference (EMI).

Keywords: radio frequency detection, electro-optic modulation, microwave photonics, slow light effect, photonic crystal waveguides, bowtie antenna, inverse taper couplers

1. INTRODUCTION Radio Frequency (RF) Electromagnetic (EM) field sensors have substantial applications for communication and sensing fields[1-5] including electromagnetic compatibility measurements, radio frequency (RF) integrated circuit testing, environmental electromagnetic interference analysis, electrical field monitoring in medical apparatuses, ballistic control, as well as detection of directional energy weapon attack. Conventional EM wave sensors based on electrical schemes normally require several discrete electronics components and large active high speed conductive probes which disturb the EM waves to be measured and make the device bulky as well. Recently, there has been growing interest in the field of RF photonics. Processing and transmission of RF signals by photonic systems offer significant advantages in terms of data throughput, the size, weight, and power (SWaP) requirements, and immunity to electromagnetic interference (EMI) [6]. Therefore, integrated photonic sensing of electromagnetic field devices in which optical signal is modulated by RF signals coupled by an antenna has been developed [7-9]. These antenna coupled modulators have some additional advantages including overcoming the high transmission loss at high frequencies, eliminating extra electronics compared to systems with separate antenna and optical modulator units, and improving the sensitivity without sacrificing the bandwidth[10].

Silicon-on-insulator (SOI) platform has attracted researchers’ attention as a platform for integration of photonic devices and complementary metal–oxide–semiconductor (CMOS) electronics [11] due to high refractive index and high transparency of silicon at communication wavelength range and CMOS process compatibility. The centro-symmetric crystal structure of silicon, however, eliminates any efficient linear electric field modulation, leaving the free carrier plasma dispersion effect as the choice for most the reported high speed silicon modulators [12-14]. Nevertheless, for the plasma dispersion based silicon modulators, the modulation bandwidth is intrinsically limited by the lifetime of the free carriers [15]. On the other hand, EO polymers as the active material in optical modulation devices have shown several advantages [16] over III-V, lithium niobate, and silicon materials including 1) large EO coefficient as high as 200 pm/V

Terahertz, RF, Millimeter, and Submillimeter-Wave Technology and Applications IX, edited by Laurence P. Sadwick, Tianxin Yang, Proc. of SPIE Vol. 9747, 97471V

© 2016 SPIE · CCC code: 0277-786X/16/$18 · doi: 10.1117/12.2213557

Proc. of SPIE Vol. 9747 97471V-1

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Proc. of SPIE Vol. 9747 97471V-2

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Proc. of SPIE Vol. 9747 97471V-3

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W1.21 W1.28W1.21i

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Figure 4. (a)Schematic of the group index taper between regular slot waveguide and slot PCW; (b) Waveguide widths and group indices in the designed group index taper.

Figure 5. Group index at different wavelength in a band engineered slot PCW.

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2.3 Design of Inverse Taper Coupler

Our group have previously designed and demonstrated a single step, through-etched sub-wavelength grating (SWG) coupler, which can be patterned together with photonic components without any additional patterning steps [22]. However, due to difficulties of patterning the sub-100nm grating structures, the typical loss we can achieved is -7 to -8 dB per grating coupler. Adopting inverse taper coupler can reduce the coupling loss down to <1 dB as reported [23]. In addition, grating couplers are inherently bandwidth-limited device with 3 dB optical bandwidth of tens of nanometers. The inverse taper typically has hundreds of nanometer of 3 dB bandwidth [23]. Hence, an inverse taper coupler design is proposed here for efficiently coupling light into the input silicon waveguide.

The schematic drawing and the geometrical parameters of the inverse taper coupler are shown in figure 6, which are optimized using commercial available FIMMWAVE software from PhotonDesign. The structure width is adiabatically tapered from 450 nm (device end) to 160 nm (fiber end). An SU-8 polymer waveguide is designed and overlaid on the silicon waveguide so that the 3 μm x 3 μm cross-section area matches with the mode area of input lensed fiber. A top view simulation result showing efficient coupling of light from the SU8 polymer input waveguide into a single mode silicon waveguide with an inverse taper of length L and a tip width of 160nm is shown in figure 7. The coupling loss per inverse taper is simulated to be less than 0.5 dB.

Figure 6. Schematic drawing and designed parameters of the inverse taper coupler

Figure 7. Top view simulation result showing mode coupling from a 3um x 3um SU8 polymer waveguide into a 160nm tip adiabatic

silicon taper and single mode waveguide

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0(mm)

2.4 Design of bowtie antenna and the antenna substrate consideration The bowtie antennas concentrate the EM wave at certain range of designed frequency and provides a strong near-field electrical enhancement. Not only is the metallic structure of the bowtie important for the resonant antenna performance, but also the substrate material properties plays a pivotal role when interacting with the EM wave. Two common materials used as substrate for antennas are silicon dioxide and SOI. The previously optimized [21] gold bowtie parameters are used here for comparing the two different substrates. The bowtie structure, shown in figure 8, consists of a conventional bowtie antenna with capacitive extension bars attached to the apex points of the bowtie. The bowtie has an arm length of 4.5mm, and a flare angle of 60 degrees. The extension bars have a length, b=300μm, width, w=10μm and a feed gap, g=10μm. The thickness of the gold film is chosen to be t=5μm.

Figure 8. Geometrical parameter of bowtie antenna with extension bars

First, we used finite element method (HFSS from ANSYS, Inc.) to simulate the S11 parameter (return loss) of the optimized structure mentioned above. The thickness of the substrate are 1 mm for the silicon dioxide substrate and 1 mm silicon handle layer and 3 μm buried oxide layer for the SOI. The simulation model in HFSS is shown in figure 9. The dielectric constant of are set as 4 for silicon dioxide and 13.9 for silicon handle layer respectively for frequency from 1-30 GHz. The loss tangent is set as 0.015 for silicon and 0.0036 for silicon dioxide. We also setup a perfect impedance matching condition of 50 ohms for the two terminal of the bowtie for the range of frequency. The simulated S11 parameters of the antenna for silicon dioxide and SOI substrates from 1-30 GHz are shown in figure 10.

Figure 9. Trimetric view of the bowtie antennas in HFSS

x y

z

α l

g wb

yx

z yt

(b)

(c)

(a)

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27) -20

-25

on SiO2on SOI

3 6 9 12 15 18 21 24 27 30

Frequency (GHz)

Figure 10. Simulation results of the bowtie antennas on SiO2 substrate and on SOI substrate

We can clearly see from S11 parameter versus frequency plot that the resonance frequency shift to lower region of 6.34 GHz from designed 8 GHz when using SOI as substrate (high dielectric constant) since resonance frequency. ∝ 1/ For the resonance frequency, the bowtie antenna on silicon dioxide has lower drop of -13.22 dB compared to -8.58 dB on silicon. That means that bowtie antenna on silicon dioxide has higher antenna gain when working as a transmitting antenna if we assume the absorption loss is negligible. We also notice that the bowtie on silicon has many undesired transmission/reflection around 15-30 GHz. The abnormal transmission/reflection signal will affect the signal detection for our designed range of frequency. When we change the thickness of the substrate in the simulation, the nadir shifts. Therefore, it should results from the interference of the multiple reflection and transmission RF waves (Fabry–Pérot-like effect) of the interfaces.

3. FABRICATION The fabrication flow of the proposed RF sensor is illustrated in figure 11. The fabrication starts with a silicon on silicon dioxide wafer with a 220 nm-thick top silicon and a 0.5 mm-thick buried oxide layer, as shown in figure 11(a). Next, the PCW is patterned by e-beam lithography (EBL) and all the top silicon region except the PCW area are etched away by reactive-ion etching (RIE). Then, a two-step ion implantation is performed on the sample, as shown in figure 11(b). After performing a rapid thermal annealing process to activate the dopants and annihilate the induced defects, we deposit a thin gold seed layer for electroplating, as shown in figure 11(c). Then, a buffer mask for the bowtie antenna is patterned on a 10 μm-thick AZ9260 photoresist using photolithography. Next, a 5 μm-thick gold film is electroplated using Techni-Gold 25ES electrolyte under a constant current of 8 mA. The AZ9260 buffer mask and gold seed layer are finally removed by lift-off and wet etching, as shown in figure 11(d). A 3 μm X 3 μm SU-8 layer for lensed fiber end-fire coupling is patterned using photolithography, as shown in figure 11(e). After cleaving, the EO polymer, SEO250, is formulated and infiltrated into the holes and the slot of the silicon PCW region by spin-coating, as shown in figure 11(f), followed by baking at 80oC in vacuum oven for 12 hours. To align the polymer chromophore and activate the EO effect of the EO polymer, the EO polymer is poled in a constant electric field of 100 V/μm at the EO polymer glass transition temperature of 130 oC, in which the bowtie antenna serves as poling electrodes. The two arm of the MZI are poled in opposite directions so that we can operate the MZI in a push-pull configuration, as shown in figure 12.

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(d) i0

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esSeesSe

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silicon dioxide substrate enables a more sensitive, more energy efficient device RF sensor. Bowtie antenna on silicon dioxide also enables better immunity to undesired noise compared to bowtie on buried oxide layer of the SOI.

Figure 14. Measurement results of the bowtie antennas

The designed inverse taper coupler is fabricated on 2 cm X 2 cm SOI wafers with buried oxide of 3 μm, and the testing layout is shown in figure 15. Seven waveguides with different silicon waveguide length are fabricated on the same chip. Two s-bends with radius of curvature 100 μm were added on each waveguide in order to suppress the stray light collected by the output fiber

Figure 15. Testing layouts of the inverse taper coupler: silicon waveguide (in white) and SU-8 overlaid layer (in orange)

The silicon waveguide is patterned using EBL and RIE. The overlaid SU-8 structure is patterned using Karl Suss MA-6 photo aligner. The SU-8 used here is MicroChem SU-8 2002 and the photolithography conditions such as spin-speed, pre-bake condition, exposure dosage, post-bake condition, and develop time are firstly optimized on silicon wafer and then inspected before we started to pattern on the silicon waveguide samples. The scanning electron microscope (SEM) images and the surface profile inspection results of optimized SU-8 structure are showed in figure 16. The optimized SU-8 pattern is 3.048 μm in width and 3.06 μm in height with side wall angle larger than 85o which are very close to the designed parameters. After the SU-8 photolithography test, we started to

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Figure 16. (Left)SEM images of the patterned SU-8 structure (Right) depth of the SU-8 structure

Figure 17. (Left)Alignment results of the SU-8 layer and silicon layer (Right) Alignment marks with vernier scales, showing the

good alignment between silicon layer and SU-8 layer.

The preliminary insertion loss measurement results for the waveguide which include two inverse taper coupler, silicon single mode waveguide loss (6628 μm long), and SU-8 waveguide loss (3560 μm long) is -4.264 dB. Using -3.6 dB/cm propagation loss for silicon single mode waveguide [25], and -1 dB/cm for SU-8 waveguide, the best result for one inverse taper is -0.74 dB, which will reduce the coupling loss from ~ -15 dB for 2 grating couplers to ~-1.5 dB for 2 inverse taper couplers.

5. DEVICE PERFORMANCE SPECULATION We can speculate the performance of the proposed integrated photonic EM wave sensor based on the results of previously developed sensor in [20] and the verification results mentioned above. The previous SOH device for the photonic of EM wave sensing of electromagnetic field at 8.4 GHz was experimentally demonstrated, with a minimum detectable electromagnetic power density of 8.4 mW/m2, corresponding to a minimum detectable electric field of 2.5 V/m [20]. Since we are using the on-chip MZI structure with push-pull configuration, the minimum detectable electromagnetic power density can be halved theoretically. Also benefiting from the γ33 value improvement from previous 135 pm/V (SEO 125) to 230 pm/V (SEO 250), if the modulation interaction length is fixed and the slow light effect remains, we should achieve 1.7X higher sensitivity. The antenna on silicon dioxide also provide 1.2X input enhancement for the EO modulation. By combing these three effect, the minimum detectable electromagnetic power density should be reduced to 2.05 mW/m2, corresponding to a minimum detectable electric field of 0.61 V/m. Besides, by replacing the low loss and insulating silicon dioxide substrate, the antenna gain of the bowtie antenna should increase which will result in further reduction of the minimum detectable electromagnetic power density. Also the optical insertion loss is reduced by ~ 4.5 dB due to the introduction of inverse taper coupler.

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6. CONCLUSIONS In summary, we have proposed a highly integrated, low loss, and high sensitivity electromagnetic wave sensor based on electro-optic (EO) polymer filled slot photonic crystal waveguides (PCW) and bowtie antenna is proposed. Design of key components is discussed and preliminary experiments are performed in order to demonstrate these key building blocks. Based on the experimental data gathered, and comparing those to our previously developed sensor devices, we speculate to achieve a minimum detectable electromagnetic power density of 2.05 mW/m2, corresponding to a minimum detectable electric field of 0.61 V/m at 8.4 GHz. The miniaturized and integrated device has several important advantages including high data throughput, reduced SWaP requirements, and immunity to electromagnetic interference (EMI).

ACKNOWLEDGEMENT

The authors would like to acknowledge the Air Force Research Laboratory (AFRL) for supporting this work under the Small Business Innovation Research (SBIR) program (Contract No. FA8650-14-C-5006), monitored by Dr. Robert Nelson and Dr. Charles Lee.

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