MICROSTRIP STRUCTURE DESIGN FOR BETTER

DIPLEXER PERFORMANCE

Bachelor of Engineering

(Electrical and Electronics Engineering)

Design and Development Project 2

SHEHAN DE FONSEKA

2012

i

DECLARATION

I hereby declare that this report entitled “Microstrip structure design for a better

diplexer performance” is the result of my own project work except for the quotations

and citations which have been duly acknowledged. I also declare that it has not been

previously or concurrently submitted for any other degree at Swinburne University of

Technology (Sarawak Campus).

..............................................

Name: Shehan De Fonseka

Date: 14th May 2012

ii

Acknowledgement

First and foremost, I would like to thank my Project Supervisor, Mr.Ting Sie King for

his valuable support and guidance in this endeavour.

Special Thanks goes to my partner, Abdul Rafay, for his effort throughout the semester.

I would also like to extend my gratitude to Chamila Premarathne, Kennedy Mrema,

Aaron Tan for their support during the fabrication process.

Last but not least, I would like to thank my parents, all my loved ones and especially my

uncle Dr. Brian N DeFonseka for the motivation and strength they have given me in

every possible way.

iii

Abstract

A Diplexer design is proposed for the GSM frequency range with suppressed 2nd

Harmonics with the use of a Dual Band Wilkinson Power Divider. The diplexer

combines two bandpass filters which will allow two different pass bands to pass

through. The design is fabricated on Microstrip and measured.

It further looks in to the aspect of using a dual band Wilkinson power divider instead of

a common port junction in the design of a diplexer. There have been many prior designs

of dual band Wilkinson power dividers, a one method has been used in this thesis to

proceed with designing the diplexer.

The simulations have been carried out on Sonnet EM and Agilent ADS 2011 softwares.

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Table of Content

1 Chapter 1: Introduction ............................................................................................. 1

1.1 Project Background ........................................................................................... 1

1.2 Problem Statement ............................................................................................ 1

1.3 Project Objectives ............................................................................................. 1

1.4 Work distribution .............................................................................................. 2

2 Chapter 2: Literature Review .................................................................................... 3

2.1 Introduction ....................................................................................................... 3

2.2 S parameters ...................................................................................................... 3

2.3 Quarter Wavelength Transmission line ............................................................ 5

2.4 Microstripline .................................................................................................... 6

2.4.1 Coupled Microstrip Lines ........................................................................... 8

2.5 Power Dividers ................................................................................................. 9

2.5.1 Three Port Network .................................................................................. 10

2.5.1.1 T Junction Power Divider .................................................................. 11

2.5.1.2 The Lossless Divider ......................................................................... 11

2.5.1.3 The Resistive power Divider ............................................................. 12

2.5.1.4 The Wilkinson Power Divider ........................................................... 13

2.5.1.4.1 Even and Odd mode Analysis ....................................................... 15

2.5.1.4.2 Even Mode .................................................................................... 16

2.5.1.4.3 Odd Mode ...................................................................................... 17

2.6 Few Important terms for a Wilkinson power divider ..................................... 18

2.7 Dual Band Wilkinson Power Divider ............................................................. 19

2.8 Band Frequencies ............................................................................................ 19

2.9 Compact Dual-band Power Divider using Branch lines and Resistors .......... 20

3 Chapter 3: Design Process and Simulation ............................................................. 22

3.1 RT Duroid 6010 .............................................................................................. 22

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3.2 Designing Compact Dual-band Power Divider using Branch lines and

Resistors ...................................................................................................................... 22

3.2.1 Simulation and Results ............................................................................. 24

3.2.2 Agilent ADS Momentum .......................................................................... 26

3.2.2.1 Working on ADS layout .................................................................... 26

3.2.2.2 ADS Component Library .................................................................. 27

3.2.3 Filters ........................................................................................................ 27

3.2.3.1 Low Pass Filter .................................................................................. 27

3.2.3.2 High Pass Filter ................................................................................. 28

3.2.4 The Diplexer Formation ............................................................................ 29

3.2.4.1 Further Optimization ......................................................................... 31

3.2.5 Optimized Results ..................................................................................... 33

3.2.6 Metclad ..................................................................................................... 34

3.2.7 Simulation for Metclad ............................................................................. 34

3.2.7.1 Dilpexer Simulation ........................................................................... 34

3.2.7.2 Simulation of the Low Pass Filter ..................................................... 35

4 Chapter 4: Fabrication and Measurement ............................................................... 36

4.1 Components Used ........................................................................................... 36

4.1.1 Vishay 180R Resistor (0402) .................................................................... 36

4.1.2 Panasonic ERJ3GEYJ101V (0603) .......................................................... 36

4.2 Test Fabrication on FR4 ................................................................................. 37

4.3 Toner Transferring .......................................................................................... 38

4.4 Fabrication on Metclad ................................................................................... 39

4.5 Measured Results for the Low Pass Filter ...................................................... 40

4.6 Measured Results for the Diplexer ................................................................. 41

4.7 Conclusion ...................................................................................................... 43

4.8 Recommendations & Future Work ................................................................. 43

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References ....................................................................................................................... 44

Appendices ...................................................................................................................... 45

Table of Figure

Figure 2.1- Two Port Network...............................................................................3

Figure 2.2- Microstripline......................................................................................6

Figure 2.3- Field lines of Microstrip.......................................................................7

Figure 2.4 - Coupled Microstriplines.......................................................................8

Figure 2.5 - Schematic Block Diagram of a n+1 port combiner and divider.................9

Figure 2.6 -Transmission Line model of lossless divider...........................................11

Figure 2.7 - A 3db Resistive power divider.............................................................12

Figure 2.8 – Wilkinson Divider in microstrip format and the transmission line format.14

Figure 2.9 - Wilkinson Power Divider Schematic.....................................................14

Figure 2.10- Wilkinson Divider in Normalized and symmetric form..........................15

Figure 2.11- Even mode Excitation........................................................................16

Figure 2.12-Odd mode Excitation..........................................................................17

Figure 2.13 - Power Divider Topology...................................................................20

Figure 3.1 - Schematic Diagram............................................................................24

Figure 3.2 - Simulation Results.............................................................................25

Figure 3.3- Modified Layout of the power divider…………………………………….26

Figure3.4- Low Pass Filter & Performance.............................................................27

Figure 3.5- Highpass filter & Performance.............................................................28

Figure 3.6- Filters connected in 90 degrees.............................................................29

Figure 3.7- Filters Connected in 60 degrees............................................................30

igure 3.8- Filters Connected in 45 degrees..............................................................30

Figure 3.9- Low Pass filter connected in 90 degrees with an extended transmission line

..........................................................................................................................31

Figure 3.10- Connected by the 14mm Transmission Line.........................................31

Figure 3.11- With two extra set of grooves at 0.2mm apart from each other...............32

Figure 3.12- Optimized Design Layout...................................................................33

Figure 3.13-Optimized Design Simulation Results...................................................33

Figure 3.14- Metclad Simulation of Diplexer..........................................................34

vii

Figure 3.15- Metclad Simulation for the Low Pass Filter.........................................35

Figure 4.1- Impedance vs Frequency curve for Vishay Thin Film Resistor.................36

Figure 4.2- Impedance vs Frequency curve for Panasonic Thick Film Resistor...........36

Figure 4.3 -Diplexer Fabricated on FR4.................................................................37

Figure 4.4 - The Fabricated lines of 0.35mm of the power divider............................37

Figure 4.5 – Divider achieved by a toner transfer.................................................... 38

Figure 4.6- Another Toner Transfer attempt........................................................... 38

Figure 4.7 - Fabricated Diplexer on Metclad...........................................................39

Figure 4.8 - Awry lines achieved by toner transfer on Metclad..................................39

Figure 4.9 -Measured Results for the Low Pass Filter..............................................40

Figure 4.10- Taking Measurements.........................................................................40

Figure 4.11-Results for the low pass Filter..............................................................41

Figure 4.12-Results for the high pass filter..............................................................41

Figure 4.13- The close up of the fabricated Diplexer................................................42

List of Tables

Table 1.1-Work Distribution ............................................................................................ 2 Table 2.1- GSM Frequencies ......................................................................................... 19 Table 3.1 Low Pass Filter .............................................................................................. 28 Table 3.2 High Pass Filter .............................................................................................. 29

1

1 CHAPTER 1: INTRODUCTION

1.1 PROJECT BACKGROUND

In an era where advanced technology in Communication industry has revolutionized the

whole world, the bar is set very high for any new product that enters the market. Mobile

Devices, Wireless Devices and Computers are all getting compact and affordable. But

with extra compactness can sometimes be a costly bargain in terms of the quality

achieved.

The occurrences of 2nd harmonics hinder the performance of a Microwave Diplexer to a

considerable extent. Furthermore losses such as return loss and insertion loss need to be

dealt with in the power distribution of the diplexer.

This thesis looks in to simulation and fabrication of Microwave Diplexer with better

harmonic performances.

1.2 PROBLEM STATEMENT

A common port junction is essential to design a diplexer. its performance and compact

size of the structure are of great concern in the power divider design.

1.3 PROJECT OBJECTIVES

• Study about Junctions and Power Dividers to join the two passband filters to

form a diplexer.

• Study about methods to reduce the effect of 2nd harmonics that occur in a

Diplexer along with high return loss and low insertion loss.

• Design and simulate a diplexer.

• Fabricate and Measure

2

1.4 WORK DISTRIBUTION

Group Member Assigned Part

Shehan De Fonseka Study and design a suitable power divider

for the Diplexer design.

Combine the Filters to form a Diplexer

and simulate for optimum performance.

Abdul Rafay Study and design bandpass filters for the

diplexer with good suppression of 2nd

harmonics. Table 1.1-Work Distribution

3

2 CHAPTER 2: LITERATURE REVIEW

2.1 INTRODUCTION

The world of microwave engineering has been evolving since its inception. Whilst

working with waves of which frequencies vary in a range of 300MHz to 300GHz of the

Electromagnetic spectrum has been a different discipline for years, it has contributed

greatly to the development of communication technology, especially with the proposals

to use microwave technology in cellular phone systems in 1970(Pozar 2005). Four

decades later, it has become an integral part of our daily lives.

The microwave diplexer is such a component that has become an integral part of the

communication industry. Thus it is important to achieve better performance while

maintaining its compact size.

2.2 S PARAMETERS

S parameters refer to the Scattering Matrix, a concept developed in the decade 1950

onwards. The Scattering Matrix is a mathematical model which enables us to

understand how RF energy propagates through a multiport network. For a RF signal that

is incident on one port, a fraction of the signal reflects while the other scatters & exit the

other ports.

S-parameters are complex mainly because the network changes both the magnitude &

the phase of the particular signals. S parameters describe the response of a N-port

network ti voltage signals at different ports. S parameters are written as “Sij” where the

first subscript (i) refers to the output port while the second subscript (j) refers to the

input port. For a N port network, there will be N2 number of parameters.

A two port network is shown below.

Figure 2.1- Two Port Network

4

𝑏1 = 𝑆11𝑎1 + 𝑆12𝑎2

𝑏2 = 𝑆21𝑎1 + 𝑆22𝑎2

a1 & a2 refer to the normalized incident wave variables while b1 & b2 refer to the

normalized reflected wave variables.

𝑎1 =𝑉1 + 𝐼1𝑍0

2�𝑍0=

𝑣𝑜𝑙𝑡𝑎𝑔𝑒 𝑤𝑎𝑣𝑒 𝑖𝑛𝑐𝑖𝑑𝑒𝑛𝑡 𝑜𝑛 𝑝𝑜𝑟𝑡 1�𝑍0

𝑎2 =𝑉2 + 𝐼2𝑍0

2�𝑍0=

𝑣𝑜𝑙𝑡𝑎𝑔𝑒 𝑤𝑎𝑣𝑒 𝑖𝑛𝑐𝑖𝑑𝑒𝑛𝑡 𝑜𝑛 𝑝𝑜𝑟𝑡 2�𝑍0

𝑏1 =𝑉1 − 𝐼1𝑍0

2�𝑍0=

𝑣𝑜𝑙𝑡𝑎𝑔𝑒 𝑤𝑎𝑣𝑒 𝑟𝑒𝑓𝑙𝑒𝑐𝑡𝑒𝑑 𝑓𝑟𝑜𝑚 𝑝𝑜𝑟𝑡 1�𝑍0

𝑏2 =𝑉2 − 𝐼2𝑍0

2�𝑍0=

𝑣𝑜𝑙𝑡𝑎𝑔𝑒 𝑤𝑎𝑣𝑒 𝑟𝑒𝑓𝑙𝑒𝑐𝑡𝑒𝑑 𝑓𝑟𝑜𝑚 𝑝𝑜𝑟𝑡 2�𝑍0

By squaring the above equations for a and b gives:

|𝑎1|2 = 𝑖𝑛𝑐𝑖𝑑𝑒𝑛𝑡 𝑝𝑜𝑤𝑒𝑟 𝑜𝑛 𝑡ℎ𝑒 𝑖𝑛𝑝𝑢𝑡 𝑜𝑓 𝑡ℎ𝑒 𝑛𝑒𝑡𝑤𝑜𝑟𝑘

|𝑎2|2 = 𝑟𝑒𝑓𝑙𝑒𝑐𝑡𝑒𝑑 𝑝𝑜𝑤𝑒𝑟 𝑜𝑛 𝑡ℎ𝑒 𝑜𝑢𝑡𝑝𝑢𝑡 𝑜𝑓 𝑡ℎ𝑒 𝑛𝑒𝑡𝑤𝑜𝑟𝑘

|𝑏1|2 = 𝑟𝑒𝑓𝑙𝑒𝑐𝑡𝑒𝑑 𝑝𝑜𝑤𝑒𝑟 𝑜𝑛 𝑡ℎ𝑒 𝑖𝑛𝑝𝑢𝑡 𝑜𝑓 𝑡ℎ𝑒 𝑛𝑒𝑡𝑤𝑜𝑟𝑘

|𝑏2|2 = 𝑖𝑛𝑐𝑖𝑑𝑒𝑛𝑡 𝑝𝑜𝑤𝑒𝑟 𝑜𝑛 𝑡ℎ𝑒 𝑜𝑢𝑡𝑝𝑢𝑡 𝑜𝑓 𝑡ℎ𝑒 𝑛𝑒𝑡𝑤𝑜𝑟𝑘

Therefore, the S parameters coefficients definitions can be given as:

𝑆11 =𝑏1

𝑎1�

𝑎2=0𝑆12 =

𝑏1

𝑎2�

𝑎1=0

𝑆21 =𝑏2

𝑎1�

𝑎2=0𝑆22 =

𝑏2

𝑎2�

𝑎1=0

Where an = 0 refers to a perfect impedance match at port n.

5

The parameter S11 and S22 are also termed the reflection coefficients while S12 and S21

are termed as transmission coefficients. In general, the S-parameters are complex

values. Thus, it is convenient to express them in terms of amplitudes and phases where

amplitudes are given in decibels (dB) which are shown as:

20 log|𝑆𝑚𝑛| 𝑑𝐵𝑚, 𝑛 = 1, 2

For filter characterization, the two parameters can be defined as:

𝐿𝐴 = −20 log|𝑆𝑚𝑛| 𝑑𝐵𝑚 𝑛 = 1, 2 (𝑚 ≠ 𝑛)

𝐿𝑅 = 20 log|𝑆𝑛𝑛| 𝑑𝐵 𝑛 = 1, 2

(where LA refers to the insertion loss between ports n and m while LR refers to the return

loss at port n.)

In transmission line, the reflection coefficient is defined as,

Γ =𝑍𝐿 − 𝑍0

𝑍𝐿 + 𝑍0= 𝑆𝑛𝑛 𝑛 = 1,2

(where ZL refers to the input impedance while Z0 refers to the terminal impedance).

Given this, the voltage standing wave ratio( VSWR) and the return loss can be related

by,

𝑉𝑆𝑊𝑅 =1 + |𝑆𝑛𝑛|1 − |𝑆𝑛𝑛|

(Jia-Sheng Hong and MJ. Lancaster, 2001)

2.3 QUARTER WAVELENGTH TRANSMISSION LINE

Quarter wave transmission line or more commonly known as “Quarter Wave

transformer” is used to match two transmission lines with different impedances. A

transmission line length of usually 𝜆4 is used. The impedance of the quarter wavelength

transmission line is the square root of the multiplication of the matching

impedances��𝑍1𝑍2�.

6

2.4 MICROSTRIPLINE

Microstrip is one of the most popular planar transmission structures used in Microwave

circuit designing. It can be manufactured with photolithography technology and can

easily be integrated with other passive or active microwave components.

Microstrip transmission line consists of a thin layer of conductor printed on top of a

grounded dielectric substrate.

The important parameters related to the microstripline structure are conductor width

(w), substrate thickness (h) , relative permittivity (ϵr) & thickness of the conductor line.

The thickness of the top conductor line is of less importance thus can be neglected. This

is mainly due to the thickness which is usually in the range of 10-20µm’s.

As the thickness of the conductor line increases, the number of electric field lines on the

side planes of the top conductor layer would increase. This will affect the characteristic

impedance of Z0 & the Effective dielectric constant (ϵeff) of the microstripline. The

Effective dielectric constant will be explained in due course.

The microstripline cannot support a pure TEM wave. In actuality, what a microstripline

can accommodate is a hybrid TM-TE wave. However in most of the practical

applications the dielectric substrate is very thin thus giving the electric fields a quasi

TEM nature.

The analysis for determining the microstripline characteristics impedance & the

propagation constant can be divided in to two groups, the quasi-static analysis method

& full wave analysis method. Full wave analysis method considers a hybrid mode of

propagation thus making the solutions provided analytically complex. This method

shows the variation of characteristic impedance and phase velocity of a microstripline

with frequency.

Figure 2.2- Microstripline

7

Quasi-static methods consider a pure TEM mode of propagation in microstripline. This

allows us to obtain a good approximation for the phase velocity (Vp), the propagation

constant (β) & the characteristic impedance ( Z0) .

𝑉𝑝 = 𝐶

�𝜖𝑒𝑓𝑓 (2.1)

𝛽 = 𝑘0 �𝜖𝑒𝑓𝑓 (2.2𝑎)

𝑘0 =2𝜋𝑓

𝑐 (2.2𝑏)

Effective Dielectric Constant (𝜖𝑒𝑓𝑓)

Since the conductor layer of microstripline lies between two different dielectric zones,

air & the dielectric substrate, the parameter Effective dielectric constant is introduced.

1 ˂ ϵr ˂ 𝜖𝑒𝑓𝑓

Microstrip has some of its field lines in the dielectric region, concentrated between the

conductor strip & the ground plane, while some fraction of it occupies the air region

above. This parameter depends on the substrate thickness (h), & the conductor width

(w).

𝜖𝑒𝑓𝑓 = 1 + 𝜖𝑟

2+

𝜖𝑟 − 12

1

�1 + 12𝑤ℎ

(2.3)

Characteristic impedance of a microstrip line can be calculated using,

Figure 2.3- Field lines of Microstrip

8

Z0 = 60

�ϵeffln �

8hw

+ w4h

� for wh

≤1 (2.4)

𝑍0 =120𝜋

�𝜖𝑒𝑓𝑓 �𝑤ℎ + 1.393 + 0.667 ln �𝑤

ℎ + 1.444�� 𝑓𝑜𝑟

𝑤ℎ

≥ 1 (2.5)

whenever the relative permittivity & the characteristic impedance has been specified,

the following equations will be used to do the calculations for the ratio between width &

height.

𝑤ℎ

= 8𝑒𝐴

𝑒2𝐴 − 2 𝑓𝑜𝑟

𝑤ℎ

≤ 2 (2.6)

where

𝐴 = 𝑍0

60 �

ϵr + 12 �

0.5

+ �ϵr − 1ϵr + 1� �0.23 +

0.11ϵr

�

𝑤ℎ

= 2𝜋

�(𝐵 − 1) − ln(2𝐵 − 1) + 𝜖𝑟−12𝜖𝑟

�ln(𝐵 − 1) + 0.39 − 0.61𝜖𝑟

�� 𝑓𝑜𝑟 𝑤ℎ

≥ 2 (2.7)

where

𝐵 = 60𝜋2

𝑍0√𝜖𝑟

2.4.1 COUPLED MICROSTRIP LINES

Coupled microstriplines are in layman’s words, two microstrip lines placed close to

each other in parallel. Coupled microstriplines are used to build Filters and Directional

couplers. This coupled line structure has the ability to transmit two quasi-TEM wave

modes, usually even & odd modes.

Figure 2.4 - Coupled Microstriplines

9

Even mode exists when there are charges of the same sign in both the microstriplines

while Odd mode exits when the signs of the charges are opposite. Each of these modes

will contain unique transmission line characteristics, namely the characteristic

impedance (Z0e, Z0o) & the propagation velocity (Vpe,Vpo). Characteristic impedance in

the even & odd modes depend on the dielectric constant (ϵr), microstripline width to

substrate height ratio (𝑤ℎ

) & the separation between the microstriplines to substrate

height ratio (𝑠ℎ).

2.5 POWER DIVIDERS

Power Dividers are passive microwave component that can be used for either power

division or power combining. A power divider may divide an input signal in to two or

more parts while the power combiner may combine two or more signals. Usually a three

port network will be used to create a T junction & other types of power dividers. Power

dividers can have more than three ports, directional couplers & hybrid use up four ports

for instance. In this thesis, we focus on a three port network. Power dividers usually

divide power equally (3db) however; unequal division can also be achieved.

Figure 2.5 - Schematic Block Diagram of a n+1 port combiner and divider

10

2.5.1 THREE PORT NETWORK

A three port network can be attributed to a T junction, which is the simplest type of a

power divider. The scattering matrix of a 3 port network consists of nine parameters.

[S] = �S11 S12 S13S21 S22 S23S31 S32 S33

�

The T junction power divider consists of 3 ports, one port is used for the input signal

while the other two allows the transmission of divided signals.

In order to have minimum reflection at each of the three ports, it’s important for the

network to be matched. A matched network is where Sij = 0 when i = j ,thus the

diagonal components from the upper left corner to the bottom right corner will equal to

zero as shown below. In this case, S11=S22=S33=0.

[S] = �0 S12 S13

S21 0 S23S31 S32 0

�

Another property of the 3 port network is the property of reciprocity. If the material that

is used to make the divider is passive & has the same electrical properties throughout

(isotropic), then the transmission between any two ports suffer the same power loss as

each other despite different propagation directions. This makes the parameters of the S

matrix symmetric for Sij=Sji.

An ideal passive network is considered to be of lossless nature, without heat or radiation

generated when propagating along. Under these circumstances, power incident on one

port should be the same as the sum of powers leaving the other ports. When a matched

network is completely lossless, it has to fulfil certain conditions:

S13∗ S23 = 0 (2.8)

S23

∗ S12 = 0 (2.9) S12

∗ S13 = 0 (2.8) |S12|2 + |S13|2 = 1 (2.10)

|S12|2 + |S23|2 = 1 (2.11)

11

|S13|2 + |S23|2 = 1 (2.12)

In an ideal scenario the sum of squares of all S parameters ought to equal one, however

in reality it can never be done. What can be done is that to bring the sum of the Squares

of the parameters to be close as possible to 1. It is not possible to design an ideal

lossless network. When looking at the above equations, it’s possible to observe that in

order to satisfy the conditions laid by the equations, at least two of S12, S13, S23 must

equal zero. But this disagrees with the conditions laid down based on Reciprocity

condition. Thus it is not possible to achieve all these properties at the same time. In

order to realize a practical design, we need to relax at least one of them(Pozar 2005).

2.5.1.1 T JUNCTION POWER DIVIDER T Junction power divider is the simplest of 3 port network power dividers that can be

used in any form of transmission medium. The T junction power divider, in the absence

of transmission line loss, cannot be matched at the same time in all ports. The variants

of T junction power dividers are discussed below.

2.5.1.2 THE LOSSLESS DIVIDER

Lossless divider can be modelled with a slight variation of T junction which has come

to known by the popular term of “Y junction”. The fields and higher order modes

associated at such a junction leads up to some energy being stored at the Junction. This

Figure 2.6 -Transmission Line model of lossless divider

12

can be attributed to a lumped susceptance of value B. For the divider to be matched to

the characteristic impedance of Z0, we can write the following equation.

Yin = jB + 1Z1

+1Z2

(2.13)

Assuming the transmission line to be of low loss or lossless nature, we can consider

there is no lumped susceptance, thus

Yin =1Z1

+1Z2

(2.14)

Although in reality, the susceptance will not equal zero, it’s effect can be minimized by

adding a reactive element to the power divider. Z1 and Z2 can be selected as to have

different power ratios, if the output lines are matched, it can be match with the input

line. However, there will be no isolation between the ports.

2.5.1.3 THE RESISTIVE POWER DIVIDER Resistive power divider is a lossy, yet can be matched at all ports. This is achieved by

using lumped element resistors in the transmission line. This explanation is for an equal

power divider, but it must be mentioned that different power ratios can also be achieved.

Figure 2.7 - A 3db Resistive power divider

13

Considering the impedance of an output port, it can be seen that the total impedance of

an output port is:

Z = Z0

3+ Z0 (2.15)

Thus the input impedance of the divider will equal,

Zin = Z0

3+

2Z0

3= Z0

This shows that the inputs are matched. While the outputs are also matched, the network

is also symmetric from all ports. Therefore S11=S22=S33=0. Assuming voltage at the

input port to be V1, we calculate the voltage V at the centre of the junction,

V = V1

2Z03

Z03 + 2Z0

3=

23

V1

The output voltages can be calculated as,

V2 = V3 = V Z0

Z03 + Z0

= 34

V = 12

V1

Therefore, S21=S31=S23=0.5, which means that the output power is -6db below the level

of the input power. The scattering matrix for this can be written as,

[S] = 12 �

0 1 11 0 11 1 0

�

The power delivered to the output of the divider can be found out to be ,P2=P3=0.25Pin

by using the basic formula for power. Thus it can be seen that half of the power

delivered is dissipated through the resistor.

2.5.1.4 THE WILKINSON POWER DIVIDER In the previous types of power dividers we discussed, we saw that each of them had

significant short comings. For instance, the lossless power divider, though lossless,

suffered from the inability of being matched at all ports not did it have the ability to

isolate output ports. The resistive divider, though can be matched at all ports, suffered

from the inability to be lossless and to achieve high isolation between the output ports

14

The Wilkinson power divider is the best option to replace the above mentioned. The

Wilkinson Power Divider was first proposed by Ernest J. Wilkinson in 1960(Wilkinson

1960). Like the resistive divider, Wilkinson power divider is matched and reciprocal,

therefore Lossy. However, when matched properly, the power can be divided without

any loss with only the reflected power being dissipated. Wilkinson power divider can be

made with different power ratios, but in analysing the Wilkinson power divider, it’s

easier to consider the equal power division or the 3db case.

The isolation resistor employed between the two output ports is one of the most

important parts of the Wilkinson power divider concept. It is the single component that

enables this model of power divider achieve what its predecessors could not.

Dissipation of energy, when signal enters through any of the output ports, happens only

through the isolation resistor. This does not affect the efficiency of the Wilkinson

network. Furthermore, the isolation resistors allows the divider achieve perfect isolation

Figure 2.8 – Wilkinson Divider in microstrip format and the transmission line

format

Figure 2.9 - Wilkinson Power Divider Schematic

15

when operating at it’s working frequency. Two methods are usually used to analyse the

Wilkinson power divider, one is the original analysis proposed by Ernest J. Wilkinson,

while the other is the even and odd mode analysis(Pozar 2005). The latter will be used

to analyse in this thesis.

2.5.1.4.1 Even and Odd mode Analysis

In order to proceed with the even and odd mode analysis, a plain circuit is derived from

the original one. Thus analysis becomes much less complicated. The derived circuit is

given below.

For simplicity, we normalize impedances to the characteristic impedance Z0. This

network is symmetric across the centre plane. The two source resistors have been given

normalized values of two so that they will form a resistor of value one when in parallel

to represent the impedance of the matched ports. The quarter wave transformer lines

have a normalized characteristic impedance of Z, while the shunt resistors contain a

value of ‘r’. For an equal split power divider, Z=20.5 & r = 2. In the even mode, Vg2

=Vg3=2V0 while in the odd mode, Vg2 =-Vg3=2V0. The superposition of these two modes

we can obtain an excitation of Vg2 = 4 V0, Vg3=0. This will be useful in finding the S

parameters.

Figure 2.10 - Wilkinson Divider in Normalized and symmetric form

16

2.5.1.4.2 Even Mode

Since Vg2 =Vg3=2V0, V2e= V3

e there will be no current flowing through the ‘r’ shunt

resistor. So there will be a short circuit across the path of the shunt resistor.

The network can be bisected with having open circuits at as shown above because no

current will flow through the transmission line between the two inputs at port 1 or the

shunt resistors r/2.

Zine =

Z2

2

V2e = V0 since Zin

e = 1

Let x=0 at port1 & x=-λ4 at port 2,

Therefore, the transmission line voltage;

V(x) = V+�e−jβx + Γejβx�; where Γ is the reflection coefficient

Equalling to the x values above,

V2e = V �−

λ4

� = jV+(1 − Γ) = V0

V1e = V(0 ) = V+(1 + Γ) = jV0

(Γ + 1)(Γ − 1)

Г is seen from port 1, looking towards the resistor of normalized 2

Γ =2 − √22 + √2

V1e = −jV0√2

Figure 2.11- Even mode Excitation

17

2.5.1.4.3 Odd Mode

For the odd mode excitation, Vg2 =-Vg3=2V0, V2o= -V3

o . The parallel connected

transmission line is λ/4 and short circuited at port 1 one while the r/2 impedance can be

seen when looking in to port 2.

Port 2 can be matched for odd mode for a value of r=2. Then,

V2o = V0

V1

o = 0 All the power in the odd mode is delivered to r/2 with no power being delivered to the

port 1.Finding the input impedance when port 2 and 3 are matched with identical loads,

Zin =12

(√2)2 = 1

S11=0 (Zin=1 for port1 )

S22=S33=0 (matched in both modes)

S12=S21= V1e+V1

o

V2e+V2

o = − j√2

S13=S31= (symmetry at ports 2 and 3)

S23=S32=0 (short/open circuit at the midplane)

When the outputs are matched, there is no power dissipating through the resistor “r”

,therefore the power divider is lossless when perfectly matched. Furthermore since

S23=S32=0, there is good isolation between the ports 2 and 3.

Figure 2.12-Odd mode Excitation

18

2.6 FEW IMPORTANT TERMS FOR A WILKINSON POWER DIVIDER

• Insertion Loss(S21=S31)

For a power divider, insertion loss refers to the additional loss due to the

splitting of the signal. The additional losses are usually caused by reflections,

dielectric absorption, radiation effects and conductor losses. Conductor losses in

high frequency devices are mainly caused by the skin effect. Losses due to

reflections also increase with increase of frequency.

• Power Divider Isolation(S23=S32)

In an ideal power divider, the output ports can be expected to be mutually

isolated. Isolation means that a signal entering port 3 does not leak in to port 2 or

vice versa. Isolation values above 15db are usually considered to be good. The

achievable isolation usually depends on the design itself. Usually, when larger

the bandwidth and higher the frequency, the difficult it is to provide good

isolation.

• Return Loss(S11=S22=S33)

The return loss and VSWR both pretty much refer to the same thing. The return

loss is an indication on how well the network is matched. Return loss and

isolation go hand in hand such that, it is difficult to attain good isolation without

good return loss performance.

19

2.7 DUAL BAND WILKINSON POWER DIVIDER

For the purpose of building a microwave diplexer, we are required to understand the

concepts of the Power Dividers. In order for us to design a microstrip structure for

better diplexer performance, the power divider junction is an essential part of the

project.

In the previous chapters, basics in to power dividers & junctions were given. It was

rather obvious that the Wilkinson power divider would be the best choice for a design of

a microwave diplexer.

However, the Wilkinson Power Divider is designed for a single band operation. With

the advancement in the communication field, multi band structures have a very higher

demand. Thus, there has been much research done about creating a Dual Band

Wilkinson Power Divider with good performance. In this thesis, we look in to two

methods that has been researched about before, and published.

The equations derived in these researches have been used for calculation and design of

this part of the project. Simulation results achieved by using Agilent ADS and Sonnet

EM have also been attached.

2.8 BAND FREQUENCIES

The frequency bands selected for this project is GSM 900/1800 bands frequently used in

Asia.

System Band Uplink (MHz) Downlink (MHz)

P-GSM-900 900 890-915 935-960

DCS-1800 1800 1710-1785 1805 -1880 Table 2.1

The operating frequencies used in this thesis are 900MHz and 1800MHz.

20

2.9 COMPACT DUAL-BAND POWER DIVIDER USING BRANCH LINES AND RESISTORS

A Wilkinson power divider structure with the use of cascaded Branch lines and two

resistors is have been reported (Law & Cheng 2008). The design of the structure

basically contains cascaded branch line sections (ZA and ZB) and resistors (RA and RB).

The divider caters to upper (f2) and lower band (f1) frequency ratio of up to 3.

𝑍𝑒𝑣𝑒𝑛,1(f = f1, f2) = 2Zs (3.19)

𝑍𝑜𝑑𝑑,1(f = f1, f2) = ZL (3.20)

The conditions stated in 3.19 and 3.20 will be used to derive the required circuit

parameters where Zs refer to the source impedance and ZL refer to the load impedance.

The Design formulas obtained in the paper are as follows:

𝑛 =2𝑍𝑠

𝑍𝐿 (3.21)

𝜀 = (𝑓2

𝑓1− 1) �

𝑓2

𝑓1+ 1�� .

𝜋2

(3.21)

k =(𝑛 − 1) tan 𝜀2 + �(𝑛 − 1)2 tan 𝜀4 + √4𝑛

2𝑛 (3.22)

Figure 2.13 - Power Divider Topology

21

𝑅2 =2𝑍0

1 − � 1 − 𝑘(𝑛 − 1)𝑘

(3.23)

𝑅1 =2𝑍0

1 + 𝑘. �

(𝑛 − 1)𝑘(1 − 𝑘) (3.24)

𝑍𝐵 = �(𝑛𝑘). 𝑍0 (3.25)

𝑍𝐴 = �𝑛𝑘

𝑍0 (3.26)

Z0 refers to the characteristic impedance of the Microstripline.

22

3 CHAPTER 3: DESIGN PROCESS AND SIMULATION

3.1 RT DUROID 6010

The Substrate RT Duroid 6010 is used in the power divider design. The RT Duroid has

a relative dielectric constant of 10.2 while the substrate thickness is given as 1.27mm.

The characteristic impedance is taken to be 50Ω.

3.2 DESIGNING COMPACT DUAL-BAND POWER DIVIDER USING BRANCH LINES AND RESISTORS

The second design is reported in the following using Cascaded sections of branch lines

and resistors(Law & Cheng 2008). The design equations are referred to 3.21 until 3.26.

The source impedance and load impedance will be equal to the characteristic impedance

of the microstripline.

𝑍𝑠 = 𝑍𝐿 = 50𝛺

𝑛 =2𝑍𝑠

𝑍𝐿= 2

𝜀 = (𝑓2

𝑓1− 1) �

𝑓2

𝑓1+ 1�� .

𝜋2

= (2 − 1) (2 + 1)⁄ . 𝜋2

=𝜋6

k =(2 − 1) tan 𝜋

62

+ �(2 − 1)2 tan 𝜋6

4+ √8

4= 0.7953

𝑅2 =2𝑍0

1 − � 1 − 𝑘(𝑛 − 1)𝑘

= 202.96𝛺

𝑅1 =2𝑍0

1 + 𝑘. �

(𝑛 − 1)𝑘(1 − 𝑘) = 107.79𝛺

𝑍𝐴 = 79.29𝛺

𝑍𝐵 = 63.06𝛺

The centre frequency of the frequencies f1(900MHz) and f2(1800MHz) is calculated as:

23

𝑓𝑜 =(𝑓1 + 𝑓2)

2= 1350𝑀𝐻𝑧 = 1.35𝐺𝐻𝑧

The width and length for the designed parameters are then calculated based on the

formulas 2.3-2.7 listed under section 2.4.

𝐴 = 79.29

60 �

10.2 + 12 �

0.5

+ �10.2 − 110.2 + 1� �0.23 +

0.1110.2

� = 3.368

𝑤ℎ

= 8𝑒3.368

𝑒2∗3.368 − 2= 0.2763 < 2

therefore, the width required for ZA transmission line is calculated as:

WZA = 0.351mm

The effective dielectric constant is calculated as:

𝜖𝑒𝑓𝑓 = 1 + 10.2

2+

10.2 − 12

1

�1 + 120.2763

= 6.386

The length can be calculated using,

𝜃 =2𝜋𝑓

𝑐 �𝜖𝑒𝑓𝑓 𝑙

𝜋2

=2𝜋 ∗ 1.35 ∗ 109𝐻𝑧

3 ∗ 108𝑚𝑠−1 √6.386 𝑙

𝑙 = 21.984𝑚𝑚

Through the same method, the design parameters for ZB transmission line are found.

WZB = 0.6996mm

𝑙 = 21.624𝑚𝑚

24

3.2.1 SIMULATION AND RESULTS

As it was stated in the previous section, the first step for the simulation is to design the

proper schematic. Once all parameters are entered, one can proceed with the simulation,

however the desired results cannot be achieved at once. In order to achieve the desired

results, tuning is required until the proper results are obtained. The tuning was done

wisely with identification of different portions of the circuit rather than trial and error

The schematic for this method is given below

Figure 3.1 - Schematic Diagram

25

Figure 3.2 - Simulation Results

Thus the results have been grouped in a table below.

Loss in dB 900MHz 1800MHz

S22=S33(Return

Loss/Transmission)

-47 -45.8

S23=S32 (Isolation) -25 -30.9

S21=S13 (Insertion Loss) -3.02 -3.02

S11 (Input matching) -24.8 -31.1 Table 3.1 Results obtained

These results give much clearer dual band characteristics around 0.9GHz and 1.8GHz.

Although the isolation and input matching occurs at around 0.8GHz instead of 0.9GHz,

the values are not that far off.

4.3.2- Simulation Results

26

3.2.2 AGILENT ADS MOMENTUM

Momentum is a part of Agilent Advanced Design Systems that provides simulation

tools for design and evaluation of modern communication systems. Momentum is an

electromagnetic simulator which computes S-parameters for generalplanar circuits, such

as microstrip, slotline, stripline, coplanar waveguide, and other topologies.

Momentum is based on a numerical discretization technique called the method of

moments. This technique solves Maxwell's electromagnetic equations for planar

structures embedded in a multilayered dielectric substrate. The simulation modes

available in Momentum (microwave and RF) are both based on this technique, but use

different technologies to achieve their results.

Momentum Microwave (Full Wave Mode) and Momentum RF differ from each other

by the Green Function formulations that are being used. Microwave mode uses full

wave Green functions, whereas RF mode simplifies the Maxwell’s equations when

characterizing the substrate. This will result in calculating L and C values that are real

(instead of complex values that are frequency dependent as in Microwave mode) and

frequency independent. Therefore, Momentum Microwave provides the most accurate

results although Momentum RF simulation can help to reduce simulation time

significantly.

3.2.2.1 WORKING ON ADS LAYOUT The designed power divider is translated in to the ADS Layout with certain

modifications to accommodate the resistors.

Figure 3.3- Modified Layout of the power divider

27

3.2.2.2 ADS COMPONENT LIBRARY High Frequency model resistors, ERJ3GEYJ 101 for 100 Ohm and ERJ3GEYJ 181 for

the 180 Ohm resistors have been used for simulations in the formation of the Diplexer

here onwards.

3.2.3 FILTERS

Conventional 3rd order Open Loop filters with grooves and stubs have been used to

maximize the filters designed for the Diplexer.

3.2.3.1 LOW PASS FILTER The low pass filter has a dimension of 42.4mm by 25.6mm.

Figure3.4- Low Pass Filter & Performance

28

Modified Open-Loop Lowpass Filter

Filter order 3

Passband 0.86GHz – 1.00GHz

Centre Frequency 0.9GHz

Fractional Bandwidth 15.6%

Insertion Loss (S21) 0.911dB

Return Loss (S11) 13.65dB Table 3.1

The harmonics have been suppressed below -20db.

3.2.3.2 HIGH PASS FILTER The High Pass filter has a dimension of 42.4mm by 11mm.

Figure 3.5- Highpass filter & Performance

29

Modified Open-Loop Highpass Filter

Filter order 3

Passband 1.749GHz – 2.019GHz

Centre Frequency 1.8GHz

Fractional Bandwidth 15%

Insertion Loss (S21) 0.001dB

Return Loss (S11) 8.888dB

Table 3.2

3.2.4 THE DIPLEXER FORMATION

Although a Y or T junction is commonly used for the formation of a Diplexer, in this

project we have tried a Dual Band Wilkinson Power divider for a junction. The Filters

designed above has to be connected to the Wilkinson power divider, so that best results

can be obtained by simulation. Several connection modes have been used in deciding

the optimum diplexer performance in this thesis.

Figure 3.6- Filters connected in 90 degrees

30

Figure 3.7- Filters Connected in 60 degrees

Figure 3.8- Filters Connected in 45 degrees

The figures 3.7 and 3.8 go on to show that none of the connections have the

performance required from the Diplexer. However, while simulating for differently

connected diplexers, it was found out that better performance may be obtained by

increasing the length the transmission line connecting the Low Pass Filter to the Dual

Band Wilkinson power divider.

31

Figure 3.9- Low Pass filter connected in 90 degrees with an extended transmission line

The transmission line which was originally the size of 8mm, has been extended by

another 8mm’s, immediately there is a significant improvement on the performance of

the Diplexer. With this design in mind, the filters have been connected in 60 degrees

and 45 degrees, however the best results were produced when connected in 90 degrees.

Thus the diplexer above given was chosen for further optimization including changing

the length of the transmission line connecting the Low Pass Filter from the Dual Band

Wilkinson Power divider.

3.2.4.1 FURTHER OPTIMIZATION

Figure 3.10- Connected by the 14mm Transmission Line

32

By changing the connected Transmission line length, it was found at that better results

can be obtained at a Distance of 14mm’s.

Despite the being reasonably closer to the diplexer performance required, there is a

surge in the 2nd Harmonics. Since the Wilkinson Power Divider itself is a separate

microwave device with harmonics, it will add up to the existing harmonics thus

hindering the better performance. In order to investigate how the performance can be

made better, several changes were made particularly to the low pass filter such as

changing the port positions, adding up extra stubs, adding more grooves..etc.

Figure 3.11- With two extra set of grooves at 0.2mm apart from each other

33

3.2.5 OPTIMIZED RESULTS

Figure 3.12- Optimized Design Layout

Figure 3.13-Optimized Design Simulation Results

The final design produces an Insertion loss of -0.02db and -0.295db at Low Pass and

High Pass frequencies respectively while the return loss is at -8.712 and -5.456 at Low

Pass and High Pass frequencies.

34

3.2.6 METCLAD

Metclad is another type of substrate with a dielectric constant of 10.8 and a thickness of

1.55mm. Metclad has been used to fabricate the final design in this theses instead of RT

Duroid 6010 due to financial constraints.

3.2.7 SIMULATION FOR METCLAD

Following is the simulation results for the Diplexer with the use of Metclad as the

substrate. In this design, the width of the port 1 entrance transmission line has been

doubled from 0.35mm to 0.7mm due to practical requirenment of port connection. This

will have a slight effect on the results.

3.2.7.1 DILPEXER SIMULATION

Figure 3.14- Metclad Simulation of Diplexer

35

3.2.7.2 SIMULATION OF THE LOW PASS FILTER

Figure 3.15- Metclad Simulation for the Low Pass Filter

36

4 CHAPTER 4: FABRICATION AND MEASUREMENT

4.1 COMPONENTS USED

4.1.1 VISHAY 180R RESISTOR (0402)

Vishay SMD wraparound thin film resistor with tolerance of 2%, up to 50GHz

Figure 4.1- Impedance vs Frequency curve for Vishay Thin Film Resistor

4.1.2 PANASONIC ERJ3GEYJ101V (0603)

Panasonic SMD wraparound thick film resistor with 5% tolerance, can withstand up to

20GHz.

Figure 4.2- Impedance vs Frequency curve for Panasonic Thick Film Resistor

37

4.2 TEST FABRICATION ON FR4

The testing for the Diplexer design was carried out by way of toner transfer method on

to a FR4 substrate.

Figure 4.3 -Diplexer Fabricated on FR4

The fabricated line widths of the power divider were quite close to the actual design

value of 0.35mm.

Figure 4.4 - The Fabricated lines of 0.35mm of the power divider

38

4.3 TONER TRANSFERRING

Toner transferring was done by printing the design on a Pulsar toner paper via a Laser

Printer available from the library. First and Formost, the quality of the print was not

extremely good. To obtain better results, it would be helpful to have access to a better

Laser Printer. Using the laminator to transfer toner on to the Metclad was proven

trickier than transferring on to FR4. The methods of Laminating, Ironing and a

Combination of that were tried out, neither provided a satisfactory transfer. Shown in

figure 4.6 is how the Power divider looked after a toner transferring.

Figure 4.5 – Divider achieved by a toner transfer

It is important to note that the toner transfer selected to proceed with, was not the best

toner transfer that was made comparatively but was used due to the time constraints.

Figure 4.6- Another Toner Transfer attempt

39

4.4 FABRICATION ON METCLAD

Toner transferring using the Laminator on to the Metclad was very difficult in

comparison with the toner transfer on to the FR4. Metclad requires to cool down once

the toner transfer is made before being put in to cold water. This will allow the unwoven

fibreglass structure to absorb the toner on to the copper layer.

Figure 4.7 - Fabricated Diplexer on Metclad

When taking a closer look, it was obvious how the power divide has not been properly

transferred on to the board with lines being awry.

Figure 4.8 - Awry lines achieved by toner transfer on Metclad

40

4.5 MEASURED RESULTS FOR THE LOW PASS FILTER

Figure 4.9 -Measured Results for the Low Pass Filter

Measured results show a return loss (S11) of -11dB while the insertion loss (S21) is

around -2.4dB. Although the Return Loss value is similar to the simulated value, the

insertion loss is quite high. 2nd Harmonics have been suppressed below 20dB (-21dB).

The insertion loss can also considered low once cable loss is deducted. The centre

frequency is shifted to around 0.95GHz instead of the 0.9GHz mark.

Figure 4.10- Taking Measurements

41

4.6 MEASURED RESULTS FOR THE DIPLEXER

Figure 4.11-Results for the LowPass Filter

The measured Lowpass filter clearly shows that there is something wrong with the

fabricated Diplexer. There is a high amount of losses occurring as seen by the S21

Insertion loss being at -4.3443dB. The return loss values are in higher scale than that of

the Insertion loss which is unacceptable.

Figure 4.12-Results for the high pass filter

42

The measured result for the high pass filter is quite far off from the simulated results as

well. As in the previous case, the Insertion Loss -8.0134 dB which is far away from the

desired 0.5 dB.

Heavy losses achieved in the diplexer can be attributed to the inaccurate fabrication of

the power divider of the Diplexer.

Figure 4.13- The close up of the fabricated Diplexer

As portrayed in figure 4.13, there are several instances where the power divider has not

maintained the width of 0.35 uniformly. This is the main cause of the inaccurate results

obtained by fabrication.

43

4.7 CONCLUSION

This thesis looked in to microstrip structure design for a better performing diplexer,

simulations with good performance were achieved by using a dual band Wilkinson

power divider instead of a common port junction design. The fabricated results were

quite far off from the simulated results due to complications in Fabrication. Inability to

maintain the width of the Power Divider resulted in high losses, thus Insertion loss

being further away from 0.5 dB. In a broader sense performance could have been

bettered if not for the fabrication limitations that prevented the designs to have a

minimum line width of 0.3mm.

4.8 RECOMMENDATIONS & FUTURE WORK

It is important to research more about different Fabrication methods which will help

achieve much more accurate results on different substrates in future. Better quality

printing and laminating can help achieve a good toner transfer for better fabrication.

Furthermore, the use of dual band Wilkinson power divider in a Diplexer can be

researched by looking in to different designs of the Wilkinson Power Divider. This

project has taken a step forward by using a Wilkinson power divider instead of a

common port junction to form a diplexer, with further research and better fabrication

options, better results can be achieved.

It is also recommended to obtain professional software such as Microwave Office or

Agilent ADS, which will provide accurate simulations and enable students working on

RF/Microwave related projects, achieve better results with higher productivity.

44

References

Chatim, RH 2005, MODIFIED WILKINSON POWER COMBINER FOR APPLICATIONS IN THE MILLIMETER-WAVE RANGE, Master Thesis thesis, DEPARTMENT OF RF-TECHNIQUES / COMMUNICATION SYSTEMS, UNIVERSITY OF KASSEL,GERMANY, pp. 74. Harty, DD 2010, NOVEL DESIGN OF A WIDEBAND RIBCAGE-DIPOLE ARRAY AND ITS FEEDING NETWORK, thesis, WORCESTER POLYTECHNIC INSTITUTE, pp. 97. Jianpeng, W, Jia, N, Yong-Xin, G & Dagang, F 2009, 'Miniaturized Microstrip Wilkinson Power Divider With Harmonic Suppression', Microwave and Wireless Components Letters, IEEE, vol. 19, no. 7, pp. 440-442, Law, C & Cheng, K 2008, 'Compact dual-band power divider design using branch-lines and resistors Only,' Microwave Conference, 2008. APMC 2008. Asia-Pacific, 1-4. Lei, W, Zengguang, S, Yilmaz, H & Berroth, M 2006, 'A dual-frequency wilkinson power divider', Microwave Theory and Techniques, IEEE Transactions on, vol. 54, no. 1, pp. 278-284, Mohra, ASS 2008, 'Compact dual band Wilkinson power divider,' Radio Science Conference, 2008. NRSC 2008. National, 1-7. Pozar, DM (ed.) 2005, Microwave Engineering, John Wiley & Sons,Inc. Wilkinson, EJ 1960, 'An N-Way Hybrid Power Divider', Microwave Theory and Techniques, IRE Transactions on, vol. 8, no. 1, pp. 116-118, Zhang, Y, Tang, X, Fan, Y, Ooi Ban, L, Leong Mook, S & Mouthaan, K 2008, 'A Miniaturized Wideband Wilkinson Power Divider,' Electronics Packaging Technology Conference, 2008. EPTC 2008. 10th, 271-274. (Chatim 2005; Harty 2010; Jianpeng et al. 2009; Lei et al. 2006; Mohra 2008; Zhang et

al. 2008)

45

Appendices

Appendix A Agilent Advanced Design System Agilent ADS is a software that provides an integrated design environment to RF

electronic designers. Agilent ADS supports designing of a schematic, layout, time

domain and frequency domain simulation and electromagnetic field simulation. Agilent

ADS is one of the most popular EM simulator softwares, and one that is being used in

the industry regularly.

• Getting started with Agilent ADS is very easy, the user is required to create a

new project in the user directory provided upon installation.

• Once the new project is created, the user may be able to select from a drop down

menu on the top left hand corner to select different mediums, components,

functions. etc.

Figure 1

46

• Once the user has selected what is required, he place components on the

schematic. Furthermore, once a circuit has been built, if you are required to

produce the layout, there is a simple function of generating the layout, which

will automatically generate the layout. This may be done vice versa as well,

where you create the layout 1st and generate the layout later.

Figure 2

Figure 3

47

Appendix B Sonnet Lite EM Simulator The sonnet lite is a free limited feature version of Sonnet’s more professional suites.

Sonnet provide full wave simulations for planar 3D circuits. It also allows for design

and analysis in coupled transmission lines, microstrip matching networks, for planar

couplers and splitter analysis etc. This software also can be integrated with Agilent

ADS through a small bridging software, thus allowing to import and export layouts and

analyse.

• To get started, go to Edit Project, select New Geometry. A plain page with a

square box with dots will appear.

Figure 4

• Then, go to Circuit and select Units. Here the units used for the length and

frequency are defined, values for Dielectric Layers and Box Size can be

assigned as these are listed under Circuit as well.

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Figure 5

Figure 6

Figure 7

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• During the design, the ports must be touching the box wall as the box wall act as

ground for the port, and thus the distance between the resonator and the box wall

should be two or three times the substrate height. This is to avoid the

interference of the simulation box with the structure.

Appendix C

Fabrication Process Basic steps in fabrication

1. Import the SONNET file in Corel Draw. Make sure in Corel Draw that the page

size is set as “Letter” if your toner transfer paper is letter size. Initially only the

outline of the image will be imported we have to fill it with black colour and

remove the outline.

2. Print the layout onto the toner transfer paper. Always print directly from Corel

Draw. Never save file in .pdf format and print, because the dimensions gets

disturbed.

3. From the toner transfer paper, the toner is transferred to the dielectric board

using the laminating machine.

4. Go through the laminating machine for several times

5. After laminating, cool the dielectric board and then soak it in water for 5

minutes. Peel the toner transfer paper gently.

6. Let the surface dry naturally. Do not dry with tissues or clothes. This will cause

the toner to wear off and also some particles from the tissue might appear on the

board.

7. After the board is dried, run it under the water to remove the chemical layer left

from the toner paper.

8. Put a layer of green film onto the layout side and laminate again, at least 20

times. Make sure there are no wrinkles. This will create an additional green layer

on the toner which will protect the toner from etching chemical. Peel the green

film after laminating.

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9. The board is then etched inside an etching machine with ferric chloride at a

temperature of about Co150 .

10. It’s very important to check the board every few minutes to avoid over etching.

After the area besides the coated tracks is totally etched, wash it with water and

dry it.

11. Remove the coating with acetone.

12. Solder the ports. The best option is to use solder paste, it will make soldering

much easier.