S/CLC Compensation Topology Analysis and Circular Coil ...

496 IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION, VOL. 3, NO. 2, JUNE 2017

S/CLC Compensation Topology Analysis andCircular Coil Design for Wireless

Power TransferYijie Wang, Senior Member, IEEE, Yousu Yao, Xiaosheng Liu, Member, IEEE, and Dianguo Xu, Fellow, IEEE

Abstract— Wireless power transfer (WPT) has drawn a lot ofattention due to its inherent advantages and great potential invarious applications. This paper proposes a novel compensationtopology composed of one compensation inductor and threecompensation capacitors. The newly proposed compensationtopology, named as S/CLC, provides the advantages of constantvoltage output and easy achievement of zero phase angle andzero voltage switching. It is also free from the constraintsimposed by the loosely coupled transformer (LCT) parameters.All the above-mentioned merits are theoretically analyzed andverified by simulation and experiment. Compared to double-sided LCC compensation topology, S/CLC compensation topologyneeds less compensation components, meaning lower cost, smallersize, higher power density, and greater potential in practicalapplications. The optimization of a circular pad is conductedas well since the properties of the LCT have a significantimpact on the performance of a WPT system. A circular padis manufactured in terms of the conclusions obtained fromthe optimization. The coupling coefficient Circular pad design,constant voltage output (CVO), wireless power transfer (WPT),zero phase angle (ZPA), zero voltage switching (ZVS).of thecircular pad is quite appealing, demonstrating the validity ofthe optimization.

Index Terms— Circular pad design, constant voltageoutput (CVO), wireless power transfer (WPT), zero phaseangle (ZPA), zero voltage switching (ZVS).

I. INTRODUCTION

IN RECENT 30 years, wireless power transfer (WPT) hasbeen widely studied due to its advantages of low main-

tenance, high reliability, flexibility, safety, and environment-friendly features [1]. Up to date, it has been applied in variousscenarios, such as cell phone, tooth brush, medical implants,and noncontact charging for electrical vehicles [2], and it’scontinually finding new applications.

There are many research fields in WPT, such as com-pensation topology and circuit analyses, coil design tech-niques for larger coupling coefficient, optimization for highefficiency [3], control methods, peripheral object detection,and safety issue [4]. Among these fields, compensation topol-ogy and circuit analyses are of fundamental role since itsdetermination of resonant frequency, power factor, outputcharacteristics. Coil design techniques for larger coupling

Manuscript received December 1, 2016; accepted January 6, 2017. Date ofpublication January 10, 2017; date of current version June 14, 2017. This workwas supported by the Fundamental Research Funds for the Central Universitiesunder Grant HIT.BRETIII.201510.

The authors are with the School of Electrical Engineering andAutomation, Harbin Institute of Technology, Harbin 150001, China(e-mail: [email protected]; [email protected]; [email protected];[email protected]).

Digital Object Identifier 10.1109/TTE.2017.2651067

coefficient are also very important since a larger coupling coef-ficient means higher power level and system efficiency, lowerpower loss.

In accordance with the literature published, four basiccompensation topologies, SS, SP, PS, and PP, where thefirst S or P represents series or parallel compensation ofthe primary coil and the second S or P stands for seriesor parallel compensation for secondary coil, are mostlyresearched [5]–[11]. Among the four basic compensationtopologies, SS and SP are more applicable for WPT systemsince the input of the compensation topology is usually squarewave voltage. The voltage of the primary parallel compensa-tion capacitor rises to source voltage immediately when thevoltage polarity changes, causing instant large current andimpairing capacitor lifespan.

The resonant capacitor for SS and SP compensation topolo-gies can be designed in accordance to three ways. The firstway is designing the capacitor to resonate with the coilleakage inductance. The second way is setting the capacitorto resonate with the coil self-inductance. The last way ischoosing the secondary compensation capacitor to resonatewith the secondary coil while selecting the primary capacitorto eliminate the reactive part of the circuit seen by the source.For SS compensation topology, the second way is equivalent tothe third way. SS compensation topology provides a constantvoltage output (CVO) characteristic if the capacitor is designedby the first way, but it offers a constant current output (CCO)characteristic if the second capacitor design way is adopted.CVO and CCO operations in this paper indicate constantvoltage and current, both at the end load. SP compensationtopology shows a CCO characteristic if the capacitor isdesigned by the first way. If the capacitor is designed bythe second or third way, the topology possesses neither CVOnor CCO characteristic. Only SS compensation topology withfirst capacitor design method, named as SS compensationtopology for short hereafter, will be compared with the newlyproposed compensation topology since both of them show aCVO characteristic.

SS compensation topology has two drawbacks. First, it isconstrained by the parameters of the loosely coupled trans-former (LCT). If the original dc input voltage and systemoperation frequency are given, the maximum output powerof SS compensated WPT system cannot be altered with-out replacing the LCT. Second, regulating system powerfactor to desirable value under the constraints of transferpower, operation frequency and LCT parameters is quitedifficult.

2332-7782 © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

WANG et al.: S/CLC COMPENSATION TOPOLOGY ANALYSIS AND CIRCULAR COIL DESIGN FOR WPT 497

Except four basic compensation topologies, several novelcompensation topologies with some appealing characteristicshave been proposed in recent years, such as SP/S [12],S/SP [13]–[15], LCL [16], [17], and LCC [18]–[23].Villa et al. [12] propose a new compensation topology namedSP/S, which can transfer rated power even with high misalign-ment. This paper focuses on misalignment tolerance becauseit is applied in dynamic charging system, where high mis-alignment is inevitable. Both load current and voltage changewith the load, indicating neither CCO nor CVO characteris-tic. This is adverse to the simplification of control circuit.Hou et al. [13] present another novel compensation topologycalled S/PS. It can be seen as the combination of SS andSP. Experimental results have validated its advantages of zerophase angle (ZPA) and insensitivity. Similar to SS compensa-tion topology, S/SP compensation topology is also constrainedby LCT parameters. Hou et al. [14], [15] develop a preciseanalysis method to analyze the characteristics of a coupling-insensitive S/SP compensated WPT system. Compared withthe analytical results in [13], the precision has been improved.Nevertheless, the aforementioned shortcoming still exists sincethe compensation topology and parameter tuning method areunchanged. Keeling et al. [16] analyze the effects of tuningdeviation in the resonant tank of an LCL inductively coupledpower transfer pickup. It can readily compensate for 20%inaccuracy in the tuning capacitor. The system performanceis appealing but additional elements are added, deterioratingsystem efficiency and raising system cost. Su et al. [17] studythe best performance of LCL/P compensation topology bymodeling and optimizing. The system performance is quitesatisfactory, but the optimization process is time-consumingand complicated.

The double-sided LCC compensation topology has beenwidely researched due to its advantages of ZPA, zero volt-age switching (ZVS), load-independent operation frequencyand output current. It is first proposed in [18], where thetuning method is also given. The tuning method to real-ize ZVS for the primary-side switches is also elaborated.Li et al. [19] integrate the additional inductors with themain coils to reduce system size and improve system powerdensity. Zhu et al. [20] analyze double-sided LCC compen-sation topology from the perspective of ideal symmetricalT-type network. The method to achieve ZVS condition isalso proposed. Li et al. [21] conduct a comparison studybetween SS and double-sided LCC compensation topologies.They compare these two compensation topologies in termsof output power displacements, impacts of load variation,voltage, and current stresses on compensation componentsand misalignment tolerance. On the whole, double-sided LCCcompensation topology is better. Zhang and Mi [22] providea comprehensive review of existing compensation topologiesfor the WPT system. Individual passive resonant networks aresummarized and employed to analyze the output characteristic,realization of ZPA and soft switching. This paper illustrates theessence of constant current or voltage output characteristic ofall existing compensation topologies, including double-sidedLCC. Double-sided LCC compensation topology possessesa lot of appealing characteristics. However, it needs two

compensation inductors and four compensation capacitors,resulting in larger size, higher cost, and lower efficiency ofthe system.

To the best of the authors’ knowledge, the WPT system hasat least four requirements¯ZPA and ZVS, CCO or CVO, freefrom LCT parameters and less compensation components—on compensation topology. According to previous analysis,none of the existing compensation topologies can satisfythe four requirements simultaneously. This paper proposes anovel compensation topology called S/CLC. It provides thecharacteristics of easy achievement of ZPA, ZVS, and CVO.It’s also free from the constraints imposed by LCT parameters.Compared with double-sided LCC compensation topology, itneeds less compensation components, indicating lower cost,smaller size, higher power density, and efficiency.

Apart from compensation topology, coil design techniquesare also very important in designing a high-performanceWPT system. The coil design techniques can be dividedinto three categories. The first is based on magneticanalysis software packages, such as JMAG [24], [25],ANSYS Maxwell [26]–[28], and ELF/MAGIC [29]. Thesesoftware packages can easily figure out the self-inductance,mutual inductance, and magnetic field distribution of theLCT. Then the LCT with best comprehensive properties canbe chosen. The second is in accordance with Biot–Savart’slaw [27] or Neumann’s formula [5], [28], [30]. However, thecalculations of the magnetic flux intensity B or the mutualinductance M are extremely complex and time-consuming.Therefore, all the studies still calculate B or M by magneticanalysis software packages, even though the formulas tocalculate B or M are given. The third is in terms of Ohm’slaw for magnetic circuit [31]–[33]. Researchers concentrateon building precise magnetic circuit model of the LCT.Nishimura et al. [31] designed and built a flat transformer fora full-bridge resonant converter which transferred several W ofpower across a large, variable human skin gap. Although themagnetic circuit of the transcutaneous large air gap transformerwas given, the magnetic circuit was too simple, leading to lowpersuasion. The magnetic circuit of core-type transcutaneousenergy transmission system given in [32] is similar to thatin [31]. The precision of the magnetic circuit is still notconvincing though it has been improved. Wei et al. [33] giveaccurate reluctance circuit of a contactless transformer. On thebasis of the reluctance circuit, the contactless transformeris optimized. Nevertheless, the calculation of the reluctancecircuit is quite complicated, limiting its practical application.Besides, the maximum air gap considered in this paper is only14 mm, which is quite small and differs greatly from the airgap of the WPT system proposed here.

This paper is organized as follows. Section I reviewedprevious studies on compensation topology and LCT designapproaches of WPT system, followed by the analyses aboutthe appealing characteristics of newly proposed compensationtopology S/CLC in Section II. Section III focuses on theoptimization of a circular pad. A practical circular pad ismanufactured under the guide of the conclusions obtainedfrom the optimization. The analyses on S/CLC compensationtopology are verified by PSpice simulation in Section IV.


Fig. 1. S/CLC compensation topology.

The simulation results agree well with the calculated results,validating the correctness of theoretical analyses. In Section V,a WPT prototype was built, where the practical experimentswere carried out. The experimental results match well withthe theoretical results. The validity of theoretical analyses isverified again. Section VI concludes this paper.

II. S/CLC COMPENSATION TOPOLOGY ANALYSIS

The diagram of newly proposed S/CLC compensation topol-ogy is shown in Fig. 1. Uin is original dc input voltage source.Q1–Q4 are four MOSFETs, which are employed to createhigh-frequency alternating current. C1 is series compensationcapacitor of primary winding. C1 corresponds to the S ofS/CLC compensation topology as it is in series with LP, theself-inductance of primary coil. LS is the self-inductance ofsecondary coil. k stands for the coupling coefficient betweenprimary and secondary coils. C2 is parallel compensationcapacitor of secondary winding, which refers to the first Cof S/CLC compensation topology. L1 is secondary seriescompensation inductor, and it corresponds to the L of S/CLCcompensation topology. C3 is in parallel with the rectifier,which consists of four diodes D1–D4. C3 is called phaseshift capacitor in this paper because the only function ofC3 is to regulate the input impedance angle of the system.C3 represents the second C of S/CLC compensation topology.L F is filter inductor and CF is filter capacitor. RL is practicalresistive load. uAB is output voltage of the inverter while uabrepresents input voltage of the rectifier.

The circuit of the diode rectifier, filter inductor and capac-itor, and resistive load, circled by the red dashed line inFig. 1, is replaced by its equivalent resistance RE for analysissimplification. According to [34], RE can be calculated by

RE = π2

8RL . (1)

The CVO characteristic of S/CLC compensation topology,implying the load current does not change with the load, willbe first given by analyzing the first-order harmonic of thesquare voltage waveform (uAB) at the switching frequency.The original dc input voltage source and the inverter can bereplaced by a sinusoidal voltage source with switching fre-quency. The parasitic resistances on the inductors and capac-itors are neglected for simplicity of analysis. The accuracy ofthe approximations will be verified by practical experimentsin the later section. Replacing LCT with its T-type model,the analytical circuit of Fig. 1 can be derived, as shown

Fig. 2. Analytical circuit of S/CLC compensation topology.

in Fig. 2. The apostrophe symbols indicate the variables ofthe secondary side have been converted to the primary side.In light of [18], the turns ratio of the secondary to primaryside is defined as

n =√

LS

LP. (2)

UAB and U′ab are the first-order harmonics of uAB and

uab, respectively. LPL and L ′SL are the leakage inductances

of primary and secondary coils respectively while LM is themutual inductance. ULM represents the voltage over the mutualinductance. C ′

2a and C ′2b are split by C ′

2, the primary-convertedcapacitance of C2. The relationship between C ′

2a , C ′2b, and C ′

2can be expressed as follows:

C ′2a + C ′

2b = C ′2. (3)

The split is done for analysis simplicity. Both C ′2a and C ′

2bare part of C ′

2. Only one capacitor C2 is needed in practicalsystem. C1 and LPL make up a series resonant tank while L ′

SLand C ′

2a form an LC resonant tank. I′LC is the output currentof the LC resonant tank. C ′

2b and L ′1 form a CL resonant tank.

All resonant angular frequencies equal ωs, the switchingangular frequency. Then the following equation can be yielded:

ω2s = 1

LPLC1= 1

L ′SLC ′

2a= 1

L ′1C ′

2b. (4)

The variables in Fig. 2 can be derived by the followingequations: ⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

LPL = (1 − k) · LP

LM = k · LP

L ′SL = (1 − k) · LS/n2

C ′2a + C ′

2b = n2 · C2

L ′1 = L1/n2

C ′3 = n2 · C3

U ′ab = Uab/n

R′E = RE/n2.

(5)

Since C1 and LPL make up a series resonant tank, andthe resonant angular frequency equals the switching angularfrequency, the impedance of the series resonant tank is zero.Consequently, the voltage over the mutual inductance ULMequals UAB.

According to constant voltage input (CVI) to CCO charac-teristic of an LC resonant tank, I′LC can be calculated by

I′LC = −jUAB

√C ′

2a

L ′SL

. (6)


In terms of the constant current input to CVO characteristicof a CL resonant tank, U′

ab can be yielded as follows:

U′ab = −jI′

LC

√L ′

1

C ′2b

= −UAB

√C ′

2a

L ′SL

· L ′1

C ′2b

. (7)

Substituting (4) into (7), the primary-converted input voltageof the rectifier can be rewritten as follows:

U′ab = −UAB · L ′

1

L ′SL

. (8)

Then the power consumed by RE can be obtained

PRE =(

UABL ′1

L ′SL

)2

· 1

R′E

(9)

where UAB is the Root Mean Square (RMS) amplitudeof UAB. Substituting (1), (2), and (5) into (9), PRE can berewritten as follows:

PRE = 8L21U2

AB

π2 (1 − k)2 LPLS· 1

RL. (10)

Actually, the power calculated by (10) is consumed by RL.Therefore, the voltage on RL can be obtained

URL = 2√

2L1UAB

π (1 − k)· 1√

LP LS. (11)

Generally, UAB is preset and unchangeable. For a givenload, the system output power can be conveniently alteredby changing the values of compensation components in termsof (11). For four conventional and S/SP, SP/S compensationtopologies, to alter the system output power, both the values ofcompensation components and the LCT need to be replaced.However, making a new LCT with desirable parameters is timeand cost consuming. S/CLC is better than these topologiesfrom the perspective of power adjustment convenience.

According to (11), the system output voltage is irrelevantwith the load, referred to as CVO characteristic in this paper.This characteristic is called load-independent output charac-teristic in some publications. It is also very important since itwill significantly simplify the design of control circuit.

The achievement of ZPA and ZVS is analyzed as follows.Based on fundamental circuit theory, the input impedance ofS/CLC compensated WPT system can be obtained as follows:

Z in =[(

R′E//

1

jωC ′3

+ jωL ′1

)//

1

jωC ′2

+ jωL ′SL

]

// jωLM + jωLPL + 1

jωC1(12)

where the operator “//” represents the parallel calculation ofimpedance. Considering that C1, C ′

2a, and C ′2b resonate with

LPL, L ′SL, and L ′

1, respectively, all at switching frequency, (12)can be simplified as (13) as shown at the bottom of this page.

To achieve ZPA, the real part of the denominator of (13)should be zero, that is

LMC ′2 R′

E + L ′SLC ′

2b R′E − ω2

s L ′1 LMC ′

2aC ′3-ZPA R′

E = 0. (14)

C ′3 in (13) has been replaced with C ′

3−ZPA in (14) to indicateit is employed to realize ZPA. Then C ′

3−ZPA can be solved

C ′3-ZPA = LMC ′

2 + L ′SLC ′

2b

ω2s L ′

1 LMC ′2a

. (15)

What really wanted is ZVS, which can lower power loss andimprove system efficiency. To achieve ZVS, the system inputimpedance should be inductive [18]. Assuming the desirableinput impedance angle is θ , then the corresponding value ofC ′

3−ZVS can be derived

C ′3-ZVS = LMC ′

2 R′E +L ′

SLC ′2b R′

E −ωsLM L ′1C ′

2a tan(θ)

ω2s LML ′

1C ′2a R′

E. (16)

Substitute (15) into (16), C ′3−ZVS can be rewritten as

follows:

C ′3-ZVS = C ′

3-ZPA − tan (θ)

ωs R′E

. (17)

In accordance with (17), the larger θ (0° < θ < 90° ),the smaller C ′

3−ZVS. In other words, smaller C ′3−ZVS results

in larger input impedance angle, hence easier achievementof ZVS. Equation (17) also indicates a positive correlationbetween R′

E and C ′3−ZVS. It means the smaller R′

E , thesmaller C ′

3−ZVS. To ensure the achievement of ZVS under allconditions, C ′

3−ZVS should be chosen under worst situation.Hence, the formula to calculate C ′

3−ZVS can be rewritten as

C ′3-ZVS = C ′

3-ZPA − tan (θ)

ωs R′E−min

(18)

where R′E−min represents minimum primary-converted equiv-

alent resistance.The input impedance angle of the system can be easily

altered by only changing the value of C3 in terms of previousanalyses on the achievement of ZPA and ZVS. This featureis desirable since it facilitates circuit debugging. This methodto regulate the input impedance angle does no harm to theresonant tank. It is comparatively independent in a sense.

III. CIRCULAR COIL DESIGN

The performance of a WPT system mainly depends onthe compensation topology characteristic and LCT property.Section II has analyzed the CVO characteristic of S/CLCcompensation topology, the other performance-related factor,LCT property, is analyzed as follows.

Many types of coil have been proposed up to now, such ascircular pad, DD coupler, square coils, DDQ coupler, four-square coils. Experimental results show that the couplingcoefficients of circular pad and DD coupler are higher whilethe costs are lower [35], resulting in various applications ofthem.

The proposed WPT system is designed for solar arraydrive assembly (SADA), where the primary rotates with

Z in = jωs L ′SL LMC ′

2b R′E(

LMC ′2 R′

E + L ′SLC ′

2b R′E − ω2

s L ′1 LMC ′

2aC ′3 R′

E

) + jωs L ′1 LMC ′

2a

. (13)


Fig. 3. Diagram of relative rotation between primary and secondary coil.

Fig. 4. Three different core structures. (a) Ferrite bars structure with identicalstrips. (b) Ferrite bars structure with different strips. (c) Ferrite panel structure.

respect to the secondary. The relative rotation is illustrated byFig. 3. Compared to DD coupler, circular pad is more suitablefor this scenario. In this application, no misalignment existsbetween the primary and secondary coils throughout theoperation, hence no coupling coefficient variation. Therefore,it is unnecessary to study the misalignment tolerance ofthe compensation topology here. The circular pad is mainlycomposed of two parts: ferrite core and coil. The optimizationon each part will be conducted with the finite element analysissoftware ANSYS Maxwell.

A. Ferrite Core of Circular Pad

What follows is the influence of ferrite core structure on thecoupling coefficient. Three different core structures, shown inFig. 4, are studied. The simulation parameters are listed inTable I, while the simulation results are given by Table II.

According to the simulation results, all self-inductance,mutual inductance and coupling coefficient increase with thecoverage area of ferrite when the thickness is identical. More-over, the differences between the simulation results of threecore structures are minor. All simulations will be conductedwith circular ferrite panel for simplicity.

The effect of core diameter on the coupling coefficientis studied. The simulation parameters are the same as thoseof Fig. 4(c) except the diameter of ferrite panel. Simulationresults are shown in Fig. 5. The bigger the diameter of ferrite

TABLE I

SIMULATION PARAMETERS OF CORE STRUCTURES ONCOUPLING COEFFICIENT

TABLE II

SIMULATION RESULTS OF PREVIOUS THREE CORE STRUCTURES

Fig. 5. Simulation results of core diameter versus the coupling coefficient.

panel, the higher the coupling coefficient. The coupling coef-ficient shows a strong positive relationship with the diameterof ferrite panel.

The effect of core thickness on the coupling coefficient isalso studied. The simulation parameters are identical to thoseof Fig. 4(c) except the thickness of ferrite panel. Simula-tion results are exhibited in Fig. 6. The coupling coefficientincreases with the thickness of ferrite core, but the incrementis minor. The coupling coefficient is 0.57561 when the core is4mm thick. It increases to 0.58447 when the core is 10 mmthick. The latter is only 1.54% larger than the former butthe core volume and mass increase by 150%. Therefore, athinner ferrite core is better. The thickness must be largerthan the critical value where magnetic saturation happens.


Fig. 6. Simulation results of core thickness versus the coupling coefficient.

Fig. 7. Two kinds of coil structure. (a) Concentrated winding. (b) Distributedwinding.

The availability also needs to be considered, which maysignificantly affect the cost.

B. Coil of Circular Pad

There are two kinds of winding structure: concentratedwinding and distributed winding, shown in Fig. 7. Accord-ing to [33], the latter leads to higher coupling coefficient.Therefore, distributed winding is employed in proposed WPTsystem.

The diagram of proposed circular coil is shown in Fig. 8.It has four main dimension-related parameters: inner diam-eter Di, outer diameter Do, height H and turns Nt . Theeffect of turns on coupling coefficient is first studied. Thesimulation parameters are the same as those of Fig. 4(c) exceptturns of coil. Simulation results are listed in Table III. Boththe self-inductance and mutual inductance are approximatelyproportional to the square of turns, highly consistent with thetheoretical calculation formulas of self-inductance and mutualinductance. The coupling coefficient has nothing to do withturns of coil. Therefore, Nt is fixed to 25 in the followingsimulations.

What follows is the effect of H on the coupling coefficient.H equals the diameter of the Litz wire since distributedwinding is employed. Apart from height of coil, all other

Fig. 8. Diagram of circular coil.

TABLE III

SIMULATION RESULTS OF TURNS ON COUPLING COEFFICIENT

Fig. 9. Effect of H on the coupling coefficient.

simulation parameters are identical to those of Fig. 4(c). Thesimulation results are displayed in Fig. 9. The bigger theheight of coil, the smaller the coupling coefficient. It is easyto understand since bigger height leads to larger distancebetween ferrite panels. The height of coil equals the diameterof Litz wire because distributed coil is employed. To gethighest coupling coefficient, the Litz wire should be selectedwith minimum diameter under the premise of meeting therequirements of current.

The simulation results of the coupling coefficient againstinner and outer diameter are shown in Fig. 10, where Dfprefers to the diameter of ferrite panel. The other simulationparameters are the same as those of Fig. 4(c). When Di isconstant, the coupling coefficient rises with Do. When thedifference between Do and Di is constant, the bigger the Di,the higher the coupling coefficient. When the sum of Do andDi equals unity, the coupling coefficient increases with Difirst and decreases later. The highest coupling coefficient isachieved when Di/Dfp is 0.2. When the ratio increases to 0.3,the coupling coefficient drops little, but the employed copper


Fig. 10. Simulation results of the coupling coefficient against inner and outerdiameter.

decreases by 5.2%. Correspondingly, both the cost and massof the system have a considerable fall.

C. Practical Circular Pad Design

Space craft has harsh requirements on the size and massof WPT system. The dimension and mass of each circularpad must be smaller than 200 × 200 × 15 mm3 and 1500 g.First of all, the ferrite structure should be determined. In termsof the conclusions obtained before, ferrite panel structure isoptimum. However, circular ferrite panel with relatively largediameter is difficult to manufacture due to its fragility, andthe cost is quite high because the demand is comparativelysmall. In addition, few corporations produce circular ferritecore, and the dimensions are not satisfactory. Therefore, ferritebars structure with identical strips is finally chosen due to itscost and availability.

Second, appropriate Litz wire should be selected. The sizeof finally selected Litz wire is �0.1 mm × 500. It canconduct about 10 A current, significantly larger than whatis really needed, but we still choose this Litz wire becauselarge coverage and small self-inductance need to be achievedsimultaneously. In fact, the coil can be wound with multi par-allel thin Litz wires. This can not only achieve large coverageand small self-inductance at the same time, but also reducethe height of coil. However, it needs high manufacturingtechniques, difficult to realize in the laboratory.

Next, the thickness of ferrite bars is set to be 10 mmsince the diameter of selected Litz wire is 3 mm, and thethickness of the container, where circular pad is put in, isabout 2 mm. Then, a simulation like Fig. 10 is carried out.Dfp is decreased to 200 mm to meet dimension requirements.Simulation results indicate that the highest coupling coef-ficient is achieved when Di/Dfp is 0.25 and (Do–Di)/Dfpequals 0.75. The highest coupling coefficient is 0.41012.According to simulation results, the inner and outer diametersof the coil are 50 and 200 mm, respectively.

What follows is selection of the length and width offerrite bar. Roughly speaking, the coupling coefficient ispositive related to the total coverage of ferrite bar. Therefore,both the length and width should not be too big. They had

Fig. 11. Output waveform of the square-wave voltage source.

TABLE IV

SOME CRITICAL PARAMETERS ADOPTED IN THE SIMULATION

better less than 80 and 25 mm, respectively. A 70 × 18 mm2

ferrite bar is finally selected. The number of ferrite bar in eachside is set to be 10.

Last but not least, the mass requirement needs to bechecked. The densities of Litz wire and ferrite bar are8.9 and 4.8 g/cm3, respectively. According to some simplecalculations, the masses of the Litz wire and ferrite panel foreach circular pad are 343.1 and 604.8 g, respectively. Eachcontainer weighs about 200 g. The total mass of a circularpad is 1147.9 g, which is less than 1500 g and meets themass requirement. If the total mass is more than the requiredmass, the core thickness should be deceased to meet the massrequirement. If the mass requirement still cannot be satisfied,the dimensions of the coil need to be altered to reduce coilmass. The finally manufactured circular pad is shown in Fig.18, numbered by 3. The measured coupling coefficient is0.3618, which is quite close to simulated result if the dif-ference between core structures is considered, demonstratingthe validity of circular pad optimization.

IV. SIMULATION VERIFICATION

All the simulations in this section are related to electriccircuit, having nothing to do with magnetic circuit. Theadopted simulation software is PSpice from Cadence DesignSystems, Inc. The simulation circuit is similar to the circuitshown in Fig. 1. The original dc input voltage source andthe inverter are replaced by a square wave voltage sourcewhile the diode rectifier in secondary side are persisted. Theoutput waveform of the square-wave voltage source is shownin Fig. 11. Some critical parameters adopted in the simulationare listed in Table IV.

When the input voltage shown in Fig. 11 and the parameterslisted in Table IV are adopted, the waveforms of input voltage


Fig. 12. Waveforms of input voltage and current when ZVS is achieved.

Fig. 13. Waveforms of input voltage and current when ZPA is achieved.

and current can be obtained, shown in Fig. 12, indicating anachievement of ZVS. The leading angle of the fundamentalcomponents of the current and voltage is 38.7°, demonstratingthe validity of theoretical analyses about ZVS. The leadingangle obtained by simulation is 8.7° larger than that obtainedby theoretical analyses. It is mainly caused by the nonsinu-soidal input voltage, rectifier, filter inductor and capacitor.

ZPA can be realized if C3 is enlarged to 77.38 nF, calculatedby (15). The waveforms of input voltage and current areexhibited in Fig. 13. The angle between the current and thevoltage is 13.8°. The difference between the simulation andtheoretical analyses has been interpreted in last paragraph.

Through the comparison between Figs. 12 and 13, theinput impedance angle can be readily changed by alteringC3. The smaller C3, the larger input impedance angle. Thischaracteristic makes circuit debugging very simple, reducingsystem cost indirectly.

Apart from the advantage of easy debugging, the proposedcompensation topology is free from the constraints imposed bythe LCT parameters. Fig. 14 displays system output voltageagainst input voltage when different L1, C2, C3 combinationsare adopted. Based on (11), system output voltage dependson L1, UAB, k, LP, and LS. UAB is proportional to inputvoltage. C2 is changed during the simulation since C2b needsto resonate with L1. C3 is changed to keep input impedanceangle constant. Fig. 14 verifies the correctness of theoreticalanalysis on system output voltage. The system output voltage,hence the system output power, can be readily changed byonly altering the values of compensation components. We do

Fig. 14. System output voltage against input voltage when different L1, C2,and C3 are adopted.

Fig. 15. Waveform of output voltage when a sudden load change happens.

not need to manufacture a new LCT with desirable parameters.This characteristic is cost and time saving.

The CVI-CVO characteristic is also simulated by PSpice.At the beginning, the resistive load RL is 80.49 �. Then theload is decreased to 60.22 � at 20 ms. The filter capacitanceis increased to 470 μH to reduce the oscillation when the loadalters. The waveform of output voltage is shown in Fig. 15.When RL is 80.49 �, the output voltage is 70.01 V. When RLchanges to 60.22 �, the output voltage decreases to 69.81 V.The load decreases by 25.2%, but the output voltage onlydecreases by 0.29%, demonstrating an excellent CVI-CVOcharacteristic of S/CLC compensation topology. Theoreticalanalysis implies that the output voltage does not vary with theload. The output voltage in Fig. 15 varies with the load, causedby the nonsinusoidal input voltage, rectifier, and filter inductorand capacitor. The transient process continues for only 14 msand the peak-to-peak value of the oscillation is only 0.4 V,doing no harm to system performance for practical application.

The double-sided LCC compensation topology is widelyresearched and employed because this compensation topologypossesses many advantages, which have been elaborated inSection I. All these merits exist in S/CLC compensationtopology. However, S/CLC compensation topology only needsone inductor and three capacitors while double-sided LCCcompensation topology needs two inductors and four capaci-tors. S/CLC compensation topology is better than double-sidedLCC from the perspective of cost, system size, and powerdensity.

The misalignment tolerance of the proposed system isanalyzed. The blue dashed line in Fig. 16 shows the simulation


Fig. 16. Simulation results of load voltage against coupling coefficient.

Fig. 17. Simulation profiles of system efficiency versus coupling coefficient.

results of load voltage against coupling coefficient when allsimulation parameters are identical to those listed in Table IVexcept coupling coefficient. When the coupling coefficientdecreases from 0.3618 to 0.24 (33.7% decrease), the loadvoltage decreases from 70.01 to 59.46 V (15.07% decrease).Though the descent of load voltage is much lower than thatof coupling coefficient, the decrease is still so big that cannotbe neglected.

The red solid line in Fig. 16 shows the simulation resultswhen the system is tuned at a coupling coefficient of 0.32.In this scenario, C1, C2, and C3 are 52.63, 85.69, and 76.54 nF,respectively. The other employed parameters, except couplingcoefficient, are the same as those given by Table IV. Thesimulation results indicate that the variation of load voltageis quite small when the coupling coefficient decreases from0.3618 to 0.24, showing an extremely good misalignmenttolerance capability.

The misalignment tolerance capability of S/CLC compen-sation topology has been studied when WPT system is tunedat different coupling coefficients. Although the misalignmenttolerance capability when the system is tuned at a couplingcoefficient of 0.32 is much better than that when the systemis tuned at a coupling coefficient of 0.3618, the practicalsystem in this research is tuned at highest coupling coefficientsince in SADA application, misalignment will not happen.If the newly proposed compensation topology is applied indynamic charging for electric vehicle, the system should be

TABLE V

ESRS OF EMPLOYED COILS IN S/CLC COMPENSATION TOPOLOGY

Fig. 18. S/CLC compensated WPT prototype.

tuned when an appropriate misalignment happens to improvesystem misalignment tolerance.

What follows is the impact of misalignment on systemefficiency. The profiles of system efficiency versus couplingcoefficient are shown in Fig. 17, while the equivalent seriesresistances (ESRs) of coils (primary coil, secondary coil, andcompensation inductor) are taken into consideration. The bluedashed line and red solid line are obtained when the system istuned at a coupling coefficient of 0.3618 and 0.32, respectively.The ESR of coil can be divided into two parts: ESRcl andESRhl. ESRcl refers to the ESR caused by copper loss whileESRhl refers to the ESR caused by hysteresis loss. ESRcl’s ofcoils are obtained by practical measurement while ESRhl’s areobtained in terms of [36]. They are given in Table V. The ESRsof compensation capacitors are neglected since compared toESRs of coils, they are quite small.

When the system is tuned at a coupling coefficient of0.3618 (case 1), the system efficiency increases with couplingcoefficient. If the system is tuned at a coupling coefficient of0.32(case 2), the system efficiency rises first and drops laterwhen the coupling coefficient varies from 0.24 to 0.3618. Theefficiency of case 1 is higher than that of case 2 whereas thevariation of system efficiency of case 2 is smaller than that ofcase 1. Anyway, the system efficiencies corresponding to bothcases are quite high and the variations are quite small.

V. EXPERIMENTAL VERIFICATION

To further verify the theoretical analysis, an S/CLC com-pensated WPT prototype is built and shown in Fig. 18. It’smade up of six parts: dc input voltage source, primary Printed


TABLE VI

CRITICAL PARAMETERS OF THE COMPONENTS EMPLOYED INTHE PROTOTYPE

TABLE VII

DETAILED PARAMETERS OF THE LCT

Fig. 19. Output voltage and current waveforms of the inverter when theparameters listed in Table VI are adopted.

Circuit Board (PCB), LCT, secondary PCB, resistive load, andoscilloscope. They are numbered from 1 to 6 in order. Criticalparameters employed in the prototype are listed in Table VI,and the detailed parameters of the LCT are given by TableVII.

Fig. 19 shows the output voltage and current waveforms ofthe inverter when the parameters listed in Table VI are adopted.It’s obvious that ZVS has been achieved. The angle that thecurrent leads the voltage is about 15°, much less than presetangle (30° ). Though the analysis about the leading angle andthe value of the phase shift capacitor is not very accurate dueto the influence of nonsinusoidal input voltage, dead time,rectifier, filter inductor and capacitor, (16) can guide the phaseshift capacitor design. The current and voltage profiles acrossother power conversion stages are shown in Fig. 20.

In light of (15), C3−ZPA should be 77.38 nF. However, whena capacitance of 77.38 nF is employed, the current leads thevoltage, indicating a capacitance of 77.38 nF is too big toachieve ZPA. Therefore, the capacitance is reduced graduallyto find the right value. When the capacitance equals 71.86nF, ZPA is achieved. The corresponding voltage and current

Fig. 20. Current and voltage profiles of (a) dc input, (b) primary coil,(c) secondary coil, (d) rectifier output, and (e) load.

Fig. 21. Output voltage and current waveforms of the inverter when C3−ZPAis employed.

waveforms are shown in Fig. 21.The advantage of easily changed output voltage of S/CLC

compensation topology has also been verified by experiment.Fig. 22 shows the profiles of output voltage versus input volt-age when different compensation parameters are employed.The output voltage is approximately proportional to input volt-age, validating the correctness of the analysis on system outputvoltage to some extent. In addition, the output voltage can bereadily altered by changing some compensation components.This characteristic makes the compensation topology free fromthe constraints imposed by the LCT parameters.

S/CLC compensated WPT system offers a characteristic ofCVI-CVO as well. A sudden load change is employed toillustrate this characteristic. The output voltage when suddenload change happens is displayed in Fig. 23. At the beginning,the resistive load is 80.49 �, and the output voltage is 73.75 V.Then the load is reduced to 60.22 �, the output voltage fallsto 70.5 V. The output voltage reduces by 4.4% while theload drops by 25.2%. Compared to the simulation results, thedescent of output voltage is higher, resulting from inaccuracy


Fig. 22. Profiles of output voltage versus input voltage when differentcompensation parameters are employed.

Fig. 23. Output voltage when sudden load change happens.

of compensation capacitors. Even so, compared to the decreaseof load, the decrease of output voltage is minor, and the systemcan be approximately seen as a CVI-CVO system. The systemshould be an accurate CVI-CVO system if the input voltageis sinusoidal, and the load is directly in parallel with C3.

The system efficiency is also derived by measuring the inputand output power of the system. The output power is 67.57 Wwhile the input power is 74.79 W, implying a system efficiencyof 90.35%, which is quite high for an 100-W WPT system.

VI. CONCLUSION

A novel compensation topology—S/CLC—for WPT systemis proposed in this paper. It provides excellent CVI-CVOcharacteristic, significantly simplifying control circuit design.When load drops by 25.2%, the output voltage only decreasesby 4.6% under the condition of inaccurate compensationparameters. Compared to SS compensation topology, proposedtopology shows the advantages of easy achievement of ZPA,ZVS and being free from the constraints imposed by the LCTparameters. Compared to double-sided LCC compensationtopology, proposed topology needs less compensation compo-nents, reducing system cost, size and improving system powerdensity and efficiency. All abovementioned characteristics andadvantages have been analyzed in theory, and verified by bothsimulation and experiment.

The optimization of a circular pad is also studied. Theinfluences of ferrite structure, core size and coil dimension-related parameters on the coupling coefficient have beeninvestigated by ANSYS Maxwell. Circular ferrite panel isoptimum structure. The bigger the diameter and thickness ofthe panel, the higher the coupling coefficient. Nevertheless, theeffect of thickness on the coupling coefficient is quite poor.

Turns of coil is irrelevant with the coupling coefficient, whichcan be fixed to certain value throughout the optimization. Theheight of coil has obvious impact on the coupling coefficientsince it is directly related to the distance between ferrite cores.With respect to a circular pad with a diameter of 200 mm andan air gap of 50 mm, the best inner and outer diameter for thecoil are 50 and 200 mm, respectively.

REFERENCES

[1] C.-S. Wang, G. A. Covic, and O. H. Stielau, “Investigating an LCL loadresonant inverter for inductive power transfer applications,” IEEE Trans.Power Electron., vol. 19, no. 4, pp. 995–1002, Jul. 2004.

[2] W. Zhou and H. Ma, “Design considerations of compensation topologiesin ICPT system,” in Proc. 22nd Annu. IEEE Appl. Power Electron. Conf.Expo. (APEC), Anaheim, CA, USA, Feb. 2007, pp. 985–990.

[3] J. P. K. Sampath, D. M. Vilathgamuwa, and A. Alphones, “Efficiencyenhancement for dynamic wireless power transfer system with seg-mented transmitter array,” IEEE Trans. Transport. Electrific., vol. 2,no. 1, pp. 76–85, Mar. 2016.

[4] Q. Zhu, Y. Zhang, C. Liao, Y. Guo, L. Wang, and L. Wang, “Null-coupled electromagnetic field cancelling coil for wireless power transfersystem,” IEEE Trans. Transport. Electrific., vol. PP, no. 99, pp. 1–1,2016, doi: 10.1109/TTE.2016.2633798

[5] J. Sallan, J. L. Villa, A. Llombart, and J. F. Sanz, “Optimal design ofICPT systems applied to electric vehicle battery charge,” IEEE Trans.Ind. Electron., vol. 56, no. 6, pp. 2140–2149, Jun. 2009.

[6] C.-S. Wang, G. A. Covic, and O. H. Stielau, “Power transfer capabilityand bifurcation phenomena of loosely coupled inductive power transfersystems,” IEEE Trans. Ind. Electron., vol. 51, no. 1, pp. 148–157,Feb. 2004.

[7] A. J. Moradewicz and M. P. Kazmierkowski, “Contactless energytransfer system with FPGA-controlled resonant converter,” IEEE Trans.Ind. Electron., vol. 57, no. 9, pp. 3181–3190, Sep. 2010.

[8] S. Valtchev, B. Borges, K. Brandisky, and J. B. Klaassens, “Resonantcontactless energy transfer with improved efficiency,” IEEE Trans.Power Electron., vol. 24, no. 3, pp. 685–699, Mar. 2009.

[9] C.-S. Wang, O. H. Stielau, and G. A. Covic, “Design considerations for acontactless electric vehicle battery charger,” IEEE Trans. Ind. Electron.,vol. 52, no. 5, pp. 1308–1314, Oct. 2005.

[10] J. L. Villa, J. Sallán, A. Llombart, and J. F. Sanz, “Design of a highfrequency inductively coupled power transfer system for electric vehiclebattery charge,” Appl. Energy, vol. 86, no. 3, pp. 355–363, Mar. 2009.

[11] W. Zhang, S.-C. Wong, C. K. Tse, and Q. Chen, “Analysis and com-parison of secondary series- and parallel-compensated inductive powertransfer systems operating for optimal efficiency and load-independentvoltage-transfer ratio,” IEEE Trans. Power Electron., vol. 29, no. 6,pp. 2979–2990, Jun. 2014.

[12] J. L. Villa, J. Sallan, J. F. S. Osorio, and A. Llombart, “High-misalignment tolerant compensation topology for ICPT systems,” IEEETrans. Ind. Electron., vol. 59, no. 2, pp. 945–951, Feb. 2012.

[13] J. Hou, Q. Chen, S.-C. Wong, C. K. Tse, and X. Ruan, “Analysisand control of series/series-parallel compensated resonant converter forcontactless power transfer,” IEEE J. Emerg. Sel. Topics Power Electron.,vol. 3, no. 1, pp. 124–136, Mar. 2015.

[14] J. Hou, Q. Chen, X. Ren, X. Ruan, S.-C. Wong, and C. K. Tse, “Precisecharacteristics analysis of series/series-parallel compensated contactlessresonant converter,” IEEE J. Emerg. Sel. Topics Power Electron., vol. 3,no. 1, pp. 101–110, Mar. 2015.

[15] J. Hou, Q. Chen, X. Ren, S.-C. Wong, and C. K. Tse, “Steady-state analysis of series/series-parallel compensated contactless resonantconverter,” in Proc. IEEE 79th Veh. Technol. Conf. (VTC), Seoul,South Korea, May 2014, pp. 1–5.

[16] N. Keeling, G. A. Covic, F. Hao, L. George, and J. T. Boys, “Variabletuning in LCL compensated contactless power transfer pickups,” in Proc.IEEE Energy Convers. Congr. Expo. (ECCE), San Jose, CA, USA,Sep. 2009, pp. 1826–1832.

[17] Y. Su, C. Tang, S. Wu, and Y. Sun, “Research of LCL resonant inverterin wireless power transfer system,” in Proc. Int. Conf. Power Syst.Technol. (PowerCon), Chongqing, China, Oct. 2006, pp. 1–6.

[18] S. Li, W. Li, J. Deng, T. D. Nguyen, and C. C. Mi, “A double-sidedLCC compensation network and its tuning method for wireless powertransfer,” IEEE Trans. Veh. Technol., vol. 64, no. 6, pp. 2261–2273,Jun. 2015.


[19] W. Li, H. Zhao, S. Li, J. Deng, T. Kan, and C. C. Mi, “IntegratedLCC compensation topology for wireless charger in electric and plug-in electric vehicles,” IEEE Trans. Ind. Electron., vol. 62, no. 7,pp. 4215–4225, Jul. 2015.

[20] Q. Zhu, L. Wang, Y. Guo, C. Liao, and F. Li, “Applying LCC compen-sation network to dynamic wireless EV charging system,” IEEE Trans.Ind. Electron., vol. 63, no. 10, pp. 6557–6567, Oct. 2016.

[21] W. Li, H. Zhao, J. Deng, S. Li, and C. C. Mi, “Comparison study on SSand double-sided LCC compensation topologies for EV/PHEV wirelesschargers,” IEEE Trans. Veh. Technol., vol. 65, no. 6, pp. 4429–4439,Jun. 2016.

[22] W. Zhang and C. C. Mi, “Compensation topologies of high-powerwireless power transfer systems,” IEEE Trans. Veh. Technol., vol. 65,no. 6, pp. 4768–4778, Jun. 2016.

[23] J. M. Miller and A. Daga, “Elements of wireless power transfer essentialto high power charging of heavy duty vehicles,” IEEE Trans. Transport.Electrific., vol. 1, no. 1, pp. 26–39, Jun. 2015.

[24] M. Budhia, G. A. Covic, and J. T. Boys, “Design and optimizationof circular magnetic structures for lumped inductive power transfersystems,” IEEE Trans. Power Electron., vol. 26, no. 11, pp. 3096–3108,Nov. 2011.

[25] M. Budhia, J. T. Boys, G. A. Covic, and C.-Y. Huang, “Developmentof a single-sided flux magnetic coupler for electric vehicle IPT chargingsystems,” IEEE Trans. Ind. Electron., vol. 60, no. 1, pp. 318–328,Jan. 2013.

[26] W. Zhang, J. C. White, A. M. Abraham, and C. C. Mi, “Loosely coupledtransformer structure and interoperability study for EV wireless chargingsystems,” IEEE Trans. Power Electron., vol. 30, no. 11, pp. 6356–6367,Nov. 2015.

[27] C. Hu, Y. Sun, C. S. Tang, and Z. H. Wang, “Optimal design ofelectromagnetic coupling mechanism for ICPT system,” in Proc. 8thIEEE Conf. Ind. Electron. Appl. (ICIEA), Melbourne, VIC, Australia,Jun. 2013, pp. 1233–1237.

[28] W.-J. Li, X.-K. Li, J.-K. Zhang, G.-R. Zhu, M. Xie, and X.-S. Li,“Research on loosely coupled transformer’s structure for ICPT system,”in Proc. Int. Conf. Ind. Inform.-Comput. Technol., Intell. Technol., Ind.Inf. Integr. (ICIICII), Wuhan, China, Dec. 2015, pp. 272–275.

[29] H. Sakamoto, K. Harada, S. Washimiya, K. Takehara, Y. Matsuo, andF. Nakao, “Large air-gap coupler for inductive charger [for electricvehicles],” IEEE Trans. Magn., vol. 35, no. 5, pp. 3526–3528, Sep. 1999.

[30] V. K. Sharma and M. Kumar, “Optimization of design parameters for 45W ICPT system,” in Proc. Students Conf. Eng. Syst. (SCES), Allahabad,India, Apr. 2013, pp. 1–6.

[31] T. H. Nishimura, T. Eguchi, K. Hirachi, Y. Maejima, K. Kuwana,and M. Saito, “A large air gap flat transformer for a transcutaneousenergy transmission system,” in Proc. 25th Annu. IEEE Power Electron.Specialists Conf. (PESC) Rec., vol. 2. Taipei, Taiwan, Jun. 1994,pp. 1323–1329.

[32] K. Iwawaki, M. Watada, S. Takatani, and Y.-S. Um, “The design of core-type transcutaneous energy transmission systems for artificial heart,”in Proc. 30th Annu. Conf. IEEE Ind. Electron. Soc. (IECON), vol. 1.Nov. 2004, pp. 948–952.

[33] Z. Wei, C. Qianhong, S. C. Wong, M. Tse, and C. Lingling, “Reluctancecircuit and optimization of a novel contactless transformer,” (in Chinese),Trans. China Electrotech. Soc., vol. 30, no. 27, pp. 108–116, Sep. 2010.

[34] R. L. Steigerwald, “A comparison of half-bridge resonant convertertopologies,” IEEE Trans. Power Electron., vol. 3, no. 2, pp. 174–182,Apr. 1988.

[35] K. Knaisch and P. Gratzfeld, “Comparison of magnetic couplers forinductive electric vehicle charging using accurate numerical simula-tion and statistical methods,” in Proc. 5th Int. Elect. Drives Prod.Conf. (EDPC), Nuremberg, Germany, Sep. 2015, pp. 1–10.

[36] D. Voglitsis, T. Todorceviá, V. Prasanth, and P. Bauer, “Loss modeland control stability of bidirectional LCL-IPT system,” in Proc. 4th Int.Elect. Drives Prod. Conf. (EDPC), Nuremberg, Germany, Sep. 2014,pp. 1–8.

Yijie Wang (S’09–M’15–SM’15) was born inHeilongjiang, China, in 1982. He received the B.S.,M.S., and Ph.D. degrees in electrical engineeringfrom the Harbin Institute of Technology, Harbin,China, in 2005, 2007, and 2012, respectively.

From 2012 to 2014, he was a Lecturer with theDepartment of Electrical and Electronics Engineer-ing, Harbin Institute of Technology, where he hasbeen an Associate Professor since 2015. His currentresearch interests include dc-dc converters, soft-switching power converters, power factor correction

circuits, digital control electronic ballasts, and LED lighting systems.

Yousu Yao was born in Jiangsu, China, in 1991.He received the B.S. degree in electrical engineeringfrom the Harbin Institute of Technology, Harbin,China, in 2014, where he is currently pursuing thePh.D. degree.

His current research interests include wirelesspower transfer.

Xiaosheng Liu (M’14) was born in Heilongjiang,China, in 1966. He received the B.S. and M.S.degrees in electrical engineering and the Ph.D.degree in mechatronics engineering from the HarbinInstitute of Technology, Harbin, China, in 1988,1993, and 1999, respectively.

He has been a Professor with the Departmentof Electrical Engineering, Harbin Institute of Tech-nology, since 2006. His current research interestsinclude power line communication and its routingmethods, communication networks and control tech-

nology, and information and communication of smart grids.

Dianguo Xu (M’97–SM’12–F’17) was born in Hei-longjiang, China, in 1960. He received the B.S.degree in control engineering from the Harbin Ship-building Engineering Institute, Harbin, China, in1981, and the M.S. and Ph.D. degrees in electricalengineering from the Harbin Institute of Technology,Harbin, in 1984 and 1990, respectively.

He has been a Professor with the Departmentof Electrical Engineering, Harbin Institute of Tech-nology, since 1994. His current research interestsinclude robotics, lighting electronics, power quality

mitigation, consumer electronics, power electronics, and motor drives.Dr. Xu is a member of the China Electrotechnical Society and the China

Power Supply Society.

本文献由“学霸图书馆-文献云下载”收集自网络，仅供学习交流使用。

学霸图书馆（www.xuebalib.com）是一个“整合众多图书馆数据库资源，

提供一站式文献检索和下载服务”的24 小时在线不限IP

图书馆。

图书馆致力于便利、促进学习与科研，提供最强文献下载服务。

图书馆导航：

图书馆首页文献云下载图书馆入口外文数据库大全疑难文献辅助工具

http://www.xuebalib.com/cloud/

http://www.xuebalib.com/

http://www.xuebalib.com/cloud/


http://www.xuebalib.com/vip.html

http://www.xuebalib.com/db.php

http://www.xuebalib.com/zixun/2014-08-15/44.html


S/CLC Compensation Topology Analysis and Circular Coil ...

Documents

Transcript of S/CLC Compensation Topology Analysis and Circular Coil ...