Seminar Report onDesign of Low Phase-Noise

Microwave OscillatorSubmitted By

UPENDRA HATIYA - 12EC63R07SHILPI RUCHI KERKETTA - 12EC63R05

BHARAT BHUSHAN 12EC63R0M.Tech. 1st Year (RF & MW)

RF & Microwave EngineeringDepartment of Electronics & Electrical Communication

Engineering

IIT Kharagpur

Seminar Report onDesign of Low Phase-Noise

Microwave OscillatorSubmitted By

UPENDRA HATIYA - 12EC63R07SHILPI RUCHI KERKETTA - 12EC63R05

BHARAT BHUSHAN 12EC63R0M.Tech. 1st Year (RF & MW)

RF & Microwave EngineeringDepartment of Electronics & Electrical Communication

Engineering

IIT Kharagpur

Seminar Report onDesign of Low Phase-Noise

Microwave OscillatorSubmitted By

UPENDRA HATIYA - 12EC63R07SHILPI RUCHI KERKETTA - 12EC63R05

BHARAT BHUSHAN 12EC63R0M.Tech. 1st Year (RF & MW)

RF & Microwave EngineeringDepartment of Electronics & Electrical Communication

Engineering

IIT Kharagpur

Phase noise sudden change in the phase of signal which corresponds to highfrequency components

Considering a sinusoidal signal of certain amplitude in time domain

Its frequency component in frequency domain. A peak at the frequency value.

A Amplitude change occurs at a time t1 because of an additive noise and changesthe phase of signal i.e. phase noise

This phase change is change in amplitude. This amplitude will then decay tocorrect itself and this decay corresponding to time t1 to t2.

In frequency this decay corresponds to a high frequency component. And its valueis related to the time taken by the signal to decay and again come to the previousfashion.

The resistive component presents in the design is responsible of the fast or slowdecay of the amplitudes. As shown in the time domain representation of waveform. We can easily understand that the decay cause due to high Q is less ascompare to lower Q.

Response due to low Q value of design and it shows that the frequency componentgenerated is more which is undesired because high frequency corresponding tohigh power so more power will loss in case of high frequency components.

But as we use high Q design the change in phase due to noise will reduce slowlyand have less frequency component so that we will get less energy loss.

Data sheet of transistor used i.e. BPF504. Just by having these values in mind wehave to bias transistor so that it would be in saturation.

Basic amplifier circuit after proper biasing and connections with VCC and GND

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Osc. starts with noise i.e. white Gaussian noise contains almost all frequencycomponents but of very low amplitude.

After that according to the dimension of resonator and the coupling inbetween resonators, some components of frequency starts transferring inbetween resonator.

The amplifier increase the frequency component of that noise, whichbandpassed through the resonator.

As we are using feedback then the amplified signal is divided and some ofits part again sent to the resonator so that barkhausen criteria would befulfilled for oscillation.

At the output port of basic amplifier circuit we need to use a bandpassing structurewhich will act as selective feedback design and make the oscillation sustain.

Butterworth Lowpass Prototype FiltersFor Butterworth or maximally flat lowpass prototype filters having a transferfunctiongiven in figure1 with an insertion loss LAr = 3.01 dB at the cutoff c = 1, theelement|S21 (j)|2 = 1/( 1+2nvalues as referring to

Lowpass Prototype Filters for all poles filters with a) a ladder networkstructure & b) its dual

this may be computed byg0 = 1.0gi = 2 sin((2i-1)/2n) for i = 1 to ngn+1 = 1.0the two-port Butterworth filters considered here are always symmetrical

in network structure, namely, g0 = gn+1, g1 = gn and so on.To determine the degree of a Butterworth lowpass prototype, a specification thatis usually the minimum stopband attenuation LAs dB at = s for s > 1 isgiven.Hencen log(10.1LAS 1)/2logsFor example, if LAs = 40 dB and s = 2, n 6.644, i.e., a 7-pole (n = 7)Butterworthprototype should be chosen.

Stub Bandpass Filters

Synchronously Tuned Coupled-Resonator CircuitsElectric CouplingAn equivalent lumped-element circuit model for electrically coupledRF/microwave resonators is given in Figure below, where L and C are the self-inductance and self capacitance, so that (LC)1/2 equals the angular resonantfrequency of uncoupled resonators, and Cm represents the mutual capacitance. Asmentioned earlier, if the coupled structure is a distributed element, the lumped-element circuit equivalence is valid on a narrow-band basis, namely, near itsresonance. The same comment is applicable for the other coupled structuresdiscussed later. Now, if we look into reference planes T1T1 and T2T2, we cansee a two-port network that may be described by the following set of equations:I1 = jCV1 jCmV2I2 = jCV2 jCmV1

in which a sinusoidal waveform is assumed. It might be well to mention thatimplies that the self-capacitance C is the capacitance seen in one resonant loop of

when the capacitance in the adjacent loop is shorted out. creasing voltage inresonant loop 2 and loop 1, respectively. four Y parametersY11 = Y22 = jCY12 = Y21 = jCmcan easily be found from definitions.

(a) Synchronously tuned coupled resonator circuit with electric coupling. (b) Analternativeform of the equivalent circuit with an admittance inverter J = Cm to represent thecoupling.According to the network theory an alternative form of the equivalent circuitin Figure can be obtained and is shown in Figure. This form yields the same two-port parameters as those of the circuit of Figure3 (a), but it is more convenient forour discussions. Actually, it can be shown that the electric coupling between thetwo resonant loops is represented by an admittance inverter J = Cm. If thesymmetry plane TT in Figure 3(b) is replaced by an electric wall (or a shortcircuit), the resultant circuit has a resonant frequency

This resonant frequency is lower than that of an uncoupled single resonator. Aphysical explanation is that the coupling effect enhances the capability to storecharge of the single resonator when the electric wall is inserted in the symmetricalplane of the coupled structure. Similarly, replacing the symmetry plane in Figure3(b) by a magnetic wall (or an open circuit) results in a single resonant circuithaving a resonant frequency

In this case, the coupling effect reduces the capability to store charge so that theresonantfrequency is increased.K E

which is not only identical to the definition of ratio of the coupled electric energyto the stored energy of uncoupled single resonator, but also consistent with thecoupling coefficient defined by figure above for coupled-resonator filter.

FORMULATION FOR EXTRACTING EXTERNAL QUALITY FACTOR QeTwo typical input/output (I/O) coupling structures for coupled microstrip resonatorfilters, namely the tapped line and the coupled line structures, are shown with themicrostrip open-loop resonator, though other types of resonators may be used (seeFigure 4(a). For the tapped line coupling, usually a 50 ohm feed line is directlytapped onto the I/O resonator, and the coupling or the external quality factor iscontrolledby the tapping position t, as indicated in Figure 5(a)0(a). For example, the

Typical I/O coupling structures for coupled resonator filters. (a) Tapped-linecoupling. (b) Coupled-line coupling.smaller the t, the closer is the tapped line to a virtual grounding of the resonator,which results in a weaker coupling or a larger external quality factor. The couplingof the coupled line stricture in Figure 4(a)(b) can be found from the coupling gap gand the line width w. Normally, a smaller gap and a narrower line result in astronger I/O coupling or a smaller external quality factor of the resonator.

Doubly Loaded ResonatorAlthough the Qe is defined for a singly loaded resonator, if the resonator issymmetrical,one could add another symmetrical load or port to form a two-port network, asFigure 6 shows, where TT represents the symmetrical plane and the single LCresonator has been separated into two symmetrical parts. When the symmetricalplane TT is short-circuited, we have

where Yino and S11o are the odd-mode input admittance and reflection coefficientat port 1, respectively. On the other hand, replacing the TT plane with an opencircuit yields the corresponding parameters for the even mode:

where _0 = 1/(LC) and the approximation (2 02 )/2 with = 0 + has been made.

Equivalent circuit of the I/O resonator with double loading.

whose magnitude is given by

Shown in Figure 7 is a plot of |S21| against /0. At resonance, = 0 and thus|S21| reaches its maximum value, namely |S21(0)| = 1. When the frequency shiftssuch that

the value of |S21| has fallen to 0.707 (or 3 dB) of its maximum value

Resonant amplitude response of S21 for the circuitWhere 3 dB is the bandwidth for which the attenuation for S21 is up 3 dB fromthatat resonance, as indicated in Figure 7. Define a doubly loaded external qualityfactor Qe as

Using above equation to extract the Qee first, then the singly loaded externalquality factor Qe is simply the twice of Qe.

Typical coupling structures of coupled resonators with (a) electric coupling,(b) magnetic coupling, (c) and (d) mixed coupling.These coupled structures result from different orientations of a pair ofopen-loop resonators, which are separated by a spacing s. It is obvious that anycoupling in those structures is proximity coupling, which is, basically, throughfringe fields. The nature and the extent of the fringe fields determine the natureand the strength of the coupling. It can be shown that at resonance of thefundamentalmode, each of the open-loop resonators has the maximum electric fielddensity at the side with an open gap, and the maximum magnetic field density atthe opposite side. Because the fringe field exhibits an exponentially decaying

character outside the region, the electric fringe field is stronger near the sidehaving the maximum electric field distribution, whereas the magnetic fringe fieldis stronger near the side having the maximum magnetic field distribution. Itfollows that the electric coupling can be obtained if the open sides of two coupledresonators are proximately placed, as Figure shows, and the magnetic coupling canbe obtained if the sides with the maximum magnetic field of two coupledresonators are proximately placed, as Figure shows. For the coupling structures inFigure the electric and magnetic fringe fields at the coupled sides may havecomparative distributions, so that both electric and the magnetic couplings occur.In this case the coupling may be referred to as mixed coupling. However, it will bedemonstrated later that these two coupling structures exhibit distinguishingcoupling characteristics.

Typical resonant responses of coupled resonator structures. (a) For the structure inFigure( a). (b) For the structure in Figure (b).

The center frequency and fractional bandwidth of the filter are set to 2 GHzand 4%, respectively. The normalized frequency of the attenuations pole is selectedto a = 2. Hence, the element values of the low-pass prototype filter are g0 =1,g1 = 0.95449, g2 = 1.38235, J1 = -0.16271, J2 = 1.06062. The couplingcoefficients and external quality factor are calculated as M12 = M34 = 0.03482,M23 = 0.03069, M14 = 0.0068, and Qe = 23.86.

Complex quality factor Qsc and group delay.

OSCILLATOR DESIGN CIRCUIT USING TWO POLECOUPLED RESONATOR AS FEEDBACK (BANDPASSING ELEMENT)

REFERENCES :-[1] Chao-Hsiung Tseng and Chih-Lin Chang, Design of Low Phase-Noise Microwave Oscillator and Wideband VCO Based onMicrostrip Combline Bandpass Filters, IEEE transactions onmicrowave theory and techniques, vol. 60, no. 10, pp. 3151-3160, oct2012.[2] J.-S. Hong and M. J. Lancaster, Microstrip Filter for RF/MicrowaveApplication. New York: Wiley, 2001.