DATA SHEET - Thierry LEQUEU · 2009. 3. 12. · or cylinder, manufactured from ferrite or...

MAGNETIC PRODUCTS

DATA SHEET

Supersedes data of 1998 Mar 25File under Magnetic Products, MA01

1998 Oct 01

ApplicationsSoft Ferrites

1998 Oct 01 2

Philips Components

Soft Ferrites Applications

APPLICATIONS

Introduction

Soft ferrite cores are used wherever effective couplingbetween an electric current and a magnetic flux isrequired. They form an essential part of inductors andtransformers used in today’s main application areas:

• Telecommunications

• Power conversion

• Interference suppression.

The function that the soft magnetic material performs maybe one or more of the following:

FILTERING

Filter network with well defined pass-band.

High Q-values for selectivity and good temperaturestability.

Material requirements:

• Low losses

• Defined temperature factor to compensate temperaturedrift of capacitor

• Very stable with time.

Preferred materials: 4C6, 3D3, 3H1, 3H3, 3B7.

INTERFERENCE SUPPRESSION

Unwanted high frequency signals are blocked, wantedsignals can pass. With the increasing use of electronicequipment it is of vital importance to suppress interferingsignals.


• High impedance in covered frequency range.

Preferred materials: 3S1, 4S2, 3S3, 3S4, 4C65, 4A11,4A15, 3B1, 4B1, 3C11, 3E25, 3E5.

Fig.1 Filter application.

andbook, halfpage

MBW403

attenuation (dB)

frequency

Fig.2 Suppression application.

handbook, halfpage

MBW404

U

frequency

U1

U2

U1 U2

load

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Philips Components


DELAYING PULSES

The inductor will block current until saturated. Leadingedge is delayed depending on design of magnetic circuit.


• High permeability (µi).

Preferred materials: 3E25, 3E5, 3E6, 3E7.

STORAGE OF ENERGY

An inductor stores energy and delivers it to the load duringthe off-time of a Switched Mode Power Supply (SMPS).


• High saturation level (Bs).

Preferred materials: 3C15, 3C30, 3C85, 3C90,2P-iron powder.

Fig.3 Pulse delay application.

handbook, halfpage

MBW405

U

time

U1 U2

U1 U2

delay

Fig.4 Smoothing/storage application.

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MBW406

U

time

load

U1 U2

SMPS

U1

U2

1998 Oct 01 4

Philips Components


PULSE TRANSFORMERS/GENERAL PURPOSE TRANSFORMERS

Pulse or AC signals are transmitted and if requiredtransformed to a higher or lower voltage level. Alsogalvanic separation to fulfil safety requirements andimpedance matching are provided.


• High permeability

• Low hysteresis factor for low signal distortion

• Low DC sensitivity.

Preferred materials: 3B7, 3B8, 3C81, 3H1, 3H3, 3E1, 3E4,3E25, 3E5, 3E6.

POWER TRANSFORMERS

A power transformer transmits energy, transforms voltageto the required level and provides galvanic separation(safety).


• Low power losses

• High saturation (Bs).

Preferred materials: 3C15, 3C30, 3C81, 3C85, 3C90, 3F3,3F4, 4F1.

Fig.5 Pulse and general purpose transformer.

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MBW407

U

time

U1 U2

U1

U20

DC level

Fig.6 Power transformer application.

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MBW408

U

time

loadU1

U2

U1

U2

0

1998 Oct 01 5

Philips Components


TUNING

LC filers are often used to tune circuits in audio, video andmeasuring equipment. A very narrow bandwidth is oftennot wanted.


• Moderate losses up to high frequency

• Reasonable temperature stability.

Preferred materials: 3D3, 4A11, 4B1, 4D2, 4E1.

Fig.7 Tuning application.

olumns

CBW344

Z

frequency

bandwidth

L C

1998 Oct 01 6

Philips Components


Ferrites for Telecommunications

Telecommunications is the first important branch oftechnology where ferrites have been used on a largescale. Today, against many predictions, it still is animportant market for ferrite cores.

Most important applications are in:

• Filter inductors

• Pulse and matching transformers.

FILTER COILS

P cores, RM cores and X cores have been developedspecially for this application.

The P core is the oldest design. It is still rather popularbecause the closed shape provides excellent magneticscreening.

RM cores are a later design, leading to a more economicusage of the surface area on the PCB. X cores also havethis advantage plus more room to get the leads out ofthe coil.

For filter coils, the following design parameters areimportant:

• Precise inductance value

• Low losses, high Q value

• High stability over periods of time

• Fixed temperature dependence.

Q VALUE

The quality factor (Q) of a filter coil should generally be ashigh as possible. For this reason filter materials such as3H1, 3B7, 3B9, 3H3, 3D3 and 4C6 have low magneticlosses in their frequency ranges (100 kHz, 2 MHz and10 MHz respectively).

Losses in a coil can be divided into:

• Winding losses, due to the DC resistance of the wireeddy-current losses in the wire electric losses ininsulation

• Core losses, due to hysteresis losses in the corematerial eddy-current and residual losses in the corematerial.

Losses appear as series resistances in the coil:

Rtot

L----------

R0

L-------

Rec

L---------

Rd

L-------

Rh

L-------

Re r+

L------------- Ω H⁄( )+ + + +=

As a general rule, maximum Q is obtained when the sumof the winding losses is made equal to the sum of the corelosses.

DC resistive losses

The DC resistive losses in a winding are given by:

The space (copper) factor fCu depends on wire diameter,the amount of insulation and the method of winding.

Eddy-current losses in the winding

Eddy-current losses in a winding are given by:

Where CwCu is the eddy-current loss factor for the windingand depends on the dimensions of the coil former andcore, and VCu is the volume of conductor in mm3, d is thediameter of a single wire in mm.

Dielectric losses

The capacitances associated with the coil are not loss free.They have a loss factor which also increases the effectivecoil resistance:

Hysteresis losses

The effective series resistance due to hysteresis losses iscalculated from the core hysteresis constant, the peak fluxdensity, the effective permeability and the operatingfrequency:

Eddy-current and residual losses

The effective series resistance due to eddy-current andresidual losses is calculated from the loss factor:

R0

L-------

1µe------=

1fCu--------× cons t Ω H⁄( )tan×

Rec

L---------

CwCuVCuf2d

2

µe------------------------------------ Ω H⁄( )=

Rd

L------- ω3

LC 2Q---- δctan+

Ω H⁄( )=

Rh

L------- ωηBBµe Ω H⁄( )=

Re r+

L------------- ωµe δtan µi⁄( ) Ω H⁄( )=

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Philips Components


INDUCTOR DESIGN

The specification of an inductor usually includes:

• The inductance

• Minimum Q at the operating frequency

• Applied voltage

• Maximum size

• Maximum and minimum temperature coefficient

• Range of inductance adjustment.

To satisfy these requirements, the designer has thechoice of:

• Core size

• Material grade

• AL value

• Type of conductor (solid or bunched)

• Type of adjuster.

FREQUENCY, CORE TYPE AND MATERIAL GRADE

The operating frequency is a useful guide to the choice ofcore type and material.

• Frequencies below 20 kHz:the highest Q will be obtained with large, highinductance-factor cores of 3B7, 3H1 or 3H3 material.Winding wire should be solid, with minimum-thicknessinsulation.

Note: high inductance factors are associated with hightemperature coefficients of inductance.

• Frequencies between 20 kHz and 200 kHz:high Q will generally be obtained with a core also in 3B7,3H1 or 3H3. Maximum Q will not necessarily beobtained from the large-size core, particularly at higherfrequencies, so the choice of inductance factor is lessimportant. Bunched, stranded conductors should beused to reduce eddy-current losses in the copper.Above 50 kHz, the strands should not be thickerthan 0.07 mm.

• Frequencies between 200 kHz and 2 MHz:use a core of 3D3 material. Bunched conductors ofmaximum strand diameter 0.04 mm are recommended.

• Frequencies between 2 MHz and 12 MHz:use a core of 4C6. Bunched conductors of maximumstrand diameter 0.04 mm are recommended forfrequencies up to 5 MHz. Solid conductors should beused at frequencies between 5 MHz and 12 MHz.

SIGNAL LEVEL

In most applications, the signal voltage is low. It is goodpractice to keep wherever possible the operating fluxdensity of the core below 1 mT, at which level the effect ofhysteresis is usually negligible. At higher flux densities,it may be necessary to allow for some hysteresis loss andinductance change.

The following expression for third harmonic voltage U3

may be used as a guide to the amount of distortion:

For low distortion, materials with small hysteresis lossfactors should be used (e.g. 3H3).

DC POLARIZATION

The effect of a steady, superimposed magnetic field due toan external field or a DC component of the winding currentis to reduce the inductance value of an inductor. As withother characteristics, the amount of the decrease dependson the value of the effective permeability. The effect can bereduced by using a gapped core or by choosing a lowerpermeability material.

AL VALUE

Since the air gap in ferrite cores can be ground to anylength, any value of AL can be provided within the limits setby the core size. In practice, the range of AL values hasbeen standardized with values chosen to cover themajority of application requirements.

If a core set is provided with an asymmetrical air gap, thisair gap is ground in the upper half. This half is marked withthe ferrite grade and AL value.

Most pre-adjusted cores are provided with aninjection-moulded nut for the adjuster.

Continuously variable adjusters can be supplied forpre-adjusted cores of most AL values. These are speciallyrecommended for filter coils. Maximum adjustment rangeis 10% to 30%, depending on core type and adjuster.

The AL factor is the inductance per turn squared (in nH) fora given core:

The measured AL value of a core will depend slightly onthe coil used for this measurement.

U3

U1------- 0.6 δhtan=

L N2

AL× nH( )=

1998 Oct 01 8

Philips Components


For very low AL values (e.g. 16 to 25) the contribution ofthe stray inductance will be quite high, resulting in amarked influence of the position of the coil in the core andits number of turns.

INDUCTANCE ADJUSTERS

A major feature of a filter core assembly is its adjustmentmechanism. It allows the cores to be set to a very accuratevalue (0.1%).

The inductance adjustment is achieved by inserting a tubeor cylinder, manufactured from ferrite or carbonyl-ironpowder into the central hole of the core. This acts as apartial magnetic shunt across the air gap. The adjusterconsists of this tube moulded into a thermoplastic carrierwhich has been threaded at one end. This screws into anut which is injection moulded or cemented into the lowerhalf of the core. The magnetic tubes are centre-lessground to give very close diameter tolerances.

INDUCTANCE STABILITY

The stability of a correctly assembled inductor dependsmainly on the stability of the ferrite's permeability.

The permeability of a ferrite material may change withtemperature, time, mechanical pressure, magneticpolarization and other factors. The most importantchanges affecting the inductance stability of theassembly are:

• Variation of permeability with temperature(temperature coefficient)

• Variation of permeability with time (disaccommodation).

Changes in inductance may also occur due to:

• Movement of the adjuster after final setting

• Movement of the coil former

• Relative movement of the core halves

• Movement of the mechanical components of theassembly.

Small movements of this kind are usually caused bychanges in temperature, mechanical vibration or shock.

The achievement of acceptable long-term inductancestability is mainly a matter of careful assembly and suitablestabilizing treatment before final adjustment. If the inductoris to be used in a critical circuit, it should be artificially agedby temperature cycling. The long-term change ininductance of an assembly so treated should not begreater than 500 × 10−6, assuming an ambienttemperature between 25 °C and 40 °C that does not varyby more than 15 °K.

The change in inductance of an RM core assembly usingclips with earth pins when subjected to “IEC 60068-2-6,test Fc” (vibration conditions) is less than 1000 × 10−6.Such severe conditions are unlikely to be encountered inpractice.

Bump tests of RM-core assemblies with earth pins(“IEC 60068-2-29, test Eb”) have also been carried out.The observed change in the inductance of RM6-R cores of3H1 material was less than 300 × 10−6.

Figure 8 shows the principle outline of a typical adjuster.

Fig.8 Inductance adjuster outline.

handbook, halfpage

CBW345

ferrite oriron powdertube

precision thread

special design head

1998 Oct 01 9

Philips Components


The data sheets include lists of recommended adjustersfor the AL values in various grades. The table also lists themaximum inductance variation. In some cases, the choiceof adjuster is optional and depends on the application.For that reason, a suggestion is given for minimum,average and maximum inductance variation whereapplicable.

Figure 9 shows a typical curve of a specified adjuster in acore set, pre-adjusted on AL.

Dependent on size, the screw-head of the adjuster issuited for tools of M1.4, M1.7 and M2.6. An adjusting tool,combining M1.4 and M1.7 is available (catalogue number4322 058 0326) as well as a tool combining M2 and M2.6(catalogue number 43222 058 0327). For customers whowish to make the adjuster tool themselves, the fouroutlines are depicted in Fig.10.

The threads of both the nut and the adjuster have closetolerances (4H) to allow smooth rotation without backlashor friction.

Fig.9 Typical curve of a specified adjusterin a core set, pre-adjusted on AL.

handbook, halfpage

0 4 12

24

0

8

16

20

4

12

MBE954

82 106number of revolutions

∆LL

(%)

Fig.10 Dimensions (in mm) of tools for adjusters.

handbook, halfpage

0.8+ 0.2

0∅ 3.6± 0.05

0.9+ 0.2

00.7 ± 0.03

1.1 + 0.20

MBE955

handbook, halfpage

0.6 + 0.10

0− 0.1∅ 1.5

0.8+ 0.2

00.4 ± 0.03

1.1 + 0.20

MBE956

handbook, halfpage

∅ 4.6± 0.05

1+ 0.2

0

0.9+ 0.2

00.9 ± 0.03

1.1+ 0.2

0

MBE957

handbook, halfpage

MBE958

∅ 2.3± 0.05

0.7+ 0.1

0

0.9+ 0.2

00.6 ± 0.03

1.1+ 0.2

0

1998 Oct 01 10

Philips Components


PULSE AND SIGNAL TRANSFORMERS

Pulse and signal transformers, also known as widebandtransformers, are frequently used in communicationsystems, including modern digital networks such as,for example ISDN and XDSL.

They provide impedance matching and DC isolation ortransform signal amplitudes. Signal power levels areusually low. In order to transmit analog signals or digitalpulses without much distortion, good widebandcharacteristics are needed.

The principal function of the transformer core is to provideoptimum coupling between the windings.

The general equivalent circuit of a signal transformer isshown in Fig.11.

The elements of the circuit depicted in Fig.11 may bedefined as follows:

Es = source voltage

Rs = source resistance

Rw = total winding resistance = R1 + R2, where R1 isthe primary winding resistance and R2 is the secondarywinding resistance referred to the primary

L = total leakage inductance = the primary inductancewith the secondary shorted

Lp = open circuit inductance

Rp = the shunt loss resistance representing thecore loss

N1, N2 = the primary and referred secondary self orstray capacitance respectively

Rb = load resistance referred to the primary turns ratio.

A high permeability core with polished pole faces results ina large flux contribution, improving the coupling. Opencircuit inductance will be high, leakage inductance is keptlow compared to this main inductance.

Ring cores are very suitable since they have no air gapand make full use of the high permeability of the ferrite.

The frequency response of a practical transformer isshown in Fig.12.

Fig.11 Simplified equivalent circuit of a transformer.

handbook, halfpage

CBW346

Rw

Rs

Es

L

Lp

Rp

C2

Rb

C1 N1 N2

Fig.12 Transmission characteristicof a wideband transformer.

handbook, halfpage

MBW411

insertionloss

frequency

LF

region

mid-band

region

HF

region

LFdroop

HFdroop

mid-bandattenuation

1998 Oct 01 11

Philips Components


The corresponding distortion of a rectangular pulse by thesame circuit is shown in Fig.13.

The shunt inductance (Lp) is responsible for the lowfrequency droop in the analog transformer since itsreactance progressively shunts the circuit as thefrequency decreases. In the case of the pulse transformer,the shunt inductance causes the top of the pulse to droop,because, during the pulse, the magnetizing current in Lprises approximately linearly with time causing anincreasing voltage drop across the source resistance.

Fig.13 An ideal rectangular pulse and themain pulse distortions that may beintroduced by a transformer.

handbook, halfpage

MBW412

pulseamplitude

pulseamplitude

overshootleadingedge

td

topof pulse

0.1

0.9droop

trailing edge

td tftr

The winding resistance is the main cause of the mid-bandattenuation in low frequency analog transformers. In apulse transformer, it attenuates the output pulse butusually has little effect on the pulse distortion.

The high frequency droop of an analog transformer may bedue to either the increasing series reactance of theleakage inductance or the decreasing shunt reactance ofthe self-capacitances, or a combination of both as thefrequency increases. In a pulse transformer, the leakageinductance, self-capacitances and the source or loadresistance combine to slow down, or otherwise distort theleading and trailing edge responses.

Suitable core types for this application in the materialgrades 3E25, 3E1, 3E4, 3E5 and 3E6 are:

• P cores

• RM cores

• EP cores

• Ring cores

• Small ER cores

• Small E cores.

If the signal is superimposed on a DC current, coresaturation my become a problem. In that case, a lowerpermeability material grade such as 3H1, 3B7, 3B8,3C81 or 3C85 is recommended.

1998 Oct 01 12

Philips Components


Ferrites for Power conversion

Power conversion is a major application area for modernferrites. Originally designed for use as line outputtransformers in television receivers, power cores are nowbeing used in a wide range of applications. Theintroduction of Switched Mode Power Supplies (SMPS)has stimulated the development of a number of new ferritegrades and core shapes to be used in the manufacture ofpower transformers, output chokes and input filters.

Power transformers and inductors generally operate underloss or saturation limited conditions which require specialpower ferrites with high saturation levels and low losses.

Output chokes must tolerate high DC currents; this meansa gapped magnetic circuit or a special material with veryhigh saturation level such as iron powder.

Input chokes prevent mains pollution generated by theSMPS. Therefore grades are used which providemaximum blocking impedances at the switchingfrequencies.

SWITCHED MODE POWER SUPPLY CIRCUITS

The basic arrangement of a Switched Mode Power Supply(SMPS) is shown in Fig.14.

In this configuration, the power input is rectified andfiltered, and the resulting DC voltage is chopped by aswitch at a high frequency. The chopped waveform isapplied to the primary of a transformer and the secondaryoutput is rectified and filtered to give the requiredDC output. The output voltage is sensed by a controlcircuit which supplies a correction signal to the drive circuitto vary the ON/OFF time of the switched waveform andcompensate for any change at the output.

Numerous circuit designs can be used to convert DC inputvoltage to the required DC output voltage. Therequirements for the transformer or inductor dependlargely on the choice of this circuit technology.

If the circuits are analyzed in this way, three basicconverter designs can be distinguished, based upon themagnetic converting device.

These are:

• Flyback converters

• Forward converters, and

• Push-pull converters.

Fig.14 Block diagram of aSwitched Mode Power Supply.

handbook, halfpage

MBE959

RECTIFIER CONVERTER

CONTROLCIRCUIT

DRIVECIRUIT

1998 Oct 01 13

Philips Components


FLYBACK CONVERTER

Figure 15 shows the basic circuit of a flyback converterand its associated waveforms.

When the switch is closed (transistor conducts), the supplyvoltage is connected across the inductor and the outputdiode is non-conducting. The current rises linearly, storingenergy, until the switch is opened. When this happens, thevoltage across the inductor reverses and the storedenergy is transferred into the output capacitor and load.By varying the conduction time of the transistor at a givenfrequency the amount of energy stored in the inductorduring each ON cycle can be controlled. This allows theoutput of the SMPS to be controlled and changed.

This basic circuit can be developed into a practical circuitusing an inductor with two windings (see Fig.16).

In a flyback converter, all the energy to be transferred tothe output capacitor and load is, at first, stored in theinductor. It is therefore possible to obtain line isolation byadding a secondary winding to the inductor (although aninductor with more than one winding appears in schematicdiagrams as a transformer, it is referred to as an inductorin accordance with its function).

Another advantage of the flyback converter is that nosmoothing choke is required in the output circuit. This isimportant in high-voltage supplies and in power supplieswith a number of output circuits (see Fig.17).

Fig.15 Basic circuit of a DC to DC flybackconverter with associated waveforms.

handbook, halfpage

MBE960

IL

L

VL

Ic

Vi

ID Io

Vi

Vi

0

Vo

VL

current flow

switch closedswitch open

0

0

0

Ic

ID

IL

Fig.16 Practical flyback converter circuits.

handbook, halfpage

CBW347

1998 Oct 01 14

Philips Components


A disadvantage of this type of converter is that the outputcapacitor is charged only during the transistor’s OFF cycle.Hence the output capacitor ripple current is high whencompared with the other types of converters.

Another disadvantage of the flyback converter concernsthe energy stored in the inductor. The inductor is driven inone direction only; this requires a larger core in a flybackdesign than for an equivalent design using a forward orpush-pull converter.

FORWARD CONVERTER

The basic circuit of the forward converter, together withits associated voltage and current waveforms is shownin Fig.18.

When the switch is closed (transistor conducts), thecurrent rises linearly and flows through the inductor intothe capacitor and the load. During the ON cycle, energy istransferred to the output and stored in the inductor ‘L’.When the switch is opened, the energy stored in theinductor causes the current to continue to flow to theoutput via the diode.

As with the flyback converter, the amount of energy storedin the inductor can be varied by controlling the ON/OFFcycles. This provides control of the output of the forwardconverter.

A more practical forward converter circuit with aline-isolation transformer is shown in Fig.19.

Fig.17 Multiple output flyback converter circuit.

handbook, halfpage

MBE962

Fig.18 Basic circuit of a DC to DC forwardconverter with associated waveforms.

handbook, halfpage

CBW340

IDVL

Ic

ViIoVo

Vi

0

Vo

VL

current flow

transistor conductingtransistor cut-off

0

0

0

Ic

ID

IL

S

L IL

load

closed openswitch

Vi − Vo

Ic + ID = IL

IL(av) = Io(av)

+

−

+

−

Fig.19 Forward converter with line isolation.

handbook, halfpage

MBE964

ID1

IL

T

L

IcViID2

D2 Vo

1998 Oct 01 15

Philips Components


PUSH-PULL CONVERTER

The basic circuit of the push-pull converter, with voltageand current waveforms is shown in Fig.20.

The push-pull converter is an arrangement of two forwardconverters operating in antiphase (push-pull action). Withswitch S1 closed (Fig.20a) diode D2 conducts and energy

is simultaneously stored in the inductor and supplied to theload. With S1 and S2 open (Fig.20b), the energy stored inthe inductor continues to support the load current via theparallel diodes D1 and D2, which are now acting asflywheel diodes. When switch S2 closes (Fig.20c), diodeD1 continues to conduct, diode D2 stops conducting andthe process repeats itself.

Fig.20 Basic circuit of a DC to DC push-pull converter with associated waveforms.

handbook, halfpage

CBW348

Vi

Vo load

S1

S2

L

D1

D2

ID2

Ic1

handbook, halfpage

CBW349

Vi

Vo load

S1

S2

L

IL

D1

D2

ID2

ID1

handbook, halfpage

CBW350

Vi

Vo load

S1

S2

L

D1

D2

ID1

Ic2

handbook, halfpage

CBW351

Ic1

Ic2

ID1

ID2

IL

0

0

0

S1

switch closed

ID1+ ID2 = IL

Io

S2 S1

0

0

δTT/2

a. b.

c. d.

1998 Oct 01 16

Philips Components


A push-pull converter circuit doubles the frequency of theripple current in the output filter and, therefore, reduces theoutput ripple voltage. A further advantage of the push-pulloperation is that the transformer core is excited alternatelyin both directions in contrast to both the forward andflyback converters. Therefore, for the same operatingconditions and power throughput, a push-pull converterdesign can use a smaller transformer core.

Multiple outputs can be constructed by using severalsecondary windings, each with its own output diodes,inductor and smoothing capacitor.

CONVERTER SELECTION

In each of the three basic converter designs there areseveral different circuit possibilities. In the flyback andforward converters, single and two-transistor designs canbe used. If two transistors are used, they will switchsimultaneously. This type of circuit preference isdetermined by the allowable collector-emitter voltage andcollector current of the transistor. In push-pull converterdesigns, the primary of the transformer can be connectedin several ways (see Fig.22).

Depending upon how the transformer primary is driven,it is possible to differentiate between single-ended(see Fig.22a), push-pull (see Fig.22b) and full-bridgecircuits (see Fig.22c). Decisions on circuit details aredetermined by the transistor capabilities.

Fig.21 Conventional push-pull converter circuit.

handbook, halfpage

MBG003

Vi

Vo

TR1

TR2

L

D1 IL

ID1

ID2

D2

Fig.22 Several push-pull converter circuits.

CBW352

Vo

TR1

TR2

L

D1 IL

Ic

D2

Vo

TR1

TR2

L

D1 IL

Ic

Ic D2

Ic

Vo

TR2

TR4

L

D1 IL

Ic

Ic D2

TR1

TR3

Ic

Ic

a.

b.

c.

1998 Oct 01 17

Philips Components


For a practical converter design, the first selection thatshould be considered is the type of converter circuit to use.To aid in this initial converter circuit selection, Fig.23 offersa rough guide to the type of converter, its output voltageand power capability. This selection has to be consideredalong with other requirements, including line isolation,ripple content, overall efficiency, multiple outputs, etc.

Table 1 summarizes the most significant properties of aconverter design. It shows the relative strengths andweaknesses of the three types of converters with regard tothese characteristics.

For a high performance, high power, single output supply,where ripple is well below 1%, the push-pull design is theobvious choice. For smaller power versions of this type ofsupply, the forward, or double-forward converter providesa useful alternative to push-pull converter.

In high-voltage supplies, the flyback converter is the mostsuitable circuit and should be considered as a preference.In multiple-output supplies, the flyback converter is againnormally the first choice because it avoids the necessity ofproviding a number of output windings on the inductor,together with a single diode and capacitor for each.

Fig.23 Converter circuit selection as a function ofoutput voltage and throughput power.

handbook, halfpage

MBW413

102

103102

103

1

Po (W)101

10

Vo(V)

flyback converters

flyback &forward

converters

forwardconverters

forw

ard

& p

ush-

pull

conv

erte

rs

Table 1 Converter design selection chart (I)

Note

1. ‘+’ = favourable; ‘0’ = average; ‘−’ = unfavourable.

FUNCTIONTYPE OF CONVERTER CIRCUIT(1)

FLYBACK FORWARD PUSH-PULL

Circuitsimplicity

+ 0 −

Number ofcomponents

+ 0 −

Drivecircuitry

+ 0 −

Output ripple − 0 +

Chokevolume

not required 0 +

Transformervolume

− 0 +

Mainsisolation

+ − +

High power − 0 +

High voltage + 0 0

Multipleoutputs

+ 0 0

1998 Oct 01 18

Philips Components


CORE SELECTION

Table 2 shows which core type could be consideredsuitable for the different types of converter design.

The power-handling capability of a given core isdetermined by frequency and material grade, its geometryand available winding area, and by other factors whichdepend on the specific application.

Table 2 Converter design selection chart (II)

Note

1. ‘+’ = favourable; ‘0’ = average; ‘−’ = unfavourable.

FUNCTIONTYPE OF CONVERTER CIRCUIT(1)

FLYBACK FORWARD PUSH-PULL

E cores + + 0

Planar E cores − + 0

EFD cores − + +

ETD cores 0 + +

EC cores − 0 +

U cores + 0 0

RM cores 0 + 0

EP cores − + 0

P cores − + 0

Ring cores − + +

Operating frequency

The preferred operating frequency of a Switched ModePower Supply is greater than 20 kHz to avoid audiblenoise from the transformer. With modern power ferritesthe practical upper limit has shifted to well over 1 MHz.

Ambient temperature

Ambient temperature, together with the maximum coretemperature, determines the maximum temperature rise,which in turn fixes the permissible total power dissipationin the transformer. Normally, a maximum ambienttemperature of 60 °C has been assumed. This allowsa 40 °C temperature rise from the ambient to the centreof the transformer for a maximum core temperature of100 °C. There is a tendency however towards highertemperatures to increase power throughput.

Flux density

To avoid saturation in the cores the flux density in theminimum cross-section must not exceed the saturationflux density of the material at 100 °C. The allowable totalflux is the product of this flux density and the minimum corearea and must not be exceeded even under transientconditions, that is, when a load is suddenly applied at thepower supply output, and maximum duty factor occurstogether with maximum supply voltage. Understeady-state conditions, where maximum duty factoroccurs with minimum supply voltage, the flux is reducedfrom its absolute maximum permissible value by the ratioof the minimum to maximum supply voltage (at all highersupply voltages the voltage control loop reduces the dutyfactor and keeps the steady-state flux constant).The minimum to maximum supply voltage ratio isnormally taken as 1 : 1.72 for most applications.

1998 Oct 01 19

Philips Components


SELECTING THE CORRECT CORE TYPE

The choice of a core type for a specific design depends onthe design considerations and also on the personalpreference of the designer. Table 3 gives an overview ofcore types as a function of power throughput and this maybe useful to the designer for an initial selection.

Each of the core types has been developed for a specificapplication, therefore they all have advantages anddrawbacks depending on, for example, converter type andwinding technique.

Table 3 Power throughput for different core typesat 100 kHz switching frequency

POWER RANGE(W)

CORE TYPE

<5 RM4; P11/7; R14; EF12.6; U10

5 to 10 RM5; P14/8

10 to 20RM6; E20; P18/11; R23; U15;EFD15

20 to 50RM8; P22/13; U20; RM10;ETD29; E25; R26/10; EFD20

50 to 100ETD29; ETD34; EC35; EC41;RM12; P30/19; R26/20; EFD25

100 to 200ETD34; ETD39; ETD44; EC41;EC52; RM14; P36/22; E30; R56;U25; U30; E42; EFD30

200 to 500ETD44; ETD49; E55; EC52; E42;P42/29; U37

<500 E65; EC70; U93; U100

Choice of ferrite for power transformers

A complete range of power ferrites is available for anyapplication.

3C15

Low frequency material with improved saturation level.Suitable for flyback converters e.g. Line OutputTransformers.

3C30

Medium frequency material with improved saturation level.Suitable for flyback converters e.g. Line OutputTransformers.

3C85

Medium frequency (<200 kHz) material for industrial use.

3C90

Medium frequency (<200 kHz) material for industrial use.Lower losses than 3C85.

3C94

Medium frequency material (<200 kHz).Very low losses, especially at high flux densities.

3F3

High frequency material (up to 700 kHz).Top material for modern high frequency designs.

3F35

High frequency material (up to 1 MHz).Very low losses, around 500 kHz.

3F4

High frequency material (up to 3 MHz).Specially recommended for resonant supplies.

4F1

High frequency material (up to 10 MHz).Specially recommended for resonant supplies.

1998 Oct 01 20

Philips Components


Performance factor of power ferrites

The performance factor (f × Bmax) is a measure of thepower throughput that a ferrite core can handle at a losslevel of 300 mW/cm3. This level is considered to beacceptable for a good medium size transformer design.From the graph it is clear that for low frequencies there isnot much difference between the grades, because thecores are saturation limited. At higher frequencies, thedifferences between the grades increase. There is anoptimum operating frequency for each material grade. It isevident that in order to increase power throughput orpower density a high operating frequency and a betterferrite should be chosen.

OUTPUT CHOKES

Output chokes for Switched Mode Power Supplies have tooperate with a DC load causing a bias magnetic field HDC.

In a closed ferrite circuit, this can easily lead to saturation.Power ferrites such as 3C85 or 3F3 start saturating at fieldstrengths of about 50 A/m. Permeability drops sharply, ascan be seen in the graphs of the material data section. Thechoke loses its effectiveness.

There are two remedies against this effect:

• The use of gapped ferrite cores

• The use of a material with allow permeability and highsaturation.

Fig.24 Choke waveform.

handbook, halfpage MBG004

Iac

Iac

I0

IM

1/f

I

Fig.25 Performance factor (f × Bmax) as a function of frequency for material grades 3C85, 3F3, 3F4 and 4F1.

handbook, full pagewidth

20000

0

80000MBG005

10210110−1

40000

60000

operating freq. (MHz)

f x Bmax(Hz T)

3C85

3F3

3F4

4F1

1998 Oct 01 21

Philips Components


GAPPED CORE SETS

The effect of an air gap in the circuit is that a much higherfield strength is needed to saturate a core.

For each operating condition an optimum air gap lengthcan be found. In a design, the maximum output current (I)and the value of inductance (L) necessary to smooth theripple to the required level are known. The product I2L is ameasure of the energy which is stored in the core duringone half cycle.

Using this I2L value and the graphs given for most coretypes, the proper core and air gap can be selected quicklyat a glance. There is a choke design program on the disc“Soft Ferrite Design Tools” (9398 402 12011) for a moredetailed design, including the required number of turns.This program also covers design of inductors on openmagnetic circuits like bobbin cores and rods.

Fig.26 Effect of increased gap length.


MBW414

103 104

103

10H (A/m)

104

µ∆

10

102

102

µe=1500

µe=1000

µe=500

µe=200

µe=100

µe=50

1998 Oct 01 22

Philips Components


Fig.27 I2L graph for E cores.

handbook, halfpage

101

CBW3191

10−1

10−2

10−3

10−4

10−1

air-gap (mm)

I2L(J)

E42/21/15

E41/17/12

E36/21/15

E30&31&32&34

E30/15/7

E25/13/7

E42/20

E46/23/30

E47&50

E80/38/20

E25/6

E20/10/6

E20/10/5

E19/8/9

E16/8/5

E13/6/6

E13/7/4

E19/8/5

E65/32/27

E56/24/19

E55/28/25

E55/28/21

E71/33/32

Fig.28 I2L graph for planar E cores (valid forE + E and E + PLT combinations).

handbook, halfpage

101

CBW3201

10−1

10−2

10−3

10−4

10−1

air-gap (mm)

I2L(J)

E38/8/25

E32/6/20

E22/6/16

E18/4/10

E14/3.5/5

E43/10/28

E58/11/38

E64/10/50

1998 Oct 01 23

Philips Components


Fig.29 I2L graph for EC cores.

handbook, halfpage

101

CBW3211

10−1

10−2

10−3

10−4

10−1

air-gap (mm)

I2L(J)

EC70

EC52

EC41

EC35

Fig.30 I2L graph for EFD cores.

handbook, halfpage

101

CBW322

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

EFD25

EFD20

EFD15

EFD12

EFD10

EFD30

1998 Oct 01 24

Philips Components


Fig.31 I2L graph for EP cores.

handbook, halfpage

101

CBW323

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

EP20

EP17

EP13

EP7 & EP10

Fig.32 I2L graph for ER cores.

handbook, halfpage

101

CBW3241

10−1

10−2

10−3

10−4

10−1

air-gap (mm)

I2L(J)

ER42

ER42A

ER40

ER35

ER48 & 54 & 54S

ER28 & 28L

1998 Oct 01 25

Philips Components


Fig.33 I2L graph for ETD cores.

handbook, halfpage

101

CBW3251

10−1

10−2

10−3

10−4

10−1

air-gap (mm)

I2L(J)

ETD49

ETD44

ETD39

ETD34

ETD29

ETD54

ETD59

Fig.34 I2L graph for P cores.

handbook, halfpage

101

CBW326

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

P66/56

P42/29

P36/22

P30/19

P26/16

P22/13

P18/11

P14/8

P11/7

P9/5

P7/4

1998 Oct 01 26

Philips Components


Fig.35 I2L graph for P/I cores.

handbook, halfpage

101

CBW327

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

P26/16/I

P22/13/I

P18/11/I

P14/8/I

P11/7/I

Fig.36 I2L graph for PT cores.

handbook, halfpage

101

CBW328

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

PT26/16

PT23/11

PT18/11

PT14/8

PT30/19

1998 Oct 01 27

Philips Components


Fig.37 I2L graph for PTS cores.

handbook, halfpage

101

CBW329

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

PTS26/16

PTS23/11

PTS18/11

PTS14/8

PTS30/19

Fig.38 I2L graph for PQ cores.

handbook, halfpage

101

CBW3301

10−1

10−2

10−3

10−4

10−1

air-gap (mm)

I2L(J)

PQ35/35

PQ20/16 & 20/20

PQ32/20 & 32/30

PQ26/20 & 26/25

1998 Oct 01 28

Philips Components


Fig.39 I2L graph for RM cores.

handbook, halfpage

101

CBW331

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

RM8

RM6S&R

RM5

RM4

RM10

Fig.40 I2L graph for RM/I cores.

handbook, halfpage

101

CBW332

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

RM14/I

RM12/I

RM10/I

RM8/I

RM6S/I

RM5/I

RM4/I

RM7/I

1998 Oct 01 29

Philips Components


Fig.41 I2L graph for RM/ILP cores.

handbook, halfpage

101

CBW333

10−2

10−1

10−3

10−4

10−5

10−1

air-gap (mm)

I2L(J)

RM14/ILP

RM12/ILP

RM10/ILP

RM8/ILP

RM6S/ILP

RM5/ILP

RM4/ILP

RM7/ILP

Fig.42 I2L graph for U cores.

handbook, halfpage

101

CBW3341

10−1

10−2

10−3

10−4

10−1

spacer thickness (mm)

I2L(J)

U10/8/3

U93/76/16

U67/27/14

U33/22/9

U30/25/16

U25/20/13

U25/16/6

U20/16/7

U15/11/6

U100/25

U93/30

1998 Oct 01 30

Philips Components


IRON POWDER RING CORES

Ring cores made from compressed iron powder have arather low permeability (max. 90) combined with a veryhigh saturation level (up to 1500 mT). The permeability isso low because the isolating coating on the iron particlesacts as a so called distributed air gap. Therefore, our2P ring core range can operate under bias fields of upto 2000 A/m.

INPUT FILTERS (COMMON MODE CHOKES)

Fig.43 Common mode choke.

handbook, halfpage

MBW416

I

-I

H

-H

To avoid the conduction of switching noise from a SMPSinto the mains, an input filter is generally necessary.The magnetic circuit in these filters is usually a pair ofU cores or a ring core.

Since the noise signal is mainly common mode, currentcompensation can be used to avoid saturation.

Two separate windings on the core cause opposingmagnetic fields when the load current passes throughthem (current compensation). The common mode noisesignal however, is blocked by the full inductance causedby the high permeability ferrite.

If, for some reason, current compensation is not completeor impossible, high permeability grades will saturate.In that case one of the power grades may be a bettercompromise. Another important factor in the designprocess is the frequency range of the interference signal.High permeability ferrites have a limited bandwidth as canbe seen from Fig.44.

These materials only perform well as an inductor below thefrequency where ferromagnetic resonance occurs. Abovethis cut-off frequency, a coil will have a highly resistivecharacter and the Q-factor of the LC filter circuit will belimited an thus, also the impedance. A better result couldhave been obtained with a grade having a lowerpermeability. Figure 45 provides a quick method ofchoosing the right ferrite for the job. A design program fora complete current-compensated input filter is provided onour “Soft Ferrite Design Tools” disc (9398 402 12011).

1998 Oct 01 31

Philips Components


Fig.44 Permeability as a function of frequency of different materials.


MBG006

10210110−1

104

10

103

102

f (MHz)

µi

3C85

3C11

3E25

4A11

4C65

3E5

3F3

Fig.45 Selection chart for materials used in input filters.

dbook, full pagewidth

<500 kHz

500 kHz to 3 MHz

3 MHz to 30 MHz

>30 MHz

no load current

with load current

3C11 - 3E25 - 3E5

3C85 - 3F3

3C85 - 3F3

4A11

4C65

INTERFERENCE FILTER

CBW354

1998 Oct 01 32

Philips Components


MAGNETIC REGULATORS

Saturable inductors provide a means of efficientlyregulating several independent outputs in a SMPS byblocking varying amounts of energy from the secondary ofthe transformer. This eliminates the need for feedbackbetween secondary and primary and allows improvedisolation of input and output. The circuits required are bothsimple and economic and can be easily integrated.

A schematic of a saturable inductor circuit (withoutregulation) together with associated waveforms isshown in Fig.46.

handbook, halfpage

CBW341

V

RV1SW1

100 kHz

SW1ON

SW1OFF

V/R

V1

IR

IR

IR

A

Aincreasing

H

B

B

B

C

C

D

D

−Bsat

+Bsat

saturable inductor

Fig.46 Schematic of a saturable inductor andassociated waveforms (without regulation).

TYPICAL CYCLE CHARACTERISTICS

During a typical cycle:

• Switch SW1 is closed (point A on timing diagram), theinductance of the saturable inductor limits the rate ofcurrent rise until the core becomes saturated.

• With the core saturated (point B), the only impedance tocurrent flow is the very small resistance of the inductor,which can be regarded as a short circuit with powerbeing transferred unimpeded to the load resistor.

• Switch SW1 is opened (point C). Because the saturableinductor has a rectangular B-H loop, the flux remainsunchanged even when H has fallen to zero. Since therehas been no change in flux, there is no inductance andthe current can fall instantaneously.

• Switch SW is re-closed (point D). As the flux in the coreis still saturated and remains unchanged, there is noresistance to the current flow to the load.

A schematic of a regulated circuit and its associatedwaveforms is shown in Fig.47.

In this circuit, the inductor is saturated while switch SW1 isclosed, thus reducing the period during which energy isconducted from the transformer to the load. Varying thelevel of this control current modulates the main outputvoltage waveform (see Fig.48), thus regulating the outputvoltage across the load.

The 3R1 ferrite material is an excellent alternative toamorphous metal for the cores of saturable inductorsof SMPS.

Remark:

The performance of 3R1 is comparable to that ofamorphous metal making it an excellent material forapplications such as output regulation and spikesuppression. When 3R1 ring cores are driven exactly attheir natural mechanical resonant frequencies amagneto-elastic resonance will occur.

With large flux excursions and no mechanical damping,amplitudes can become so high that the maximum tensilestress of the ferrite is exceeded. Cracks or even breakageof the ring core could be the result. It is advised not to drivethe toroidal cores at their radial resonant frequencies oreven subharmonics (e.g. half this resonant frequency).

Resonant frequencies can be calculated for any ring corewith the following formula:

where:

f = radial resonant frequency (kHz)

Do = outside diameter (mm)

Di = inside diameter (mm).

fr5700

π Do Di+

2-------------------

----------------------------- kHz=

1998 Oct 01 33

Philips Components


Fig.47 Schematic of a saturable inductor and associated waveforms (with regulation).


CBW353

V

RSW1

SW2VR1

100 kHz

ON

OFF

ON

OFF

IR

Ic

SW1

SW2

IR

Ic

increasing Ic100 mA (typ.)

10 A (typ.)

Fig.48 Typical control curve for a 3R1 ring core(size 14 × 9 × 5 mm, with 15 turns).

handbook, halfpage

0 50 250

8

6

2

0

4

MBG009

100 150 200

outputvoltage

(V)

control current (mA)

Fig.49 Properties of 3R1 ferrite material;f = 100 kHz, T = 25 °C.

handbook, halfpage

−400 −200 0 400

400

200

−200

−400

0

MBG010

200

magneticinduction

(mT)

magnetic field strength (A/m)

1998 Oct 01 34

Philips Components


Ferrites for Interference Suppression and Electromagnetic Compatibility (EMC)

Fig.50 Principles of Electromagnetic Compatibility (EMC).

andbook, full pagewidth

dB

receiver

source

EMC margin

safety marginsupplier

EMS limit

EME limit

EMS level

EME level

frequency

EMC = Electro Magnetic CompatabilityEMS = Electro Magnetic SusceptabilityEME = Electro Magnetic Emission

MBW418

With the ever increasing intensive use of electronicequipment Electromagnetic Compatibility (EMC) hasbecome an important item. Laws specify limits of the levelof interference caused by equipment (EME) and also thesensitivity of equipment to incoming interference (EMS).

Limiting curves are defined by organizations such asCISPR and FCC. Since the density of equipmentincreases, laws will become more stringent in the nearfuture.

During the design phase, problems with interference canbe avoided to some extent. Often additional suppressioncomponents such as capacitors and coils will benecessary to meet the required levels. Inductivecomponents are very effective in blocking interferingsignals, especially at high frequencies. The principles ofsuppression are shown in Fig.51.

Capacitors are used as a shunt impedance for theunwanted signal.

Unfortunately for high frequencies, most capacitors do nothave the low impedance one might expect because ofparasitic inductance or resistance.

Fig.51 Basic suppression circuits.

handbook, halfpage

U

Z

Z

Z

I s

L

i

i

U

Z

Z

Z

I s

L

i

i Z p

pI

interference source

interference source

I

I L

MBW400

1998 Oct 01 35

Philips Components


Inductors are used in series with the load impedance. Theyprovide a low impedance for the wanted signal, but a highimpedance for the interfering, unwanted, signal.

Philips have a full range of ring cores, beads, beads onwire, SMD beads, wideband chokes and cable shields tosuit every application. Rods and tubes are also often usedfor this application after they have been coiled by the user.

SAMPLE BOXES

As the design process in these areas is often based ontrial and error, we have assembled 6 differentdesigners’ sample boxes . Each box is filled with aselection from our standard ranges, which aims at aspecific application area. The boxes also contain a bookletwith full information about the products and theirapplications. These sample boxes are:

• Sample box 9: SMD beads and chokes

• Sample box 10: Cable shielding

• Sample box 11: EMI suppression products.

INTERFERENCE SUPPRESSION BEADS

A range of beads is available in two material grades,especially developed for suppression purposes.

They can easily be shifted on existing wires in theequipment:

• 3S1 for frequencies up to 30 MHz

• 4S2 for frequencies from 10 to 1000 MHz.

The material grades and beads are fully guaranteed fortheir main feature, impedance as a function of frequency.

The grade 3S1 has a high permeability and is thereforerather sensitive for DC load. In applications where a highDC current is flowing 4S2 can be a better choice(see Figs 52, 53 and 54).

Fig.52 Impedance as a function of frequencyfor material grades 3S1 and 4S2;bead size 5 × 2 × 10 mm.

handbook, halfpage150

01

MBW419

10 102 103

30

60

90

120

f (MHz)

3S1

4S2Z

(Ω)

BD5/2/10

1998 Oct 01 36

Philips Components


Fig.53 Impedance as a function of frequency at different DC levels for material grade 4S2.


0

150MBW420

103102101

30

60

90

120

ZS(Ω)

frequency (MHz)

1 A

3 A

no DC100 mA300 mA

Fig.54 Impedance as a function of frequency at different DC levels for material grade 3S1.


0

100MBG011

103102101

20

40

60

80

ZS(Ω)

frequency (MHz)

1 A

2 A

no DC100 mA300 mA

1998 Oct 01 37

Philips Components


BEADS ON WIRE

This product range consists of suppression beads, alreadymounted on pre-soldered 0.6 mm wire and taped onstandard reels. These can be handled by automaticplacement machines.

SMD FERRITE BEADS

In response to market demands for smaller, lighter andmore integrated electronic devices a series of SMD beadswas added to our range. They are available in differentsizes and 2 suppression ferrite grades.

Basically these beads consist of a ferrite tube with arectangular cross-section and a flat tinned copper wirewhich is bent around the edges and forms the terminals ofthe component. This design offers many superiormechanical and electrical features.

Some examples of their impedance as a function offrequency and the influence of bias current are given in thegraphs.

Fig.55 Outline of SMD beads.

handbook, halfpage

MSB618

Fig.56 Impedance as a function offrequency for SMD beads.


01

MBW346

10 100 1000

10

20

30

40

f (MHz)

3S1

4S2Z (Ω)

BDS3/1.8/5.3

Fig.57 Impedance as a function of frequency for anSMD bead with bias current as a parameter.


01

MBW347

10 100 1000

10

20

30

40

f (MHz)

0 AZ

(Ω) 1 A2 A3 A4 A5 A

BDS3/1.8/5.3-4S2

1998 Oct 01 38

Philips Components


SMD FERRITE BEADS FOR

COMMON-MODE INTERFERENCE SUPPRESSION

Philips Components has introduced a new range of softferrite SMD beads for common-mode interferencesuppression.

With standard suppression methods in a signal path, thewanted signal is often suppressed along with theinterference, and in many modern applications (EDP forinstance) this leads to unacceptable loss of signal.

In Philips' new interference suppression beads, a pair ofconductors within a single soft ferrite block are connectedalong their lengths by an air gap.

Common-mode signals (interference signals passing inthe same direction along the input and output channels ofa device, an IC for instance) serve to reinforce themagnetic flux around both conductors and are thereforeattenuated.

In contrast, the wanted signal passing along the input andoutput channels serves to cancel the flux around theconductors and therefore passes unattenuated. Fig.58 Outline of an SMD common-mode choke.

handbook, halfpage

MBW358

Fig.59 Impedance as a function offrequency of an SMD commonmode bead with two conductors.


01

MBW348

10 100 1000

40

80

120

160

f (MHz)

1 turn

Z (Ω)

2 turns

CMS2-5.6/3/4.8-4S2

Fig.60 Impedance as a function offrequency of an SMD commonmode bead with four conductors.


01

MBW349

10 100 1000

40

80

120

160

f (MHz)

1 turn

Z (Ω)

2 turns

inner channel

outer channel

CMS4-11/3/4.8-4S2

1998 Oct 01 39

Philips Components


WIDEBAND CHOKES

Wideband chokes are wired multi-hole beads. Sincethey have up to 21⁄2 turns of wire their impedance valuesare rather high over a broad frequency range, hencetheir name.

The magnetic circuit is closed so there is little stray field.The DC resistance is very low since only a short length of0.6 mm copper wire is used.

These products already have a long service record and arestill popular for various applications.

Recently the range was extended with several new types,e.g. with isolation and taped on reel.

Fig.61 Outline of wideband chokes.


MSB614

Fig.62 Impedance as a function offrequency for a wideband choke.


01

MBW421

10 102 103

200

400

600

800

f (MHz)

3B1

Z (Ω)

4B1

WBC2.5

Fig.63 Outline of a wideband chokewith support plate.

handbook, halfpage

MSB617

1998 Oct 01 40

Philips Components


SMD WIDEBAND CHOKES

SMD wideband chokes are an alternative to a SMD beadwhen more impedance or damping is required.

The design of this product is based on our well knownrange of wideband chokes.

In these products the conductor wire is wound throughholes in a multi-hole ferrite core, thus separating themphysically and reducing coil capacitance.

The result is a high impedance over a wide frequencyrange, a welcome feature for many interference problems.

The present SMD design preserves the excellentproperties and reliability of the original wideband chokesby keeping the number of electrical interfaces to anabsolute minimum.

Fig.64 Outline of an SMD wideband choke.

handbook, halfpage

MBW359

Fig.65 Impedance as a function of frequencyfor SMD wideband chokes.


01

MBW350

10 100 1000

200

400

600

800

f (MHz)

3B1

Z (Ω)

4B1

WBS2.5-5/4.8/10

Fig.66 Insertion loss of a 3B1 SMD wideband chokeas a function of frequency (50 Ω circuit).

handbook, halfpage0

−251

MBW351

10 100 1000

−20

−15

−10

−5

f (MHz)

IL (dB)

0 A

1 A2 A

3 A

WBS2.5-5/4.8/10-3B1

1998 Oct 01 41

Philips Components


CABLE SHIELDS

New in our range are so-called cable shields. Theseproducts are an effective remedy against common-modeinterference on coaxial or flat cables. They come in severalshapes: round tubes, rectangular sleeves and split sleevesto mount on existing cable connections.

Our new suppression material 3S4 is very suitable for thisapplication. It combines a high permeability (1700) for highimpedance in the lower frequency range with an excellenthigh frequency behaviour for true wideband suppression.

Fig.67 Outline of a cable shield.

handbook, halfpage

MBW360

Fig.68 Impedance of a cable shieldas a function of frequency.


01

MBW361

10 102 103

50

100

150

200

f (MHz)

Z (Ω)

CSF38/12/25

1998 Oct 01 42

Philips Components


RODS AND TUBES

Rods and tubes are generally used to increase theinductance of a coil. The magnetic circuit is very open andtherefore the mechanical dimensions have more influenceon the inductance than the ferrite's permeability(see Fig.69) unless the rod is very slender.

In order to establish the effect of a rod on the inductanceof a coil, the following procedure should be carried out:

• Calculate the length to diameter ratio of the rod (l/d)

• Find this value on the horizontal axis and draw avertical line.

The intersection of this line with the curve of the materialpermeability gives the effective rod permeability.

The inductance of the coil, provided the winding covers thewhole length of the rod is given by:

where:

N = number of turns

A = cross sectional area of rod

I = length of coil.

L µ0µrodN

2A

l----------- H( )=

Fig.69 Rod permeability (µrod) as a function of length to diameter ratio with material permeability as a parameter.


103

102

10

21 10 102

Length / diameter ratio

µrod

µi = 10.000

5000

2000

1000

700500

400

300

200

150

100

70

40

20

10

MBW422

Philips Components – a worldwide companyAustralia: Philips Components Pty Ltd., NORTH RYDE,

Tel. +61 2 9805 4455, Fax. +61 2 9805 4466

Austria: Österreichische Philips Industrie GmbH, WIEN,Tel. +43 1 60 101 12 41, Fax. +43 1 60 101 12 11

Belarus: Philips Office Belarus, MINSK,Tel. +375 172 200 924/733, Fax. +375 172 200 773

Benelux: Philips Nederland B.V., EINDHOVEN, NL,Tel. +31 40 2783 749, Fax. +31 40 2788 399

Brazil: Philips Components, SÃO PAULO,Tel. +55 11 821 2333, Fax. +55 11 829 1849

Canada: Philips Electronics Ltd., SCARBOROUGH,Tel. +1 416 292 5161, Fax. +1 416 754 6248

China: Philips Company, SHANGHAI,Tel. +86 21 6354 1088, Fax. +86 21 6354 1060

Denmark: Philips Components A/S, COPENHAGEN S,Tel. +45 32 883 333, Fax. +45 31 571 949

Finland: Philips Components, ESPOO,Tel. +358 9 615 800, Fax. +358 9 615 80510

France: Philips Composants, SURESNES,Tel. +33 1 4099 6161, Fax. +33 1 4099 6493

Germany: Philips Components GmbH, HAMBURG,Tel. +49 40 2489-0, Fax. +49 40 2489 1400

Greece: Philips Hellas S.A., TAVROS,Tel. +30 1 4894 339/+30 1 4894 239, Fax. +30 1 4814 240

Hong Kong: Philips Hong Kong, KOWLOON,Tel. +852 2784 3000, Fax. +852 2784 3003

India: Philips India Ltd., MUMBAI,Tel. +91 22 4930 311, Fax. +91 22 4930 966/4950 304

Indonesia: P.T. Philips Development Corp., JAKARTA,Tel. +62 21 794 0040, Fax. +62 21 794 0080

Ireland: Philips Electronics (Ireland) Ltd., DUBLIN,Tel. +353 1 7640 203, Fax. +353 1 7640 210

Israel: Rapac Electronics Ltd., TEL AVIV,Tel. +972 3 6450 444, Fax. +972 3 6491 007

Italy: Philips Components S.r.l., MILANO,Tel. +39 2 6752 2531, Fax. +39 2 6752 2557

Japan: Philips Japan Ltd., TOKYO,Tel. +81 3 3740 5135, Fax. +81 3 3740 5035

Korea (Republic of): Philips Electronics (Korea) Ltd., SEOUL,Tel. +82 2 709 1472, Fax. +82 2 709 1480

Malaysia: Philips Malaysia SDN Berhad,Components Division, PULAU PINANG,Tel. +60 3 750 5213, Fax. +60 3 757 4880

Mexico: Philips Components, EL PASO, U.S.A.,Tel. +52 915 772 4020, Fax. +52 915 772 4332

New Zealand: Philips New Zealand Ltd., AUCKLAND,Tel. +64 9 815 4000, Fax. +64 9 849 7811

Norway: Norsk A/S Philips, OSLO,Tel. +47 22 74 8000, Fax. +47 22 74 8341

Pakistan: Philips Electrical Industries of Pakistan Ltd., KARACHI,Tel. +92 21 587 4641-49, Fax. +92 21 577 035/+92 21 587 4546

Philippines: Philips Semiconductors Philippines Inc.,METRO MANILA, Tel. +63 2 816 6345, Fax. +63 2 817 3474

Poland: Philips Poland Sp. z.o.o., WARSZAWA,Tel. +48 22 612 2594, Fax. +48 22 612 2327

Portugal: Philips Portuguesa S.A.,Philips Components: LINDA-A-VELHA,Tel. +351 1 416 3160/416 3333, Fax. +351 1 416 3174/416 3366

Russia: Philips Russia, MOSCOW,Tel. +7 95 755 6918, Fax. +7 95 755 6919

Singapore: Philips Singapore Pte Ltd., SINGAPORE,Tel. +65 350 2000, Fax. +65 355 1758

South Africa: S.A. Philips Pty Ltd., JOHANNESBURG,Tel. +27 11 470 5911, Fax. +27 11 470 5494

Spain: Philips Components, BARCELONA,Tel. +34 93 301 63 12, Fax. +34 93 301 42 43

Sweden: Philips Components AB, STOCKHOLM,Tel. +46 8 5985 2000, Fax. +46 8 5985 2745

Switzerland: Philips Components AG, ZÜRICH,Tel. +41 1 488 22 11, Fax. +41 1 481 7730

Taiwan: Philips Taiwan Ltd., TAIPEI,Tel. +886 2 2134 2900, Fax. +886 2 2134 2929

Thailand: Philips Electronics (Thailand) Ltd., BANGKOK,Tel. +66 2 745 4090, Fax. +66 2 398 0793

Turkey: Türk Philips Ticaret A.S., GÜLTEPE/ISTANBUL,Tel. +90 212 279 2770, Fax. +90 212 282 6707

United Kingdom: Philips Components Ltd., DORKING,Tel. +44 1306 512 000, Fax. +44 1306 512 345

United States:

• Display Components, ANN ARBOR, MI,Tel. +1 734 996 9400, Fax. +1 734 761 2776

• Magnetic Products, SAUGERTIES, NY,Tel. +1 914 246 2811, Fax. +1 914 246 0487

• Passive Components, SAN JOSE, CA,Tel. +1 408 570 5600, Fax. +1 408 570 5700

Yugoslavia (Federal Republic of): Philips Components, BELGRADE,Tel. +381 11 625344/373, Fax. +381 11 635 777

Internet :

• Display Components: www.dc.comp.philips.com• Passive Components: www.passives.comp.philips.com

For all other countries apply to:Philips Components, Building BF-1, P.O. Box 218, 5600 MD EINDHOVEN,The Netherlands, Fax. +31-40-27 23 903.

COD19 © Philips Electronics N.V. 1998

All rights are reserved. Reproduction in whole or in part is prohibited without theprior written consent of the copyright owner.The information presented in this document does not form part of any quotation orcontract, is believed to be accurate and reliable and may be changed without notice.No liability will be accepted by the publisher for any consequence of its use. Publicationthereof does not convey nor imply any license under patent- or other industrial orintellectual property rights.

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DATA SHEET - Thierry LEQUEU · 2009. 3. 12. · or cylinder, manufactured from ferrite or...

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