ANTENAS Y PROPAGACION

JANUARY 2010 VOLUME 58 NUMBER 1 IETPAK (ISSN 0018-926X)

Editorial . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . T. S. Bird 2

PAPERS

AntennasA Dual-Linearly-Polarized MEMS-Reconfigurable Antenna for Narrowband MIMO Communication Systems . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. Grau, J. Romeu, M.-J. Lee, S. Blanch, L. Jofre, and F. De Flaviis 4A Lumped Circuit for Wideband Impedance Matching of a Non-Resonant, Short Dipole or Monopole Antenna . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . V. Iyer, S. N. Makarov, D. D. Harty, F. Nekoogar, and R. Ludwig 18A Simple Ultrawideband Planar Rectangular Printed Antenna With Band Dispensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . K. G. Thomas and M. Sreenivasan 27Novel Broadband Circularly Polarized Cavity-Backed Aperture Antenna With Traveling Wave Excitation . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . K.-F. Hung and Y.-C. Lin 35Frequency Selective Surfaces for Extended Bandwidth Backing Reflector Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Pasian, S. Monni, A. Neto, M. Ettorre, and G. Gerini 43ArraysAntenna Modeling Based on a Multiple Spherical Wave Expansion Method: Application to an Antenna Array . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Serhir, P. Besnier, and M. Drissi 51An External Calibration Scheme for DBF Antenna Arrays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . H. Pawlak and A. F. Jacob 59Study and Design of a Differentially-Fed Tapered Slot Antenna Array . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . E. de Lera Acedo, E. Garca, V. Gonzlez-Posadas, J. L. Vzquez-Roy, R. Maaskant, and D. Segovia 68Metallic Wire Array as Low-Effective Index of Refraction Medium for Directive Antenna Application . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . R. Zhou, H. Zhang, and H. Xin 79ElectromagneticsA Novel Analysis of Microstrip Structures Using the Gaussian Greens Function Method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. M. Tajdini and A. A. Shishegar 88Application of Kummers Transformation to the Efficient Computation of the 3-D Greens Function With 1-D

Periodicity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. L. Fructos, R. R. Boix, and F. Mesa 95

(Contents Continued on p. 1)

(Contents Continued from Front Cover)

Electromagnetic Scattering by an Infinite Elliptic Dielectric Cylinder With Small Eccentricity Using PerturbativeAnalysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. D. Tsogkas, J. A. Roumeliotis, and S. P. Savaidis 107

Numerical MethodsImproved Model-Based Parameter Estimation Approach for Accelerated Periodic Method of Moments Solutions With

Application to the Analysis of Convoluted Frequency Selected Surfaces and Metamaterials . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . X. Wang and D. H. Werner 122

Pareto Optimal Microwave Filter Design Using Multiobjective Differential Evolution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. K. Goudos and J. N. Sahalos 132

Scattering and ImagingA Sparsity Regularization Approach to the Electromagnetic Inverse Scattering Problem .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. W. Winters, B. D. Van Veen, and S. C. Hagness 145Adaptive CLEAN With Target Refocusing for Through-Wall Image Improvement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P. C. Chang, R. J. Burkholder, and J. L. Volakis 155WirelessA Comprehensive Channel Model for UWB Multisensor Multiantenna Body Area Networks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. van Roy, C. Oestges, F. Horlin, and P. De Doncker 163Path-Loss Characteristics of Urban Wireless Channels . . . . K. T. Herring, J. W. Holloway, D. H. Staelin, and D. W. Bliss 171A MIMO Propagation Channel Model in a Random Medium .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. Ishimaru, S. Jaruwatanadilok, J. A. Ritcey, and Y. Kuga 178Effect of Optical Loss and Antenna Separation in 2 2 MIMO Fiber-Radio Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. Kobyakov, M. Sauer, A. Ngoma, and J. H. Winters 187Experimental Evaluation of MIMO Capacity and Correlation for Narrowband Body-Centric Wireless Channels . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . I. Khan and P. S. Hall 195

COMMUNICATIONS

Application of Characteristic Modes and Non-Foster Multiport Loading to the Design of Broadband Antennas . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . K. A. Obeidat, B. D. Raines, and R. G. Rojas 203

77 GHz Stepped Lens With Sectorial Radiation Pattern as Primary Feed of a Lens Based CATR .. . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . M. Multari, J. Lanteri, J. L. Le Sonn, L. Brochier, C. Pichot, C. Migliaccio, J. L. Desvilles, and P. Feil 207

A Single-Layer Ultrawideband Microstrip Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Q. Wu, R. Jin, and J. Geng 211Influence of the Finite Slot Thickness on RLSA Antenna Design . . . . . . . . . . . . . . . . . . A. Mazzinghi, A. Freni, and M. Albani 215Efficient Determination of the Poles and Residues of Spectral Domain Multilayered Greens Functions That are Relevant

in Far-Field Calculations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. L. Fructos, R. R. Boix, R. Rodrguez-Berral, and F. Mesa 218Electromagnetic Scattering From a Slotted Conducting Wedge . . . . . . . . . . . . . . . . . . . . J. J. Kim, H. J. Eom, and K. C. Hwang 222The Optimal Spatially-Smoothed Source Patterns for the Pseudospectral Time-Domain Method . . . . . . . . . . . . . . . . . . . . Z. Lin 227Experimental Microwave Validation of Level Set Reconstruction Algorithm .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. A. Woten, M. R. Hajihashemi, A. M. Hassan, and M. El-Shenawee 230Scattering of Electromagnetic Waves From a Rectangular Plate Using an Enhanced Stationary Phase Method

Approximation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . C. G. Moschovitis, K. T. Karakatselos, E. G. Papkelis, H. T. Anastassiu, I. C. Ouranos, A. Tzoulis, and P. V. Frangos 233

Experimental Characterization of UWB On-Body Radio Channel in Indoor Environment Considering DifferentAntennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. Sani, A. Alomainy, G. Palikaras, Y. Nechayev, Y. Hao, C. Parini, and P. S. Hall 238

CORRECTIONS

Corrections to Modeling Antenna Noise Temperature Due to Rain Clouds at Microwave and Millimeter-WaveFrequencies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . F. S. Marzano 242

List of Reviewers for 2009 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 243

IEEE ANTENNAS AND PROPAGATION SOCIETYAll members of the IEEE are eligible for membership in the Antennas and Propagation Society and will receive on-line access to this TRANSACTIONS through IEEE Xplore upon payment of the annual Societymembership fee of $24.00. Print subscriptions to this TRANSACTIONS are available to Society members for an additional fee of $36.00. For information on joining, write to the IEEE at the address below.Member copies of Transactions/Journals are for personal use only.

ADMINISTRATIVE COMMITTEEM. ANDO, President R. D. NEVELS, President Elect M. W. SHIELDS, Secretary-Treasurer

2010 2011 2012 2013P. DE MAAGTG. ELEFTHERIADESG. MANARAP. PATHAK*A. F. PETERSON

A. AKYURTLUW. A. DAVISH. LINGM. OKONIEWSKI

*J. T. BERNHARD

Honorary Life Members: R. C. HANSEN, W. R. STONE*Past President

Committee Chairs and RepresentativesAntenna Measurements (AMTA): S. SCHNEIDERAntennas & Wireless Propagation Letters Editor-in-Chief:

G. LAZZIApplied Computational EM Society (ACES): A. F. PETERSONAwards: A. F. PETERSONAwards and Fellows: C. A. BALANISChapter Activities: L. C. KEMPELCCIR: P. MCKENNACommittee on Man and Radiation: G. LAZZIConstitution and Bylaws: O. KILICDigital Archive Editor-in-Chief: A. Q. MARTINDistinguished Lecturers: J. C. VARDAXOGLOUEducation: D. F. KELLYEAB Continuing Education: S. R. RENGARAJANElectronic Design Automation Council: M. VOUVAKISElectronic Publications Editor-in-Chief: S. R. BESTEuropean Representatives: B. ARBESSER-RASTBURGFellows Nominations Committee: J. L. VOLAKISFinance: M. W. SHIELDS

Gold Representative: R. ADAMSHistorian: K. D. STEPHANIEEE Press Liaison: R. J. MAILLOUXIEEE Magazine Committee: W. R. STONEIEEE Public Relations Representative: W. R. STONEIEEE Social Implications of Technology: R. L. HAUPTInstitutional Listings: T. S. BIRDJoint Committee on High-Power Electromagnetics:

C. E. BAUMLong-Range Planning: C. RHOADSMagazine Editor-in-Chief: W. R. STONEMeetings Coordination: S. A. LONGMeetings Joint AP-S/URSI: M. A. JENSENMembership: S. BALASUBRAMANIAMNano Technology Council: G. W. HANSONNew Technology Directions: S. C. HAGNESSNominations: J. T. BERNHARD

PACE: J. M. JOHNSONPublications: R. J. MARHEFKARAB/TAB Transnational Committee Liaison: D. R. JACKSONRegion 10 Representative: H. NAKANOSensor Council: A. I. ZAGHOUL, T. S. BIRD, M. W. SHIELDSStandards CommitteeAntennas: M. H. FRANCISStandards CommitteePropagation: D. V. THIELTABARC Correspondent: C. A. BALANISTAB Magazines Committee: W. R. STONETAB New Technology Directions Committee: A. I. ZAGHLOULTAB Public Relations Committee: W. R. STONETAB Transactions Committee: T. S. BIRDTransactions Editor-in-Chief: T. S. BIRDTransnational Committee: D. R. JACKSONUSAB Committee on Information Policy: S. WEINUSAB R&D Committee: A. C. SCHELLUSNC/URSI : J. T. BERNHARDTWomen in Engineering Representative: P. F. WAHID

AP Transactions website: http://www.ict.csiro.au/aps AP Transactions Manuscript Central website: http://tap-ieee.manuscriptcentral.comChapter Chairs

Albuquerque: L. H. BOWENAtlanta: D. M. KOKOTOFFAustralian Capital Territory: E. S. LENSSONBeijing: D. ZUBeijing, Nanjing: W. X. ZHANGBenelux: D. VANHOENACKERBombay: M. B. PATILBuffalo: M. R. GILLETTEBulgaria: K. K. ASPARUHOVACalcutta: P. K. SAHACentral New England: B. T. PERRYChicago: F. ARYANFARCoastal Los Angeles: F. J. VILLEGASColumbus: M. A. CARRConnecticut: C. ALVAREZCroatia: Z. SIPUSCzechoslovakia: M. POLIVKADayton: A. J. TERZUOLIEastern North Carolina: T. W. NICHOLS

Eastern North Carolina: T. W. NICHOLSEgypt: H. M. EL-HENNAWYFinland: A. LUUKANENFlorida West Coast: K. A. OCONNORFoothill: F. G. FREYNEFort Worth: S. TJUATJAFrance: M. DRISSIFukuoda: M. TAGUCHIHong Kong: K. W. LEUNGHouston: J. T. WILLIAMSHuntsville: H. SCHANTZIsrael: S. AUSTERItaly: G. VECCHIJapan: N. MICHISHITAKansai: M. KOMINAMIMalaysia: E. MAZLINAMelbourne: A. M. JONESMontreal: K. WUNagoya: N. INAGAKI

New South Wales: K. P. ESSELLENorth Jersey: H. DAYALOrlando: P. F. WAHIDOttawa: Q. YEPhiladelphia: J. NACHAMKINPoland: W. KRZYSZTOFIKPortugal: C. A. FERNANDESQueensland: D. V. THIELRio de Janeiro: J. R. BERGMANNRussia, Moscow: D. M. SAZONOVRussia, Nizhny: Y. I. BELOVSt. Louis: D. MACKESan Diego: G. J. TWOMEYSanta Clara Valley/San Francisco: G. A. MANASSEROSeoul: H. J. EOMSouth Africa: P. W. VAN der WALTSouth Australia: B. D. BATESSouthern Alberta: S. BOUMAIZA

Spain: L. J. DE-HAROSpringfield: P. R. SIQUEIRASweden: A. RYDBERGSwitzerland: C. O. FUMEAUXSyracuse: E. ARVASTainan: W.-S. CHENTaipei: J-F. KIANGThailand: P. AKKARAEKTHALINToronto: G. V. ELEFTHERIADESTurkey: I. TEKINUkraine, East: A. I. NOSICHUkraine, Kharkov: N. N. KOLCHIGINUkraine, West: O. ZAMORSKAVancouver: D. G. MICHELSONVictoria: K. GHORBANIWinnipeg: V. I. OKHMATOVSKI

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATIONIs the leading international engineering journal on the general topics of electromagnetics, antennas and wave propagation. The journal is devoted to antennas, including analysis, design, development, measure-ment, and testing; radiation, propagation, and the interaction of electromagnetic waves with discrete and continuous media; and applications and systems pertinent to antennas, propagation, and sensing, suchas applied optics, millimeter- and sub-millimeter-wave techniques, antenna signal processing and control, radio astronomy, and propagation and radiation aspects of terrestrial and space-based communication,including wireless, mobile, satellite, and telecommunications. Author contributions of relevant full length papers and shorter Communications are welcomed.See inside back cover for Editorial Board.

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Digital Object Identifier 10.1109/TAP.2009.2039676

2 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 1, JANUARY 2010

EditorialReflections on 2009

R ESEARCHERS and publishers alike eagerly await the re-lease of citation data. Irrespective of the significant lim-itations of these data, they are now an integral part of appli-cations for grants, fellowships and promotions as well as indi-vidual CVs. For publishers and authors, citations equal impact.The IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATIONsImpact Factor has risen more than 160% over the past five years.For the first time, the IEEE TRANSACTIONS is ranked among thetop ten electrical engineering publications, based on the latestdata from Thomson ISI. These data show that the journal Im-pact Factor in 2008 was 2.479, compared with 1.636 in 2007.While this is pleasing, more significant is the 53% increase inthe total number of citations for the Transactions from 10 375 to15 884. The total number of papers published (about 476) hasremained roughly the same for the past three years.

Significantpublicationmilestoneswerepassedin2009withoutthe world coming to an end. Pre-processing of manuscripts wentpaperless in May when the IEEE TRANSACTIONS migrated to anon-line system.Thefirst issue tobecompletelyassembledon-linewas last November, Volume 57, No. 11. This transition was notwithout its hiccups, as some of you may have noticed. The changehas been likened to a pilot migrating from a small plane to ajumbo jet without simulator training! Whilst most changes oc-curred smoothly, others produced unexpected results: some au-thors received a publication date by email without the EditorialOffice being aware that a paper had been scheduled and paperswere queued for publication without the usual pre-processing.The publication of some papers has been delayed, but we shouldpick this up with some large issues early in 2010.

An important benefit of on-line processing is that paperscleared for publication are now available on IEEE Xplore aboutthree months ahead of the printed Transactions (click viewarticles). This has required IEEE to formally specify the dateof publication, which it recently determined will be the date anarticle is publicly available, either as an on-line pre-print viaIEEE Xplore or in print, whichever comes first. Further, by Jan-uary 1, 2011, articles in all IEEE TRANSACTIONS, journals, andletters will include the following dates in the footnotes: man-uscript received, manuscript revised, manuscript acceptance,publication and current version. The IEEE TRANSACTIONScommenced doing this last September.

The topic of pre-publications leads to a related matter. Someauthors of papers under review have been requested to eitherdisplay a notice indicating the paper is subject to IEEE copy-right or to remove it from the web site. Authors and their em-ployers have the right to post their IEEE-copyrighted materialon their own web sites without special permission, providingthe paper prominently displays a notice alerting readers to theirobligations with respect to copyrighted material and that theposted work includes an IEEE copyright notice. An example of


an acceptable notice is: This material is presented to ensuretimely dissemination of scholarly and technical work. Copy-right and all rights therein are retained by authors or by othercopyright holders. All persons copying this information are ex-pected to adhere to the terms and constraints invoked by eachauthors copyright. Please refer to http://www.ieee.org/web/publications/rights/policies.html for further details.

In another part of this issue is listed the names of reviewersfor the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATIONover the past 12 months. Without quality reviewers and Asso-ciate Editors, the IEEE TRANSACTIONS would cease to be theleading journal in this specialization of electrical engineering.Conspicuous service of our leading reviewers was recognizedwith the presentation of certificates during the Antennas andPropagation Society Symposium in Charleston last June. Thecriteria were: quality (based on reports and scores of Editorsin ScholarOne); timeliness (within 30 days); and number of re-views completed in the past 12 months (the minimum was arbi-trarily set at ten). Certificates were presented to:

Jorgen Bach Andersen Thomas Ellis Stuart Hay Hon Tat Hui Michael Jensen Tzyh-Ghuang Ma Aldo Petosa John Sahalos Kin-Lu Wong Junho Yeo

It is significant that for all these reviewers the average numberof days taken to review a paper was less than 20 days, the qualityof reviews was typically at the highest score available in Schol-arOne and all exceeded the 10 papers minimum for the year (onereviewed with more than 30). So you see some reviewers areworking very hard indeed for the Transactions!

At the same event, certificates were awarded to high per-forming Associate Editors based on similar criteria, except thatI scored timeliness for each manuscript handled. The AssociateEditors recognized for exemplary service were:

Kwok Leung Duixian Liu Robert Scharstein

I hope that the practice of recognizing reviewers and AssociateEditors will continue after my term concludes later this year.

Finally, the IEEE TRANSACTIONS ON ANTENNAS ANDPROPAGATION has a new web address http://ieeeaps.org/aps_trans/index.htm. This is on the IEEE Antennas and Prop-agation Society web site, the use of which should allowcontinuity of material from one editor to the next.

TREVOR S. BIRD, Editor-in-ChiefCSIRO ICT CentreEpping, NSW 1710, Australia

0018-926X/$26.00 2009 IEEE

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 1, JANUARY 2010 3

Trevor S. Bird (S71M76SM85F97) received the B.App.Sc., M.App.Sc., and Ph.D. degreesfrom the University of Melbourne, Melbourne, Australia, in 1971, 1973, and 1977, respectively.

From 1976 to 1978, he was a Postdoctoral Research Fellow at Queen Mary College, Universityof London, London, U.K., followed by five years as a Lecturer in the Department of Electrical En-gineering, James Cook University, North Queensland, Australia. During 1982 and 1983, he was aConsultant at Plessey Radar, U.K. In December 1983, he joined CSIRO, Sydney, NSW, Australia,where he held several positions and is currently Chief Scientist in the ICT Centre and a CSIROFellow. He is also an Adjunct Professor at Macquarie University, Sydney, Australia. He has pub-lished widely in the areas of antennas, waveguides, electromagnetics, and satellite communicationantennas, and holds 12 patents.

Dr. Bird is a Fellow of the Australian Academy of Technological and Engineering Sciences, theInstitution of Engineering and Technology (IET), London, U.K., and an Honorary Fellow of theInstitution of Engineers, Australia. In 1988, 1992, 1995, and 1996, he received the John Madsen

Medal of the Institution of Engineers, Australia, for the best paper published annually in the Journal of Electrical and ElectronicEngineering, Australia, and in 2001 he was co-recipient of the H. A. Wheeler Applications Prize Paper Award of the IEEE An-tennas and Propagation Society. He was awarded a CSIRO Medal in 1990 for the development of an Optus-B satellite spot beamantenna and again in 1998 for the multibeam antenna feed system for the Parkes radio telescope. He received an IEEE Third Mil-lennium Medal in 2000 for outstanding contributions to the IEEE New South Wales Section. Engineering projects that he played amajor role in were given awards by the Society of Satellite Professionals International (New York) in 2004, the Engineers Australiain 2001, and the Communications Research Laboratory, Japan, in 2000. In 2003, he was awarded a Centenary Medal for service toAustralian society in telecommunications, and also named Professional Engineer of the Year by the Sydney Division of EngineersAustralia. His biography is listed in Whos Who in Australia. He was a Distinguished Lecturer for the IEEE Antennas and Propa-gation Society from 1997 to 1999, Chair of the New South Wales joint AP/MTT Chapter from 1995 to 1998, and again in 2003,Chairman of the 2000 Asia Pacific Microwave Conference, Member of the New South Wales Section Committee from 1995 to2005, and was Vice-Chair and Chair of the Section, from 1999 to 2000 and 2001 to 2002, respectively. He was an Associate Editorof the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION from 2001 to 2004, a member of the Administrative Committee ofthe IEEE Antennas and Propagation Society from 2003 to 2005, and a member of the College of Experts of the Australian Re-search Council (ARC) from 2006 to 2007. He has been a member of the technical committee of numerous conferences includingJINA, ICAP, AP2000, EuCAP, and the URSI Electromagnetic Theory Symposium. He was appointed Editor-in-Chief of the IEEETRANSACTIONS ON ANTENNAS AND PROPAGATION in 2004.


A Dual-Linearly-Polarized MEMS-ReconfigurableAntenna for Narrowband MIMO Communication

SystemsAlfred Grau, Jordi Romeu, Ming-Jer Lee, Sebastian Blanch, Llus Jofre, Member, IEEE, and Franco De Flaviis

AbstractThe design and characterization is described of acompact dual-linearly-polarized reconfigurable 2-port antenna.The antenna can operate in two different selectable linear po-larization bases, thus being capable of reconfiguring/rotating itspolarization base from vertical/horizontal , to slant . The antenna has been implemented on a Quartz substrate,

and uses monolithically integrated micro-electromechanical(MEM) switches to select between the two aforementioned po-larization bases. The antenna operates at 3.8 GHz and presentsa fractional bandwidth of 1.7%. The interest of the proposedantenna is two-fold. First, in LOS scenarios, the antenna enablespolarization tracking in polarization-sensitive communicationschemes. Second, there are the gains of using it in a multiple-inputmultiple-output (MIMO) communication system employingorthogonal space-time block codes (OSTBC) to improve the diver-sity order/gain of the system in NLOS conditions. These benefitswere verified through channel measurements conducted in LOSand NLOS propagation scenarios. Despite the simplicity of theantenna, the achievable polarization matching gains (in LOS sce-narios) and diversity gains (in NLOS scenarios) are remarkable.These gains come at no expenses of introducing additional receiveports to the system (increasing the number of Radio-Frequency(RF) transceivers), rather as a result of the reconfigurable capa-bilities of the proposed antenna.

Index TermsAntenna diversity, dual-linearly polarized an-tenna, micro-electromechanical systems (MEMS), multiple-inputmultiple-output (MIMO) , polarization diversity, reconfigurableantenna.

I. INTRODUCTION

N EW technologies in communications such as soft-ware-defined radio (SDR) and radio-frequency (RF)switches implemented using micro-electromechanical systems(MEMS), present new challenges and opportunities for antennadesign. Antennas have traditionally been assumed to have afixed radiation pattern and polarization at the operating fre-

Manuscript received November 05, 2007; revised July 22, 2009. First pub-lished November 06, 2009; current version published January 04, 2010. Thiswork was supported by the National Science Foundation award ECS-0424454and Project TEC-2007-66698.

A. Grau, M.-J. Lee, and F. De Flaviis are with the University of California atIrvine, Irvine, CA 92697 USA (e-mail: [email protected]; [email protected]; [email protected]).

J. Romeu, S. Blanch, and L. Jofre are with the Universitat Politcnica deCatalunya, 08034 Barcelona, Spain (e-mail: [email protected]; [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.


quency. With the introduction of reconfigurable antennas, itis possible to dynamically change these properties. Reconfig-urable antennas work based on the principle that by alteringthe antennas physical configuration, the current density on theantenna may be controlled in a desirable manner and thereforeits radiation pattern/polarization/frequency can be changed.To change the antennas physical configuration, one can usemicroelectromechanical (MEM) switches or active devicessuch as diodes or field-effect transistors (FETs). By placingthese components in strategic locations over the geometry ofthe antenna, the current paths can be engineered in such away that the resultant radiation patterns of the antenna, or theoperating frequencies, follow some desired requirements.

There is a fundamental Wheeler-Chu-McLean limitation onthe gain as well as hardware limitations when realizing electri-cally small antennas being simultaneously efficient and broad-band. Therefore covering several frequency bands concurrentlywith a single antenna having enough efficiency and bandwidthis a major challenge. As a result, in the literature one can findreconfigurable antenna designs which do not cover all bands si-multaneously, but provide narrower instantaneous bandwidthsthat are dynamically selectable, such as in [1][3]. However,having to excite several radiation patterns and polarization statesconcurrently with a single antenna may also be impractical.Therefore, several designs of pattern [2], [4][7] and polariza-tion reconfigurable antennas exist in the literature.

In particular for polarization reconfigurable antennas, [8], [9]present the design of single-port polarization reconfigurable an-tennas, using PIN diodes, with the capability to reconfigure itspolarization from being left-handed to right-handed circular po-larization. Other designs with similar characteristics, using PINdiodes, were proposed in [10], [11]. Also [10], [12] introducetwo reconfigurable microstrip patch antennas, using PIN diodes,which are able to select between a circular and linear polariza-tion. In [13] a polarization MEMS reconfigurable antenna oper-ating at 26.6 GHz, based on a square shaped microstrip patch,and able to select between a circular and linear polarization fromthe same antenna is presented. Other polarization reconfigurableantennas using MEMS and working in the 26 GHz range havebeen proposed in [14][16]. These single-fed antennas are ingeneral used to change the antenna polarization to operate inmultiple communication standards.

We go one step further in the design of polarization recon-figurable antennas, and we present a compact dual-linearly-po-larized reconfigurable 2-port antenna. Notice that some prelim-inary simulations of the proposed antenna were first introduced

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GRAU et al.: A DUAL-LINEARLY-POLARIZED MEMS-RECONFIGURABLE ANTENNA FOR NARROWBAND MIMO 5

by the authors in [17]. In this paper, we include a complete de-scription of the proposed antenna, we present the design guide-lines and its characterization through simulated and measureddata. To the knowledge of the authors, the proposed antenna isthe first one of its kind being reported in the literature. Simula-tion and measurement results show that the antenna is capable toreconfigure/rotate its polarization base from being vertical/hor-izontal , to slant . MEM switches have been usedto reconfigure the radiation characteristics of the proposed an-tenna. The reasons are several. Traditionally diodes and field-ef-fect transistors (FETs) have relatively larger insertion losses, es-pecially at higher frequencies, in the order of 0.5 dB or larger.On the other hand, MEM switches offer a superior technology,in which the insertion losses are of the order of 0.1 dB [7], [18],[19]. Both technologies are similar in terms of losses at low fre-quencies. However, other advantages of MEMS technology overdiodes include larger isolation, an almost a negligible DC powerconsumption, and most importantly they can be monolithicallyintegrated within the antenna because they can be fabricated oncheap substrates such as PCB or Quartz. Thus MEMS switchesare seen as a powerful technology to be employed in the designof reconfigurable antennas.

In addition, many studies on reconfigurable antennas omitto evaluate the gains introduced by these antennas at systemlevel and in real propagation scenarios. In this paper, we doput emphasis on these aspects. In particular, it was foundthat the interest of the proposed antenna is two-fold. First, inline-of-sight (LOS) scenarios, the antenna has polarizationtracking capabilities, which are of particular interest when usedwith communications schemes where the polarization orien-tation/matching between the transmit and receiver antennasis critical (such as in phase array architectures). Secondly, innon-line-of-sight (NLOS) propagation scenarios, the antennacan be used to improve the diversity order (and array gain)of MIMO communications system employing orthogonalspace-time block codes (OSTBC). This is done by using het-erogeneous polarization configurations among the transmit andreceive antennas [20], [21], and also taking advantage of thefact that reflection, diffraction, and scattering affect differentlyeach polarization, thus producing signals at the receiver withuncorrelated fading statistics. These benefits are quantified ana-lytically. Finally, channel measurements conducted in LOS andNLOS propagation scenarios are used to verify the theoreticalfindings.

The paper is organized as follows: in Section II the proposedantenna is described and measured parameters are presented.Section III introduces the system and channel model used to de-scribe the system level performance of the antenna. Sections IVand V presents the theoretical benefits of using the proposed an-tenna in LOS and NLOS environments, through a phased arrayscheme and a MIMO system using OSTBCs, respectively. Per-formance (channel) measurement results are finally presentedin Section VI. Notation: Throughout this paper we use boldupper-case letters to represent matrices, bold lower-case lettersto represent vectors, and , and to denote transpose,complex conjugate and Hermitian, respectively. representsthe solid angle. where is the elevation angle with

Fig. 1. Schematic of the ORIOL antenna. Notice the two input ports andthe structure of the biasing system using radial stubs. Zoom-in of the feedingsystems showing the structure of the MEM switches, referred in the text by .

origin in the z-axis, and is the azimuth angle with origin onthe x-axis. Notice that .

II. ORIOL ANTENNA

A. DescriptionThe proposed antenna is a compact dual-linearly-polarized

reconfigurable 2-port antenna. Henceforth, we refer to it as theoctagonal reconfigurable isolated orthogonal (ORIOL) elementantenna. The ORIOL antenna consists of a single octagonal mi-crostrip patch, as shown in Fig. 1, in which its two ports al-ways excite two orthogonal polarizations (dual-polarization) ofthe radiated electric field. This is achieved by exciting the patchfrom two points located in perpendicular sides of the octagonalpatch. We refer to these two polarizations as a polarization base.Moreover, the antenna has the capability to reconfigure/rotateits polarization base in two different radiation states, that is,from being vertical/horizontal ( or ), to slant(or ), or viceversa, where no-tice that we use the reference coordinate system given in Fig. 1.Each of the ways in which a particular reconfigurable antennacan radiate is defined as a radiation state. We use index to nu-merate them. If represents the total number of radiation statesin which a reconfigurable antenna can operate, the ORIOL an-tenna is such that . We use indexto numerate the ports of the ORIOL antenna.

A detailed schematic of the structure of the ORIOL antennawith its basic dimensions is shown in Fig. 2(a). Most of the de-sign complexity in the ORIOL antenna resides in the feedingstructure, thus we proceed to describe it. Notice that each port 50

feeding line connects to a quarter wave transformer througha high-impedance line which, after a few millimeters, splits intotwo high-impedance lines that connect to the octagonal patchat two adjacent sides. The purpose of these high-impedancelines is to transform the high input antenna impedance valueseen at the edges of the octagonal patch into a 50 impedancevalue. At each location where the high-impedance quarter wavetransformer splits into two lines, we have located four switches,


Fig. 2. In (a) the basic dimensions of the ORIOL antenna are being shown and(b) represents a zoom-in of the feeding system and the series MEMS-switchstructure.

denoted by , as shown in Fig. 1.For a proper operation of the antenna, the activation of theseswitches has to guarantee that the currents do not pass trough thetwo split high-impedance lines simultaneously, but only throughone of them. I.e., the switches must be con-trolled simultaneously, to ensure that in the first radiation stateof the ORIOL antenna they be both in the electricalstate ON, thus exciting the patch from locations (from port 1)and (from port 2), as shown in Fig. 2(a). Logically, in this ra-diation state, the switches must be both in theelectrical state OFF. If we use to denote the currentdistribution excited from the input port when the ORIOLantenna is set in the radiation state, then in the radiationstate the current distributions excited on the octagonalpatch are . Simulated plots (using HFSS 10[22]) of the aforementioned current distributions are shown inFig. 3(a) and (b). In the second radiation state of the ORIOL

Fig. 3. Representations of the simulated (using HFSS 10) current distributionsassociated with each antenna-port, for the two radiation states. In particular, (a) , (b) , (c) , (d) .

antenna , the switches must be simul-taneously in the ON state and the switches inthe OFF state, such that the patch is excited from locations(from port 1) and (from port 2), as shown in Fig. 2(a). In thiscase, the excited current distributions are ,as shown in Fig. 3(c) and (d). Reconfiguring the location of theinput feeding point along the edge of the antenna, at points ,

, , , results in reconfiguring the polarization of the ORIOLantenna. Because these points are separated by an angular dis-tance of 45 , the two desired polarization bases can be excited.Table I summarizes the logic that relates the radiation states ofthe ORIOL antenna to the electrical state of the MEM switches

in order to excite a particular current distributionone the ORIOL antenna.

Let us define now as the normalized far-field radiation pattern at the port of the ORIOL antenna,when the antenna is operated in the radiation state. Noticethat, is the radiation pattern associated with the currentdistribution . In the first radiation state of the ORIOLantenna , the distribution currentsproduce the radiation patterns . The polar-ization of these radiation patterns in the -axis direction

is and thus the polarization base referred to pre-viously as the vertical/horizontal (or ), is excited. Onthe other hand, in the second radiation state of the ORIOL an-tenna , the distribution currents pro-duce the radiation patterns , which in the

-axis direction are polarized in the directions. This corresponds to the second polar-

ization base of slant . The polarization of these radiationpatterns in the -axis direction and its relation to theelectrical state of the MEM switches, is summarized in Table I.


TABLE ICORRESPONDENCE TABLE BETWEEN THE ORIOL ANTENNA RADIATION STATES AND THE REQUIRED STATE OF THE MEM SWITCHES TO EXCITE A PARTICULAR

POLARIZATION, RADIATION PATTERN AND CURRENT DISTRIBUTION

B. Structure of the Switches and Bias Networks

The feeding and biasing structures, as well as the non-idealisolation capabilities of switches, were previously found to havemeasurable effects on the polarization of the excited fields. Inour case, the switches were implemented with MEMS tech-nology. In particular, the switch used in this work is a capacitiveMEM switch whose structure is based on a double-supportedsuspended membrane over a microstrip line, and was developedduring previous research studies within our group [7], [18], [19],[23], [24]. The basic dimensions of the used MEM switch aregiven in Fig. 2(b). In addition, the gap of the metal bridge is 5

, the thickness of the membrane is 0.5 , and the diam-eter of the membrane holes is 10 . Notice that the switches

are composed of two series-MEM switches in se-ries (instead of one single series-MEM switch), as shown inFig. 1. The reason for this is to improve signal isolation andreduce the coupling of currents on the lines that need to be dis-connected, in each antenna radiation state. At 3.8 GHz, the iso-lation achieved with a single series-MEM switch was found tobe around 16 dB [18], while with two series-MEM switchesin series the isolation level was increased up to 23 dB. Giventhe above isolation level, and to compensate the effect of thefeeding and biasing structures, the exact locations of the fourinput points ( , , , , shown in Fig. 2(a)) were then variedalong the edge and optimized such that the two desired polariza-tion bases ( and slant ), could be excited. By doingthis, the produced radiated fields have a clean polarization (inthe direction) that deviates from the desired one by lessthan 2 .

Finally, notice that the proposed antenna uses a total of eightmonolithically integrated series MEM switches, which arestrategically located in the microstrip feeding structure of theantenna to achieve the desired reconfigurable capabilities. Theactuation voltage of the used MEMS was 30 V. Although thisrequired voltage can be a drawback for certain applications,electrostatically activated MEMS switches are suitable for verylow power consumption applications.

The bias networks are used to activate or deactivate the MEMswitches. These networks are composed of DC control lines andradial open stubs, which are specifically designed to assure thatthe RF signals do not penetrate on the bias networks, that is, tocreate open circuits at RF frequencies. Basically, both the ra-dial open stubs and the microstrip transmission lines that con-nect the radial stubs to either the octagonal patch or the MEMswitches, are quarter-wave transformers [25], which translatethe open-circuit of the radial stubs to the biasing points. Fig. 2(a)shows the location of the bias networks on the surroundings ofthe ORIOL antenna and its basic dimensions.

Fig. 4. Picture of the ORIOL antenna fabricated on a Quartz substrate, andzoom in of one of the four MEM switches .

C. FabricationThe ORIOL antenna was fabricated on a Quartz substrate. A

picture of the antenna and a zoom in of one of the four MEMswitches is shown in Fig. 4. The thickness of theQuartz substrate is , with relative dielectric con-stant , and dissipation factor . Thefabrication process is that developed by the authors and exten-sively described in [19]. The metal used to pattern the microstrippatch and the bias networks is gold with a thickness of 0.5 .

D. Scattering ParametersFigs. 5 and 6, show the simulated and measured scattering

parameters, at the input ports of the ORIOL antenna, over a fre-quency spanning from 3.54 GHz, in the radiation statesand , respectively. In both states, there exists a good iso-lation level between ports of about 3032 dB and the return lossis above 15 dB in any port. As commented in Section II-B, suchlarge isolation values allow the ORIOL antenna to radiate witha very small deviation from the desired polarization. In fact,as it will be shown in the next sections, the co-cross polariza-tion level is around 1830 dB. The 10 dB return loss bandwidthof the ORIOL antenna is 1.7%, and its resonant frequency is3.82 GHz. Notice that in applications with specific bandwidthrequirements, several well known techniques could be used toexpand the bandwidth of this antenna, such as using smaller per-mittivity substrates, or changing the excitation mechanism ofthe patch antenna, among other.

E. Radiation PatternsFig. 7 shows the simulated and measured normalized far-field

radiation patterns of the ORIOL antenna at any of its two ports,in the radiation states and , for the (x-axis)cut plane. Fig. 8 shows the same quantities for the(y-axis) cut plane. At both figures, in the radiation state[sub-figures (a)(b)], the co-polar component for port(a) is set in the direction , and at port (b) the co-polar


Fig. 5. Simulation (gray lines) and measurement (black lines) of the ORIOLantenna scattering parameters in the radiation state .

Fig. 6. Simulation (gray lines) and measurement (black lines) of the ORIOLantenna scattering parameters in the radiation state .

component is set in the direction . In the polarization state[sub-figures (c)(d)], at port (c) the co-polar

component is set in the direction , and at port(d) the co-polar component is set in the direction .The measured maximum gain is 4.9 dBi. This value is relativelow for a patch antenna, but it can be explained from the factthat the metal thickness (0.5 ) is a fraction of the skin depth.Notice that the gain could be easily improved by depositing athicker metal layer. Finally, the cross-polarization componentis always below 18 dB in any port in any of the two state, forthe two cut planes. The radiation patterns have been measuredat the resonant frequency of 3.82 GHz.

III. SYSTEM MODEL

A. System

In order to evaluate the performance of the ORIOL antennaat system level, we first present our system and channel model.Consider a communication system with one ORIOL antenna atthe transmitter in a fixed radiation state, let us assume in the ra-diation state , such that its polarization base is always ver-tical/horizontal . At the receiver, assume one ORIOLantenna which can change its radiation state such that it can se-lect a polarization base of or . Thus, we can definethe following two possible channel propagation states (CPSs)for our system.

CPS 1: transmit ORIOL antenna in the configura-tion and receive ORIOL antenna in the configura-tion.

CPS 2: transmit ORIOL antenna in the configura-tion and receive ORIOL antenna in the .

where we use index to numerate them. The ORIOLantenna arrangements for the two CPSs are illustrated in Fig. 9.Notice that in this work, we do not use reconfigurability at thetransmitter. We assume also perfect knowledge of the channelmatrix at the receiver only. Because the ORIOL antenna is a2-port antenna, the proposed system model can be analyzed asa MIMO communications system with transmit portsand receive ports (2 2 system). For the sequel, we useindex and to numerate the ports of the transmit and receiveORIOL antenna, respectively.

We define now as the channel matrix in thepropagation state, given by

(1)

where denotes the channel coefficient containingthe gain and phase information of the paths between thetransmit port of the ORIOL antenna and the receive portof the ORIOL antenna, during CPS .

B. Channel Model

With the Ricean K-factor, , defined as the ratio of deter-ministic to scattered power, the channel matrix can be ex-panded into

(2)

where the entries are in general correlated complexGaussian random variables with zero-mean. These variablesare used to describe the scattering nature of the (NLOS) prop-agation channel. On the other hand, the entries of aredeterministic variables which describe the LOS component ofthe channel. can be directly computed from the transmit


Fig. 7. Co (solid line) and cross (dotted line) polarization components of the normalized radiation pattern for the cut plane. In the polarization state (a), (b), at port (a) the co-polar component is set in the direction , and at port (b) the co-polar component is set in the direction . In the polarizationstate (c), (d), at port (c) the co-polar component is set in the direction , and at port (d) the co-polar component is set in thedirection

. Measurements are in black color, and simulations in gray color.

and receive antenna radiation patterns and the propagation lossfactor of the LOS component [26], [27], as follows:

(3)

where and are the angle-of-departure (AoD) and theangle-of-arrival (AoA) of the LOS path component with respectto the local coordinate systems at the transmitter and receiver,respectively, and are the maximum gain associated withthe and ports of the transmit and receive ORIOL an-tennas, respectively, and is the distance between the trans-mitter and the receiver. Notice that denotes the radiation statesof the receive ORIOL antenna, in the CPS. Unless specifiedotherwise, we use .

For the scattering component of the channel, we assume theKronecker channel model described in [28] ((2)). This channelmodel has been widely used in the literature and in several IEEEstandards (such as IEEE 802.11n) [29]. Contrary to [29], we donot model the power delay profile (i.e., narrowband assumption)and Doppler effects, since we aim to measure the performancegains due to the spatial diversity provided by the ORIOL an-tenna.

IV. POLARIZATION-TRACKING USING THE ORIOL ANTENNAIN LOS ENVIRONMENTS

Due to the capability of the ORIOL antenna to reconfigureits polarization base from being vertical/horizontal to slant

, it can be used to perform polarization tracking in polar-ization-sensitive communication schemes, such as phased arrayarchitectures. Phased array architectures have been shown to bethe optimal transmission technique when [30], [31],that is in LOS conditions where the channel matrix is given by

. The advantage of phased array schemesover other architectures is that only long-term channel stateinformation consisting of the relative location of the transmitterand receiver, needs to be sent back to the transmitter. However,one drawback of phased array using dual-linearly-polarizedantennas is the antenna orientation. Having an unexpectedpolarization misalignment between the transmit and receiveantennas may cause important losses on the Signal-to-NoiseRatio (SNR) of the system. The use of circularly polarizedantennas is sometimes proposed to overcome this limitation,because they do not require any alignment [8]. However, theseantennas have the inconvenience that the axial ratio (whichis a measure of the quality of the circularly polarized waves)tends to increase rapidly as the scanning angle from boresightincreases, thus rendering serious reductions on the SNR onthese angles. In addition, the axial-ratio bandwidth for mi-crostrip antennas is normally much smaller than the 10 dBbandwidth. On the other hand, MEMS technology enables us todesign linearly-polarized reconfigurable antennas that can trackthe polarization of the incoming waves, such as the ORIOLantenna, and thus solve the problem of antenna orientationwithout having to use circularly-polarized antennas.


Fig. 8. Same representation as in Fig. 7, but for the cut plane. Measurements are in black color, and simulations in gray color. (a) State at port ; (b) state at port ; (c) state at port ; (d) state at port .

Fig. 9. ORIOL antenna arrangements for the two channel propagation states of the proposed system. (a) Channel propagation state 1 : V-H to V-H. (b)Channel propagation state 2 : V-H to 45 degrees.

For simplicity we limit our analysis to the 2 2 systemdescribed in Section III, using one transmit and one receiveORIOL antenna. However, the conclusions are valid for phasedarray structures with an arbitrary number of transmit and receiveantennas. Let us define as the transmitted vector, givenby

(4)

where the same symbol is sent from the two ports of thetransmit ORIOL antenna. The variance of is given by

, where is the average energy per data symbolat the transmitter. On the other hand, where

is the average transmit power from all transmit ports. Thereceived spatial vector can be written as follows

(5)where is the received spatial of additive whiteGaussian noise (AWGN) vector. Using (3) and (5), the com-bined received signal can then be expressed as follows:

(6)

Notice that the operation in (6) consists of directly combiningthe signals from all the receive ports, as it is similarly done in


analog phased arrays through an external power combining net-work. Assume that and are also the maximum gaindirection of the transmit and receive antennas, and that

. Finally, using (6), the average receive SNR of thesystem in the CPS , denoted by , can be written as

(7)

where represents the noise power. To illustrate thebenefits of the ORIOL antenna, assume in this section,a third CPS, given by: CPS 3: transmit ORIOL an-tenna in the configuration and receive ORIOLantenna in the configuration. Notice that inthe CP3, ,

and . Therefore and nopower is being delivered. On the other hand, for the othertwo CPSs, it holds thatand , and therefore

. Thus, a phased array system using reconfig-urable ORIOL antennas at the receiver, will pick the radiationstate of the receiving antennas that maximizes the receive SNR,that is , thus tracking the polar-ization of the transmit antennas. Notice that since the transmitand receive ORIOL antennas can be oriented randomly, othercombinations of polarization bases between the transmitter andthe receiver may arise, however, those will produce SNR valuewithin and . To summarize, for a non-reconfigurabledual-polarized phased array architecture, the ratio of worst tobest SNR is

(8)

while for a reconfigurable dual-polarized phased array architec-ture using the ORIOL antennas, such ratio is

(9)

which means that the percentage of power received varies inbetween 50% and 100%, instead of 0% to 100% in the formercase. As shown above, the ORIOL antenna does not completelysolve the problem of polarization mismatch in linearly polar-ized phases array architectures, but proves the fact that recon-figurable antenna technology can be used to solve this problem.To solve the problem completely, one would need to use a po-larization reconfigurable antenna in which the polarization basecan be changed at smaller angular steps than 45 , which is afeasible feature.

V. DIVERSITY GAIN ENHANCEMENT USING THE ORIOL INNLOS ENVIRONMENTS

As shown in [21], [32], in NLOS propagation scenarios, re-configurable antennas (such as the ORIOL antenna) can be usedto improve the diversity order of a MIMO system using orthog-onal space-time block codes (OSTBC) [33], where OSTBCs area kind of widely used modulation schemes designed for quasi-static flat fading channels. These systems have been shown tobe excellent schemes in NLOS propagation scenarios ,because they provide maximum diversity gain with very simpledecoding complexity. In these systems, the instantaneous SNRat the input of the decoder, , can be written as [33]

(10)

where we define , with andis the average transmit power from all transmit antennas. Wedefine the average array gain (in the CPS ) of such systemas . This quantity gives us an insighton the average received signal power as a result of coherentlycombining the signals propagating through the channel fromthe transmit antennas to all the receiving antennas. Notice thatOSTBC systems, in contrast to phased array architectures, arenot sensitive to antenna orientation, because the SNR is pro-portional to the summation of the squared absolute values ofthe channel coefficients. The reconfigurable capabilities of theORIOL antenna can be used to improve the array gain (andthe diversity order) of such systems. A simple way to do it isby allowing the system to select, among the two CPS given inSection III, the CPS that maximizes the receive SNR [21]. Wedenote the optimal state by . Assume an ideal NLOS prop-agation scenario and that we use the Alamouti OSTBC [34].Assume for now that the channel matrices and , cor-responding to the two possible CPSs, are iid Gaussian randommatrices with zero-mean and unit-variance entries. Then, forthe proposed reconfigurable MIMO system using OSTBCs, theSNR is given by

(11)

where . The average SNR at thedecoder, , is henceforth given as

(12)

where it is possible to find a close-form expression for, given by

(13)


where in our case, and is thecoefficient of , in the expansion of

(14)

A more general expression of can be found in [28]for the case in which the ports of a reconfigurable antenna areallowed to be reconfigured independently of each other (as it isnot the case in the proposed antenna). For the sake of complete-ness, in [28] it is also shown that the diversity order of such asystem would be . Notice that the expression given aboveis a simplification of that given in [28].

In general, and , are correlated Gaussian randommatrices, and therefore the expression of given in (13)represents an upper-limit. In addition, the diversity order of thesystem would be larger than but smaller than [28].Correlation may appear due to the characteristics of the antennaor lack of scattering richness in the propagation channel. As-suming for now an ideal NLOS environment (uniform transmitand receive power spectrums), we can estimate the transmit andreceive correlation matrices from the measured radiation pat-terns only [28], [35]. That is, the entries of the transmit correla-tion matrix, as defined in [28], are given by

(15)

and the entries of the receive correlation matrix among CPSsand are computed by

(16)

where for simplicity we assume and . Equation (17)summarizes the receive envelope correlation values computedfrom the measured 3D normalized far-field radiation pattern.As shown, because in the proposed system using the ORIOLantenna , then and , are correlatedGaussian random matrices. However, notice that the correlationvalues are small enough ( 0.7) to provide significant diversitygain, as commented in [36], and as it was verified through mea-surements

(17)

VI. MEASUREMENT RESULTS

A. Measurement Setup

To compute the received power (or SNR) for any of the twoaforementioned architectures, one needs to collect the ampli-tude and phase information of the channel coefficients .This information can be extracted using a virtual array technique(VAT) [37]. Using this method, two antennas are moved alongdistinct linear trajectories, virtually creating the transmit andreceive arrays of a particular MIMO system. For each CPS ,the coefficients are then computed by switching amongthe ports of the transmit and receive antennas, then sending asingle tone at the operating frequency of the antennas (3.8 GHzin our case), and measuring the propagated signal with a vectornetwork analyzer connected to both the transmit and receiveantennas. In our case, the experiments were conducted withina room with several metallic objects and walls, as shown inFig. 10. For the NLOS measurements, the two robotic arms thatmove the antennas, were placed in a NLOS configuration andseparated by a distance of approximately 6 m. During the LOSmeasurements, they were placed in a LOS configuration andseparated by a distance of 1.5 m. Two ORIOL antennas wereused, one being installed in the transmit arm and the other in thereceive arm. During the measurements, the transmit and receiveORIOL antennas were moved at steps of and , respec-tively. One thousand realizations of the channel matricesand were collected. The samples of the channel matrices

and were sequentially obtained on a time-invariantchannel. To verify the time-invariant channel assumption, thecorrelation among two sequentially obtained sets of samples of

was measured and found to be 0.98.

B. Polarization-Tracking Measurements in LOS EnvironmentsIn this section the presented measurement results are

conducted in a LOS environment. Fig. 11 shows the mea-sured cumulative distribution function (CDF) of the quantity

(in dB), which is proportionalto the receive SNR, for the phased array architecture using onetransmit ORIOL antenna (with fixed radiation state ) andone receive ORIOL antenna, as described in Section IV, in eachof the CPSs. Notice that in the CPS 1, when the polarizationsof the transmit and receive antennas are perfectly aligned, thereceive power is maximum. In the CPS 3, when the polarizationmisalignment is maximum, the receive power drops about 8 dB.Notice that the receive power in the CPS 3 is not zero, whichcan be explained by the fact that the channel is not ideallyLOS, and the radiated polarizations by the ORIOL antennasmay deviate from the ideally desired ones by about 2 , ascommented in Section II-B. For the same reason, the receivepower in CPS 2 only drops about 1.75 dB below that of CPS1. However, the trend of the measurements agree well with thefindings presented in Section IV.

C. Diversity and Capacity Gain Measurements in NLOSEnvironments

The performance gains of a MIMO system using OSTBCsand reconfigurable ORIOL antennas are now verified through


TABLE IICORRESPONDENCE TABLE BETWEEN THE CPS IN CASE 1 AND THE REQUIRED STATE OF THE MEM SWITCHES TO EXCITE A PARTICULAR POLARIZATION

AND RADIATION PATTERN

Fig. 10. Layout and dimensions of the room where the LOS and NLOS mea-surements were conducted.

Fig. 11. Measured CDF of the quantity for a phased array architecture usingone transmit (with fix radiation state) and one receive ORIOL antenna, for thethree CPSs described in Section IV. These measurements were conducted in aLOS environment.

channel measurements in a NLOS environment. Two cases areconsidered:

Case 1: system described in Section III; Case 2: system described in Section III where the receive

ORIOL antenna is followed by an external switch that se-lects one of the two output ports according to the receiveSNR.

Notice that while case 1 can be analyzed as a 2 2 MIMOsystem with two CPSs, case 2 can be analyzed as a 2 1 MIMOsystem with four CPSs. Tables II and III describe the logic thatrelates the CPSs, the electrical state of the MEM switches, theactive port selected by the external switch (case 2 only), andthe excited radiation pattern and polarization, for case 1 and 2,respectively.

We first investigate the correlation properties of the measuredchannel. From the measured channel coefficients it is possibleto estimate the transmit and receive correlation matrices. In par-ticular for the receive correlation matrix, it is given by

(18)

for any value of . Equations (19) and (20)

(19)

(20)

summarize the measured receive envelope correlation for case1 and case 2, respectively, computed using (18). From our mea-surements we observe that thus and ,are correlated Gaussian random matrices. Notice that the enve-lope correlation values follow similar trends to those obtainedfrom the measured radiation patterns assuming an ideal NLOSscenario (see (17)). This is in agreement with the fact that themeasurements are conducted in a NLOS environment (non-ideal


TABLE IIICORRESPONDENCE TABLE BETWEEN THE CPS IN CASE 2 AND THE REQUIRED STATE OF THE MEM SWITCHES AND EXTERNAL SWITCH TO EXCITE A PARTICULAR

POLARIZATION AND RADIATION PATTERN

Fig. 12. Measured CDF of (in dB), for case 1. Solid lines represent theCDF curves of in each one of the CPSs, while the dashed line represents theCDF curve after selecting the optimal CPS, for each measured sample. NLOSmeasurements.

but) with rich scattering objects. In both cases, from thechannel measurements.

1) Diversity Gain: Fig. 12 and Fig. 13 show the measuredCDF of (in dB), for cases 1 and 2, respectively. In Fig. 12(case 1), the received power is that collected from the twoports and combined using maximal-ratio-combining (MRC).In Fig. 12 (case 2), the received power is that collected from asingle port after selecting the optimal radiation state and portof the receiving ORIOL antenna. In both figures, the solidlines represent the CDF curves of in each CPSs, while thedashed line represents the CDF curves of after selectingthe optimal CPS, for each measured sample.

We define now the incremental array gain, , as the ratioof received power, at a probability of , using the reconfig-urable ORIOL antenna and selecting the optimal CPS tothe received power in any of the states. can be expressedas

(21)

Notice that is in fact the diversity gain of the system asa result of using the ORIOL reconfigurable antenna. In case 1,

is equal to 1.13 dB, and in case 2 it equals 3.86 dB.

Fig. 13. Measured CDF of (in dB), for case 2. Solid lines represent theCDF curves of in each one of the CPSs, while the dashed line represents theCDF curve after selecting the optimal CPS, for each measured sample. NLOSmeasurements.

On the other hand, the average incremental array gain, ,computed as

(22)

is equal to 0.74 dB and 1.77 dB, for cases 1 and 2, respectively.These diversity gains come at no expenses of introducing ad-ditional receive ports to the system (increasing the number ofradio-frequency (RF) transceivers), but rather as a result of thereconfigurable capabilities of the ORIOL antenna. Using (13),the theoretical values of the average incremental array gain as-suming that and are iid Gaussian random matrices,are 1 dB and 2.5 dB, for cases 1 and 2, respectively. Notice thatthe measured average values are slightly below those predictedthrough theory, which has to do with the fact that in our case,the channel matrices and are correlated Gaussianrandom matrices. Also notice that the received power in case 1is larger that in case 2. This has to do with the fact that in case1 we are combining the received power from two ports usingMRC, while in case 2 only one receiving port is available afterthe external switching mechanism. On the other hand, due tothe fact that in case 2 not only the optimal radiation state of theORIOL antenna is selected but also the optimal output port, theincremental array gain is logically larger than in case 1. For thesake of completeness, the above curves are finally compared to


Fig. 14. Simulated bit-error rate vs. SNR curves for the case 1 of the proposedsystem, using the Alamouti code and binary phase-shit key (BPSK) modulation.

those of a Single-Input Single-Output (SISO) system (1 1),consisting in using only the port 1 in the transmit and receiveORIOL antennas, and not allowing the receive ORIOL antennato reconfigure its states.

In Fig. 12 and Fig. 13, each CPS guarantees a different levelof received power. This can be explained by the fact that re-flection, diffraction, and scattering affect differently each polar-ization [26]. Notice that in these figures the channel matrices

are not normalized and correspond to those directly ex-tracted from measurements.

Finally, Fig. 14 shows the bit-error (BER) rate vs. SNRcurves for the proposed system (case 1), using the Alamouticode [34] and Binary Phase-Shift Key (BPSK) modulation. Toobtain these curves we have conducted Monte Carlo simulationsusing the Kronecker channel model given in Section III-B andthe measured complex correlation values associated with thosegiven in (19). Notice that the diversity order (defined as theslope of the curves when ) of the MIMO-OSTBCsystem using the ORIOL antenna is larger than that of anon-reconfigurable system with the same number of transmitand receive ports. In fact, at a BER probability of , theimprovement on the SNR is about 2.1 dB. These curves arecompared also with that of an ideal reconfigurable system inwhich and are iid. In all the cases, the channelmatrices have been normalized according to [38]. Notice thatthe diversity order of the system using the ORIOL antenna isin between and , due to the fact that

, and in particular it is approximately equal to6, which is equivalent to the diversity order of a 2 3 MIMOsystem using OSTBCs. However, only two receive RF trans-ceivers are needed in the system using the ORIOL antenna.Therefore, from a cost perspective, reconfigurable antennas,such as the ORIOL antenna, allow us to build cheaper RF frontends.

2) Capacity Gain: Fig. 15 and Fig. 16 show the CDF of thesystem capacity (in bits/s/Hz) for cases 1 and 2, respectively.The system capacity, , assuming that the transmitted power

Fig. 15. Measured CDF of the system capacity (in bits/s/Hz) for case 1. Solidlines represent the CDF curves of the capacity in each one of the CPSs, whilethe dashed line represents the CDF curve after selecting the optimal CPS, foreach measured sample. NLOS measurements.

is equally distributed among the transmit antennas, is de-fined as

(23)

In both figures, the solid lines represent the CDF curves of thecapacity in each one of the CPSs, while the dashed line repre-sents the CDF curve after selecting the optimal CPS, for eachmeasured sample. As shown in Fig. 15 and Fig. 16, the capacitygain, computed as the increase on the system capacity at a 10%probability due to the benefits introduced by using the recon-figurable ORIOL antenna, is equal to 0.44 bits/s/Hz and 1.86bits/s/Hz, for cases 1 and 2, respectively. These capacity gainsresult from the increase in received power, as commented inSection VI-C-1. Finally, notice that when using (23), the channelmatrices have also been normalized according to [38].

VII. CONCLUSIONA compact dual-linearly-polarized reconfigurable 2-port an-

tenna was designed, fabricated and tested. Measurements haveshown that the antenna is capable of reconfiguring/rotating itspolarization base from being vertical/horizontal , toslant . The antenna has been implemented on a Quartz sub-strate, and uses monolithically integrated micro-electromechan-ical (MEM) switches to select among the two aforementionedpolarization bases. MEMS technology has been chosen in thiswork because MEM switches are able to offer small insertionloss, large isolation, almost a negligible DC power consump-tion, and most importantly they can be monolithically integratedwithin the antenna because they can be fabricated on cheap sub-strates such as PCB or Quartz. The return loss level seen at theantenna ports was found to be always above 15 dB, and theisolation among ports larger than 30 dB. The measured max-imum gain is 4.9 dBi and the fractional bandwidth 1.7%. Thesystem level performance of the ORIOL antenna has also been


Fig. 16. Measured CDF of the system capacity (in bits/s/Hz) for case 2. Solidlines represent the CDF curves of the capacity in each one of the CPSs, whilethe dashed line represents the CDF curve after selecting the optimal CPS, foreach measured sample. NLOS measurements.

investigated analytically and through measurement. In LOS sce-narios, it has been shown that the ORIOL antenna has polar-ization tracking capabilities, which are of particular interest inpolarization-sensitive applications, such as in phased arrays. InNLOS environments, when used on MIMO system employingOSTBCs, it has been probed that the proposed antenna improvesthe diversity gain/order of the system. As shown, despite thesimplicity of the antenna, the achievable polarization matchinggains (in LOS scenarios) and diversity gains (in NLOS sce-narios) are remarkable. These diversity gains come at no ex-penses of introducing additional receive ports to the system (in-creasing the number of radio-frequency (RF) transceivers), butrather as a result of the reconfigurable capabilities of the ORIOLantenna.

ACKNOWLEDGMENTThe authors are thankful for the support of the Balsells fel-

lowships and the California-Catalonia Engineering InnovationProgram 20042005.

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Alfred Grau was born in Barcelona, Spain, in 1977.He received the Telecommunications Engineeringdegree from the Universitat Politecnica de Catalunya(UPC), Barcelona, Spain in 2001 and the M.S.degree and Ph.D. degree in electrical engineeringfrom the University of California at Irvine (UCI), in2004 and 2007, respectively.

He is currently working as a Scientist at BroadcomCorporation. His research interest are in the fieldof miniature and integrated antennas, multi-portantenna (MPA) systems, MIMO wireless commu-

nication systems, software defined antennas, reconfigurable and adaptiveantennas, channel coding techniques, microelectromechanical systems(MEMS) for RF applications, and computer-aided electromagnetics.

Jordi Romeu was born in Barcelona, Spain, in 1962.He received the Ingeniero de Telecomunicacin andDoctor Ingeniero de Telecomunicacin degrees, bothfrom the Universitat Politecnica de Catalunya (UPC),in 1986 and 1991, respectively.

In 1985, he joined the Photonic and Electromag-netic Engineering Group, Signal Theory and Com-munications Department, UPC, where he is currentlya Full Professor and is engaged in research on an-tenna near-field measurements, antenna diagnostics,and antenna design. He was a Visiting Scholar at the

Antenna Laboratory, University of California, Los Angeles, in 1999, on a NATOScientific Program Scholarship, and in 2004 at University of California Irvine.He holds several patents and has published 35 refereed papers in internationaljounals and 50 conference proceedings.

Dr. Romeu was grand winner of the European IT Prize, awarded by the Euro-pean Commission, for his contributions in the development of fractal antennasin 1998.

Ming-Jer Lee received the B.S. and M.S. degreesin physics from the National Tsing Hua University,Hsin-Chu, Taiwan, R.O.C., in 1990 and 1994, respec-tively, and the Ph.D. degree in electric engineeringfrom the University of California, Irvine, in 2006.

From 1996 to 2001, he was the section head ofthe thin lm process section in Fab4 Taiwan Semi-conductor Manufacturing Co. (TSMC), Hsin-Chu,Taiwan, R.O.C. He was in charge of dielectric andmetal lm deposition including various CVD andsputter.

Sebastian Blanch was born in Barcelona, Spain, in1961. He received the Ingeniero and Doctor Inge-niero degrees in Telecommunication Engineering,both from the Polytechnic University of Catalonia(UPC), Barcelona, Spain, in 1989 and 1996, respec-tively.

In 1989, he joined the Electromagnetic and Pho-tonics Engineering Group, Signal Theory and Com-munications Department, UPC, where he is currentlyan Associate Professor. His research interests are an-tenna near field measurements, antenna diagnostics,

and antenna design.

Llus Jofre (M78) was born in Barcelona, Spain, in1956. He received the M.Sc. (Ing) and Ph.D. (DoctorIng.) degrees in electrical engineering (telecommu-nications engineering) from the Technical Universityof Catalonia (UPC), Barcelona, in 1978 and 1982, re-spectively.

From 1979 to 1980, he was a Research Assistantwith the Electrophysics Group, UPC, where heworked on the analysis and near-field measurementof antenna and scatterers. From 1981 to 1982, hewas with the Ecole Superieure dElectricite, Paris,

France, where he was involved in microwave antenna design and imagingtechniques for medical and industrial applications. In 1982, he was appointedAssociate Professor with the Communications Department, Telecommunica-tion Engineering School, UPC, where he became a Full Professor in 1989.From 1986 to 1987, he was a Visiting Fulbright Scholar at the Georgia Instituteof Technology, Atlanta, working on antennas, and electromagnetic imagingand visualization. From 1989 to 1994, he served as Director of the Telecom-munication Engineering School (UPC), and from 1994 to 2000, he was UPCVice-rector for Academic Planning. His research interests include antennas,scattering, electromagnetic imaging, and wireless communications. He haspublished more than 100 scientific and technical papers, reports, and chaptersin specialized volumes. During 2000 and 2001, he was a Visiting Professorwith the Electrical and Computer Engineering Department, Henry SamueliSchool of Engineering, University of California, Irvine, where he focused onantennas and systems miniaturization for wireless and sensing applications.

Franco De Flaviis was born in Teramo, Italy, in1963. He received the Laurea degree in electronicsengineering from the University of Ancona, Italy,in 1990, and the M.S. degree and Ph.D. degreein electrical engineering from the University ofCalifornia at Los Angeles (UCLA), in 1994 and1997 respectively.

In 1991, he was an Engineer with Alcatel workingas a researcher specializing in the area of microwavemixer design. In 1992, he was a Visiting Researcherat the UCLA working on low intermodulation mixers.

Currently, he is an Associate Professor in the Department of Electrical and Com-puter Engineering, University of California Irvine. His research interests are inthe filed of computer-aided electromagnetics for high-speed digital circuits andantennas, microelectromechanical systems (MEMS) for RF applications fabri-cated on unconventional substrates such as printed circuit board and microwavelaminates.

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