Return Loss Layout Guidelines for the Video Equalizer

June 24, 2008 Document No. 001-46578 Rev. ** 1

AN46578 Author: Mukund Krishna Associated Project: No

Associated Part Family: CYV15G010xEQ, CY270M010xEQ GET FREE SAMPLES HERE

Software Version: None Associated Application Notes: AN6003

Application Note Abstract High speed data transmission has always introduced issues of return loss, where signal integrity is affected due to changes in impedances in the signal path. The video cable equalizer recovers high speed digital video signals after transmission through long coaxial cables. This application note introduces the concept of return loss and discusses the techniques, based on circuits and PCB layouts, to improve the return loss performance of an equalizer.

Introduction The Cypress Video Equalizer is a part of Cypress’ broadcast video portfolio and provides the vital function of signal equalization. It is designed specifically to meet SMPTE 259M, 344M, 292M, and DVB-ASI specifications. The video equalizer is capable of equalizing up to 200 meters of coaxial cable at 1.485 Gbps and 350 meters at 270 Mbps.

A critical SMPTE specification for these devices is return loss, measured at the inputs of the equalizer. When signals cross from the cable to the equalizer inputs or from the cable driver outputs to the cable, there is a change in the medium’s impedance. This causes signal reflections, measured as return loss.

This application note shows how the prudent use of matching networks and certain board layout methods can improve return loss performance of the equalizer. It discusses the importance of return loss and methods to measure it. It also describes termination circuits and matching networks used specifically for the equalizer to minimize return loss. Finally, it analyzes board layout guidelines to minimize return loss with specific reference to the equalizer.

Return Loss Return loss is defined at the point of intersection of media with different impedances, as the measure of reflected energy from a signal transmitted to that point.

When an electrical signal is transmitted through a medium such as copper wire, a certain amount of the signal’s energy is reflected from the point where the impedance of the transmission medium changes. A common example of such impedance changes in real world applications is when a coaxial cable ends in a load with impedance that differs from the characteristic impedance of the cable. The larger the change in impedance, the more the signal is reflected back. Other examples of impedance discontinuities are when a

cable terminates into a connector, change in trace path or shape on a PCB, damage to cable due to long term physical duress, and improper installation of cables and connectors.

Mathematically it is defined as follows:

)](|)([|20)( LOLO ZZZZLogdBRL

Equation 1

Zo = impedance of transmission line

ZL = impedance at load

This usually results in positive dB values because the numerator is smaller than the denominator. A large magnitude indicates a small loss and vice versa.

S-Parameters

Like Y-Parameters, S-parameters (scattering parameters) are commonly used for high frequency circuits to supply information about a network. Instead of using currents and voltages, S-parameters use normalized incident and reflected waves at the port of measurement.

Figure 1. Incident and Reflected Waves in a 2-Port Network

AC

Two PortZo Zo

Zs

ZL

Input Output

a1

b1

a2

b2

Figure 1 represents a high frequency signal being fed into a 2-port network that processes it and transmits a signal into a load. On both the reception and transmission sides of the

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network, the transmission mediums have a characteristic impedance of ZO. ZS represents the source impedance and ZL represents the load impedance.

a1: Wave from source incident on 2-port device.

b1: Part of wave from source reflected by 2-port.

a2: Part of transmitted wave reflected by load and incident on output of 2-port.

b2: Part of a2 reflected by output of 2-port back to load.

The total wave traveling from the output of 2-port to the load includes the part of a2 reflected by the port’s output and the part of a1 transmitted through the 2-port network.

Similarly, the total wave traveling from the input of 2-port back to the source includes the part of a1 reflected by the port’s input and the part of a2 transmitted through the 2-port network.

This translates to the following equations:

2121111 aSaSb Equation 2

2221212 aSaSb Equation 3

S11 and S22 represent the input and output reflection

coefficients respectively. S11 is calculated by setting a2 to

zero. Similarly, S22 is calculated by setting a1 to zero. Plotting these on a Smith chart, it is possible to calculate the input and output impedances of the port.

Most commonly, the parameter used to denote differential return loss is S11, also known as SD11.

Therefore, in decibels,

0)/()_/]_([20)( 21111 whenaabrefVinrefVinVportLogdBS Equation 4

In a system with an equalizer, the outputs of the device are fed to another chip or are terminated. Therefore, to draw a comparison with Figure 1, a2 becomes zero because there is no reflection from the load.

Where Vport is the voltage measured at the port and Vin_ref is the voltage sent in at the port. The difference represents the reflected wave.

With more high speed transmission links using differential signaling as a way to build noise immunity into a system, differential return loss measured from S-parameters are also of prime importance. A later section discusses important termination and layout techniques to ensure good return loss when using differential signaling.

Importance of Return Loss for the Video Equalizer At higher frequencies such as High Definition (HD) video rates, return loss becomes very important. This is because impedance mismatches cause more signal reflections at these frequencies, which in turn distorts the original signal being transmitted to a greater extent.

The Society of Motion Picture and Television Engineers (SMPTE) places a limit on the return loss in a system based on data rate. For frequencies from 0–1.485 GHz (HD data rate), the return loss in a system must not exceed 15 dB. From 1.485–2.97 GHz (2xHD data rate), the return loss must not exceed 10 dB. With digital video transmission and distribution transitioning to HD and 2xHD, return loss in any cable, connector, or board is a very critical parameter to be understood and minimized by design.

Therefore, the return loss contributed by the equalizer, which is an endpoint of the serial link, must be better than the limits specified for the system. In addition to the device’s port and package (layout of the board, proximity of components, trace geometry, and power and ground planes), the return loss compensation network and connectors are all important factors that affect final return loss.

Measuring Return Loss This section details how single ended and differential return loss is measured and how graphs such as Smith charts are interpreted.

Network Analyzer Based Measurement

A commonly used method of measuring the return loss on a single port is to use an S-parameter based network analyzer such as the Agilent 8722ES. This analyzer is capable of measuring and graphically displaying the return loss over frequency. When using network analyzers, remember that many of them do not have 75 ohm port interfaces, which is the characteristic impedance defined for broadcast video transmission links. Most commonly, they have a 50 ohm interface; therefore, a 50 to 75 ohm converter is needed. The steps to measure return loss are as follows:

1. Calibrate the instrument according to product specifications using the correct calibration kit provided by the manufacturer or an equivalent one.

2. Set the start and stop frequencies for measurement.

3. Set the markers at the desired frequency points.

4. Connect the input port of the board to the network analyzer’s port and turn on power for the board. The board should be populated similar to any fully functional board to get a realistic measurement. For single-ended return loss measurement, connect only one port (the input port in question) to one of the ports of the network analyzer. To measure differential return loss, connect the differential input port to a pair of ports on the network analyzer as specified in the user guide of the analyzer. Most modern analyzers can directly report the S11 differential return loss parameter.

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5. Depending on the analyzer being used, measurement is displayed in a variety of ways. Graphical depictions are the logarithmic scale graph or a Smith chart.

The log scale graph (Figure 2) is very easy to read. Note the return loss values (displayed in dB on the y-axis) for each frequency point (on the x-axis in log scale). The image also provides a quick insight into the trend of return loss across the frequency range.

The Smith chart is an older graphical method, but continues to be used. Although it requires some calculation to obtain the actual return loss numbers, the Smith chart provides other useful information.

Figure 2. Log Scale Graph Showing Return Loss over Frequency

Figure 3 shows a typical Smith chart. The chart is a graphical representation of the mathematical properties of a network. It is widely used in the RF world to represent port impedances, in impedance matching, and to determine S-parameters and standing wave ratios. The circles in green are constant resistance circles and the arcs in red are constant reactance arcs. The red arcs on the positive y-axis represent inductive reactances and those below represent capacitive reactances. A point plotted on the chart is at the intersection of a constant resistance and constant reactance circle. This essentially decides the impedance represented by that point R + jX.

As an example, consider a normalized impedance of 1 + j2. This point lies at the intersection of the RL = 1 and XL = 2 circles, as shown by point A on the graph.

Smith charts are also made with an angle scale on the outside as shown in yellow. A line is drawn from the center of the chart through point A to the edge of the chart.

The complex reflection coefficient ( ) in rectangular coordinate form can be read as 0.5+j0.5. If this is converted

to the polar form, it becomes 0.707 45 . This number is also measured from the Smith chart.

The straight line through point A meets the angle scale on the perimeter of the chart; the angle can now be read. The magnitude is basically the length of the straight line from the center to point A.

Return Loss = –20 Log ( ) = –20 Log (0.707) = 3.01 dB

Many analyzers report the return loss in negative dB, which is technically correct. However, it is often reported as a positive number by convention.

Figure 3. Smith Chart Source: http://www.tmworld.com/article/CA187342.html

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Measuring Return Loss for the Equalizer

When measuring return loss (RL) contributed by the video equalizer, ensure the following:

The board is populated with the device and all other components, as it would be for full functionality.

Power is supplied to the board within limits of the device or suitable for normal operation.

For the equalizer, because the RL is measured at the inputs, there is no need to give an input signal to the device.

All unused inputs (equalizer) and outputs are terminated to 75 ohms.

TDR Based Measurement

A modern method of measuring return loss is by using the Time Domain Reflectometry (TDR) ability present in current wide-bandwidth oscilloscopes, such as the Agilent 86100C with the 54754A Differential TDR module. The final return loss measurement is usually an output of the post measurement processing software present in the oscilloscope.

In TDR, a fast rise time pulse is transmitted through the medium. When this pulse meets impedance discontinuities, a certain amount of the pulse’s energy is reflected back. This depends on the difference between the characteristic impedance of the medium and the impedance at the point of discontinuity.

This returning pulse is measured by the reflectometer and plotted against time (hence time domain). This procedure is used to determine the characteristic impedance of cables and traces, check for impedance discontinuities, and determine length of the transmission medium such as cables or traces.

Additional software present in the equipment converts the time domain measurement to a frequency domain value and reports the S-parameter values. This is represented similar to the network analyzer, with the return loss on the y-axis in dB and the frequency on the x-axis in log scale.

The steps to measure return loss using a TDR module are as follows.

1. Calibrate the TDR module of the scope using the calibration kit recommended by the manufacturer.

2. A normal calibration of the TDR module usually calibrates it to the output of the module. Often, connectors, cables, or converters are required to interface with a board. Despite this, the measurement must be taken for the specific DUT, which is the board. Therefore, the calibration software on many modern oscilloscopes allows the user to calibrate the response of the connectors, converters, and cables.

3. The response window of the TDR measurement must be adjusted to enable measuring the response of the signal after it is reflected.

4. When this is done, the DUT is connected to the port of the TDR module. The TDR module shows the response in the time and frequency domains. In the frequency domain, it displays the S-parameters on a log scale based graph similar to the network analyzer. The y-axis is the S11 in dB and x-axis is the frequency sweep in a log scale.

The accuracy of this type of measurement has been questioned, but it has now become more acceptable to measure S-parameters this way.

Techniques to Optimize Return Loss for the Equalizer

Setting the Correct State of the EQ/CD

Apart from the return loss network at its inputs, the equalizer has protection diodes for ESD protection. These have a certain parasitic capacitance. When the power supply is on, these diodes are reverse-biased and their parasitic capacitance decreases. This results in a better return loss measurement. To get accurate return loss measures, the equalizer must be powered up as it would for normal operation.

Termination

To counter repeated signal reflections in a transmission link, impedance is added to the transmission line in various ways to match the characteristic impedance of the transmission line. This is known as termination.

Single Ended Termination

Termination matches the ends of a transmission line (where the line meets a source or a load) with impedance equal to that of the line. This ensures that all waves are absorbed and reflections are minimized or avoided altogether.

Figure 4. Single Ended Source Termination

TXRX

75ohm

Rs = 75 ohms

When a circuit is source terminated, there are reflections returning from the receiver, but these get absorbed by the source terminating resistance. Often RS is less than 75 ohms; therefore, the total effective resistance of RS and the source intrinsic resistance are 75 ohms.

A disadvantage is that the signal amplitude is reduced.

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Figure 5. Single Ended End Termination

TXRX

75ohm

ZL = 75 ohms

When a circuit is end terminated, any signal arriving at the receiver is absorbed completely and there are no reflections. There is, however, power dissipated at this parallel impedance.

Differential Termination

In a differential pair, each line must be terminated to match the characteristic impedance of the line.

A common way to do it is to double the impedance and connect it across the differential pair because the midpoint of the true and complement of the differential pair is an AC ground.

The termination components must be placed as close to the receiver as possible so that another transmission line is not formed between the terminators and the receiver.

Figure 6. Termination of a Differential Line

TX RX150ohm

75ohm

75ohm

75ohm

Impedance Matching Networks The need for a matching network arises because of loss and distortion of signal due to reflections. The maximum power transfer theorem also states that for maximum power transfer to occur, the source and load impedances must match. This follows from the fact that impedance discontinuities in a signal’s transmission path cause reflections, which implies power loss.

Therefore, an impedance matching network:

Matches source and load impedances that are different.

Matches one impedance that is constant and characteristic to the medium to the other that may vary with frequency.

Matching Networks for Video Broadcasting Devices

In the video broadcasting world, the matching network for equalizers is usually a parallel network of an inductor and a resistor. This is the preferred network because it counters the effect of the pad capacitance at the serial

interfaces of these devices when they are mounted on the board.

Figure 7. Return Loss Matching Network for Video Equalizer

Coaxial

cable(Zo)

L

R

Rterm

C

Matching Network

Equalizer input

with pad

capacitance

Cpad

The schematic in Figure 7 shows the input return loss matching arrangement for serial input of the video equalizer. There is a finite capacitance to ground from the serial input due to the board layout. The capacitor between the device and the matching network is the AC coupling capacitor. The matching network comprises the inductance in parallel with the resistance, followed by a terminating resistance to ground.

At low frequencies, the pad capacitance acts as an open circuit and the inductor acts as a short. Therefore, the impedance shown into the network after the cable is Rterm. At intermediate frequencies, the equivalent termination is Rterm + (parallel equivalent of XL and R).

At high frequencies, the pad capacitance is a short. At the same time, the inductor presents an open circuit and the equivalent termination is R.

The resistor value R is selected and the inductor is tuned such that at the interest frequency (approximately 1.5 GHz for video), the impedance into the serial inputs by the cable remains more or less the same and very close to the characteristic impedance of the cable itself (75 ohms).

A rough estimate of the inductance value can be reached with the help of the pad material’s capacitance properties. Subsequently, different values of inductance can be substituted to optimize the actual return loss (as measured on a network analyzer). This provides a good selection of inductance.

Note Refer to application note AN6003, Interfacing the HOTLink II™ Transceiver with the Cypress CYV15G0101EQ/CYV15G0102EQ Equalizer and CYV15G0101CD Cable Driver for Digital Video and HOTLink II CYV15G0404 Video Demonstration Board User’s Guide for examples of termination circuits and impedance matching networks for the equalizer and cable driver.

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Board and Layout Guidelines for the Video Equalizer The equalizer’s function is to receive a video signal that has come over a long distance of cable (up to 200 meters at HD data rates) and restore the signal quality such that the downstream receiver is able to successfully use it without any degradation in video quality. The signal is at its weakest at the input of the equalizer.

Some of the key aspects to consider when designing boards for a video equalizer are as follows.

Tracks or Traces

For voltage reflections, any trace with a propagation delay of more than twice the rise or fall time of the signal is considered a transmission line. This means, it can be affected by reflections. If the trace’s propagation delay is such that the signal gets reflected and returned to the source when the signal is still rising, it gets absorbed. If the signal has already risen, this leads to distortion of the signal.

To calculate the propagation velocity of a transmission line with a known dielectric material, use the following equation:

r

CV Equation 5

V = velocity of propagation through the medium

C = speed of light.

εr = dielectric constant of dielectric material of board.

At high frequencies, a transmission line becomes equivalent to a distributed parasitic capacitance and can distort the signal.

The video transmission system is defined for a characteristic impedance of 75 ohms. If the cables that transmit the signal are 75 ohms, the traces on the boards where these signals originate or terminate should be 75 ohms to avoid discontinuity in impedance in the signal path. Track width must be calculated carefully so that tracks have the desired characteristic impedance of 75 ohms.

Figure 8 shows a cross section of the boards. εr

represents the dielectric constant of the dielectric material used in the board. ‘H’ represents the height of the dielectric material, ‘t’ represents the thickness of the trace, and ‘W’ represents the width of the trace.

These are the actual numerical factors that determine the characteristic impedance of the trace.

Figure 8. Cross Section of a Trace

ohmsTW

HnZ

r

o8.0

98.51

41.1

87 Equation 6

Figure 9. Traces Ending in Teardrop

Source: www.smtnet.com/bob-willis/pbdai.html

One basic rule is to end the trace at the pad with a ‘tear-drop’ shape instead of abruptly as shown in Figure 6. This avoids sudden impedance changes.

The kind of transmission line that has optimum signal integrity is the ‘coplanar wave-guide with lower ground’. It is a surface micro-strip transmission line surrounded by a copper area and connected to the ground with vias every 1 cm.

Current flows in a closed loop. In a PCB, the longer and more convoluted the loop is, the worse it is for signal integrity. This is because a longer or more convoluted loop results in a higher inductance in the current path, which in turn degrades the signal quality. Ideally, the ground plane should be directly below the signal trace and should have no interruptions in the return path such as vias or routing via multiple layers. In addition to signal integrity, current loops that are long produce EMI and are susceptible to EMI from other traces.

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Cutouts

When the board has bodies on the surface (using pads), with ground and power planes below, there is a stray parasitic capacitance between the planes and the bodies or pads. The pads are conducting, the board has some dielectric material, and there could be a conducting power or ground plane below. This in essence is a capacitor.

This extra capacitance adds to the effect of the input network of the equalizer (and also output network) and affects the return loss at the port. To avoid this, the ground and power planes are cut out from underneath the input network and pins (Figure 17).

Placement of Input and Output Network Components

The input network optimizes the return loss of the device ports. Therefore, to the external world, the network appears as part of the device characteristics. This means the components must be placed close to the input or output of the device.

Additionally, the device itself must be placed as close to the connectors as possible so that the trace length for carrying the signal in or out is minimum.

Figure 10. Layout Practices for Differential Signaling

w

tb

r = Dielectric constant

s

Differential Signaling

Some important practices to be followed for differential signaling layout are as follows:

Keep the differential pairs as short as possible.

Keep the pairs to the same layers and do not route through any vias as this leads to impedance changes.

Ensure a large spacing between two pairs to reduce impedance variations and any effect of a neighboring pair.

Keep the intra-pair and inter-pair spacing as constant as possible.

There are two capacitances involved: one between the top via pad and the reference plane above it and one between the bottom via pad and the reference plane below it. The

inductance is that of the barrel and contributes a series inductance to the signal path.

The effects of a via are as follows:

The capacitance slows the edge rate of the signal.

If the signal passes through several consecutive vias, the effective characteristic impedance is decreased.

The series inductance adds to the system’s inductance and affects the signal integrity by affecting characteristic impedance.

Vias and Connectors

A via is a barrel shaped connector used to transfer a signal path between different layers on a board. The via is electrically modeled as shown below.

Figure 11. Electrical Model of a Via

C_pad_layer2C_pad_layer1

L_barrel

Avoid use of vias for critical, high speed signals or try to minimize them as much as possible. This is because consecutive vias change the characteristic impedance of the medium.

For connectors, minimize the length of the connectors so that the inductance is kept to a minimum, which in turn reduces impedance discontinuities.

Debugging the Board for Return Loss

Figure 2, which illustrates the return loss measurement taken by a network analyzer, shows clear peaks and valleys. This graph is useful in determining the impedance characteristics of the board on the path of the signal.

Return loss depends directly on the difference between the characteristic impedance of the medium and the load. If at a particular frequency, a certain capacitive or inductive reactance peaks, this causes the load impedance ZL to either increase or decrease.

If this movement of ZL is such that it increases the difference between Zo and ZL, the return loss also peaks. If on the other hand, the movement is such that it decreases the difference between the two, the return loss decreases.

If the return loss profile is flat all through and not bumpy, it shows that the impedance component is not sensitive to frequency, which can only mean that the load is mostly resistive.

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Figure 12. Peaks and Valleys on a Return Loss Graph

The TDR method of measuring return loss is also useful in verifying the board design and fabrication quality. In the time domain measurement mode on the oscilloscope, connecting the inputs or outputs of a board to the TDR module’s ports result in the impedance profile of the transmission line. When configured with the correct dielectric constant and the horizontal scale is set to display length in meters, the profile on the scope shows the impedance against length. Any discontinuities in impedance show up on this profile and can be used to verify if the board is fabricated according to design and if that design results in the desired transmission line characteristics.

This response can be used to vary placement of terminating components and the value of the inductor in the return loss network.

Figure 13. TDR of a Cable Driver Board Output

Case Study: Optimizing the Equalizer Evaluation Board for Return Loss This section briefly describes the exercise to optimize the design of the video equalizer evaluation board. It showcases how the layout guidelines detailed previously is used to optimize return loss.

Figure 14. QFN Video Equalizer Evaluation Board

To start with, the board was designed similar to the one shown in Figure 14, with minor differences. The board shown in Figure 14 gives superior return loss performance and is used for this exercise.

Table 1 gives details of the six board variants used.

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Table 1. Board Variants

Version RL Network Placement

Trace Width AirGap BNC Center Pad Extended

BNC via Thermal Relief

Cutout under BNC Center Pad

3.0a Inside 14 mil 37 mil No Yes No

3.0b Outside 14 mil 37 mil No Yes Yes

3.0b+ Outside 14 mil 37 mil Yes Yes Yes

3.0c Outside 14 mil 145 mil Yes NA Yes

3.0d Inside TearDrop 37 mil No No Yes

3.0e Outside 40 mil 37 mil No No Yes

Version 3.0a denotes the base version of the 3.0 revision of boards. Compared to the previous revision, the thermal reliefs of the BNC connector’s upper and lower soldering pads were removed. In addition, direct vias to ground were added within the BNC connector to improve signal return. This is common to all variants in this revision.

The ‘Trace Width’ field denotes the width of the traces from the BNC connector to the Return Loss network and then on to the SDI inputs of the equalizer. 14 mil is the calculated width for a characteristic impedance of 75 ohms using a 2D field solver. The ‘TearDrop’ and ’40 mil’ trace widths in revisions 3.0d and 3.0e respectively are an attempt to smoothen the transition from a trace to the soldering pads of the components.

The ‘AirGap’ field refers to the distance between the BNC connector’s center pad and the closest metal plane. 37 mil is the calculated value from the 2D field solver for a characteristic impedance of 75 ohms. 145 mil is the interpreted gap from the BNC connector (Trompeter UCBBJE20) data sheet.

For versions 3.0b+ and 3.0c, the BNC connector’s center pad is extended based on the interpretation of the Trompeter specification sheet.

Versions 3.0a, 3.0b, and 3.0b+ have the BNC via thermal reliefs present to connect the pad to ground.

Except for the base version 3.0a, power and ground plane cutouts are present in all the versions.

Figure 15. Version 3.0 a

No cutouts

Figure 16. Version 3.0 b

BNC via

thermal

relief

Power and

Ground plane

cutouts

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Figure 17. Version 3.0 b+

Figure 18. Version 3.0 c

Extended Center Pad

Figure 19. Version 3.0 d

Teardrop shaped pads

Figure 20. Version 3.0 e

Test Procedure

For every board revision, two boards are built and populated with a Cypress Video Equalizer each.

The return loss measurements are taken using the network analyzer using the procedure detailed earlier.

Tests are performed at room temperature with a supply voltage of 3.3V.

Table 2. Results of Return Loss Measurement

Version Return Loss 50 Mhz–1.485 Ghz (populate with CY

device)

Return Loss 1.485 Ghz–3 Ghz (populate with CY

device)

3.0a -19.5 dB -6.5 dB

3.0b -15.8 dB -6.9 dB

3.0c -14.0 dB -6.7 dB

3.0d -27.2 dB -10.5 dB

3.0e -21.4 dB -8.2 dB

3.0b+ -19.2 dB -7.0 dB

2.0 -16.5 dB -4.5 dB

The table shows the results of the return loss measurements and version 3.0d is the clear winner. The table also shows return loss performance comparison using Gennum video equalizers on the same boards. The same board revision shows up as the best for this exercise too.

Figure 21 shows the return loss profile across frequency taken from the network analyzer of revision 3.0d board.

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Figure 21. Return Loss Profile of Version 3.0d across Frequency

The profile shows that return loss at a frequency of 1.485 GHz is approximately 27 dB and at 3.0 GHz is approximately 10 dB. Between these two frequencies, the return loss is always lower than that at 3.0 GHz.

This meets SMPTE’s return loss specifications at the relevant data rates.

In addition to this, tests with different inductor values in the return loss matching network show that an inductor value of 5.6 nH results in the best return loss performance (approximately 30 dB at 1.485 GHz).

Summary This application note demonstrates the importance of return loss to design of high speed signaling systems. Previously, only RF systems fit this category, but current video transmission rates have increased enough to make return loss a critical design specification for the video broadcasting systems. Therefore, it is important for everyone, from semiconductor manufacturers to board and system designers, to understand the implications of the choice of board layout design, components, materials, and circuit topology.

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Document History Document Title: Return Loss Layout Guidelines for the Video Equalizer

Document Number: 001-46578

Revision ECN Submission Date

Orig. of Change Description of Change

** 2518142 06/18/08 UKK New application note.

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