Using the ATF-10236 in Low Noise Amplifier Applications inÊthe UHF through 1.7 GHz Frequency...

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Using the ATF-10236 in LowNoise Amplifier Applicationsin the UHF through 1.7 GHzFrequency Range

Application Note 1076

IntroductionGaAs FET devices are typicallyused in low-noise amplifiers in themicrowave frequency regionwhere silicon transistors can’tprovide the required gain andnoise performance. There are,however, many applications in thefrequency range below 2 GHzwhere the low noise figures andhigh gain of GaAs FETs canimprove receiver sensitivity.Typical applications include lownoise amplifiers (LNAs) in the 800to 900 MHz frequency range foruse in celluar telephone and pagerapplications and spread spectrumtransceiver applications. Addi-tional applications include the1228 and 1575 MHz frequenciesused for Global PositioningSystem (GPS) applications. Otherapplications include VHF mobileradio, IMMARSAT, and WEFAX,just to name a few.

This application note describestwo low-noise amplifiers that usethe Hewlett-Packard ATF-10236low noise GaAs FET device. Bothdesigns use identical circuittopology with the only differencesbeing in the proper choice of threeinductors depending on thefrequency of operation. The

designs are centered at 900 MHzand 1575 MHz, but can be scaledfor any frequency within theregion of 400 to 1700 MHz. Eachamplifier has a usable bandwidthof about 30 to 40 percent.

Using a high-gain, high-frequencyGaAs FET at VHF poses specialproblems. Of greatest concern isthe problem of designing theamplifier for unconditionalstability. Typically, GaAs FETshave greater gain as frequency isdecreased, e.g., 25 dB maximumstable gain at 500 MHz. A secondproblem is that matching thetypical microwave GaAs FET atlower frequencies for minimumnoise figure does not necessarilyproduce minimum input VSWR.

Achieving the lowest possiblenoise figure requires matching thedevice to Γopt (the source matchrequired for minimum noisefigure). At higher microwavefrequencies this will generallyproduce a reasonable input VSWR,since Γopt and the complexconjugate of the device inputreflection coefficient S11 areusually close on the Smith Chart.At lower frequencies, specialconsideration needs to be given to

the input circuit design and to thetradeoffs required to ensure lownoise figure while still achievingmoderate gain, low VSWR andunconditional stability.

Device FamilyThis application note will discussthe use of the ATF-10236 series oflow noise GaAs FET devices.The device has a 500 micron gateperiphery and is most suitable forapplications in the VHF through 4GHz frequency range. The deviceis tested for noise figure and gainat 4 GHz where it is typicallyused in satellite TVRO applica-tions. The ATF-10136 is thepremium device being specifiedat 0.6 dB maximum noise figurewhile the ATF-10236 is specifiedat a 1.0 dB maximum and theATF-10736 is specified at a 1.4 dBmaximum all at 4 GHz. Typicallya three stage device lineup isused at C band with the ATF-10136 device being used as thefirst stage followed by the ATF-10236 followed by the ATF-10736.The higher noise figure of thesecond and third stages hasminimal effect on the over allcascade noise figure. A threestage low-noise amplifier withgreater than 30 dB of gain is

5966-0166E

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generally required at C band toovercome filter/mixer losses.

While there is considerable differ-ence in noise figure at 4 GHzbetween the three devices (i.e. 0.8dB), the difference in the 1 to 2GHz frequency range is less thanseveral tenths of a dB. In actualcircuits built on low cost FR-4dielectric material, the ATF-10236device is capable of a 0.5 dB noisefigure at 900 MHz and a 0.75 dBnoise figure at 1575 MHz. Mostcommercial applications at fre-quencies below 2 GHz generally donot require the high gain of a threestage cascade so generally a singlestage LNA is used followed by abipolar device or a silicon MMIC.Cascading the ATF-10236 with a3.5 dB noise figure MMIC such asthe MSA-0686, will still result inabout a 1 dB noise figure LNA with25 to 30 dB gain.

The Hewlett-Packard ATF-10236 issupplied in the low cost commer-cial 0.100 inch “micro-X” metal/ceramic package. Examination ofthe data sheet reveals that thedevice is capable of a 0.6 dB noisefigure at frequencies below 2 GHzwith an associated gain of greaterthan 16 dB. The noise parametersand S-parameters of this transistorare summarized in Table 1.

Design TechniqueObtaining the lowest possiblenoise figure from the devicerequires that the input matchingnetwork convert the nominal50 Ω source impedance to Γopt.This produces a deliberate imped-ance mismatch that, while minimiz-ing amplifier noise figure, pro-duces a high input VSWR. Theideal situation is where Γopt is thecomplex conjugate of S11 (i.e.,

S11*). For this condition, mini-mum noise figure is achieved whenthe device is matched for mini-mum VSWR. This situation occurspredominantly above 2 GHz andtends to diverge at lower frequen-cies, where S11 approaches 1.

High input VSWR has varyingsignificance, depending on theapplication. Most noteworthy isthe increased uncertainty of thenoise figure measurement due toreflections between the noisesource and amplifier input. Havinga noise source with a very lowoutput VSWR and one whoseVSWR has minimal change be-tween the “on” and “off” states willminimize this uncertainty. Onesuch noise source is the Hewlett-Packard HP346A with a nominal 5dB Excess Noise Ratio (ENR).Similarly, when the amplifier isconnected to a receive antenna,high input VSWR creates addeduncertainty in overall systemperformance. The effect is diffi-cult to analyze unless an isolator isplaced at the input to the amplifier.The use of an isolator, however,adds excessive loss and, at VHFfrequencies, the size of the isolatoris often prohibitively large.

To examine the alternatives,constant noise figure and constantgain circles can be constructed toassess the impact of tradingincreased noise figure for adecrease in input VSWR and acorresponding increase in ampli-fier gain. In most instances, theresult is a much higher noise figurethan really desired. One option isto use source feedback. Thissubject has already been coveredby several authors (References 1-3). Source feedback, in the formof source inductance, can improve

input VSWR with minimal noisefigure degradation. The drawbackof utilizing source inductance is again reduction of up to severaldecibels. However, GaAs FETdevices often have more gain thandesired at low frequencies, so thepenalty is not severe.

The effect of source inductance onamplifier input match is beststudied with the help of a com-puter simulation. The microwavedesign simulation program fromHewlett-Packard EEsof calledTouchstone™ is used to analyzeS11 of the amplifier with theproposed output matching net-work. S11 was measured lookingdirectly into the gate of the devicewith the source inductance addedbetween the source and ground.With the ATF-10236 at 500 MHz,adding the equivalent sourceinductance of two 0.10 inch leadscauses the value of S11 to decreasefrom 0.970 to 0.960. Angle remainsrelatively constant at about -20degrees. Γopt remains relativelyunchanged with the addition ofsource inductance. Comparing S11to Γopt at 500 MHz now showsthem to be nearly identical. Theresult is that minimum noise figureand minimum VSWR will coincidemore closely with one anotherwhen matching the device forminimum noise figure. PlottingΓopt for the ATF-10236 device from450 MHz through 2 GHz in Figure 1shows that Γopt lies very near theR/Z0=1 curve. This implies that aseries inductance will provide thenecessary match to attain mini-mum noise figure.

The simplest way to incorporatesource inductance is to use thedevice source leads. Using deviceleads as inductors produces

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mode is invaluable for determiningthe optimum lead length for a givenperformance. Table 2 shows theeffect of lead length on gain, noisefigure, stability, and input andoutput VSWR at 900 MHz. It isclear that lead lengths of 0.06 inchor less have a minor effect on noisefigure while improving input matchsubstantially. Gain does suffer, butthis is not a major concern.

An added benefit of using sourceinductance is enhanced stability asevidence by the Rollett stabilityfactor, k. Excessive sourceinductance can have an adverseeffect on stability at the higherfrequencies. In the case of the 900MHz amplifier, zero length sourceleads create potential instability atlow frequencies while longersource lead length creates poten-tial instability at high frequencies,i.e. 6.6 GHz. It is determined that0.060 inch source lead length is anoptimum choice based on allparameters. The optimum sourcelead length varies with frequencyof operation. At 1575 MHz, 0.020inch source lead length provides

effect of the lead inductance canbe analyzed by simulating theinductance as a high-impedancetransmission line. The TUNE

approximately 1.3 nH per 0.100inch of source lead, or 0.65 nH fortwo source leads in parallel. Withthe help of Touchstone™, the

Table 1. Scattering and Noise Parameters for the Hewlett-Packard ATF-10236 GaAs FET,Vds = 2 volts and Ids = 25 mAScattering Parameters: Common Source, Zo = 50 Ω

Freq S11 S21 S21 S22GHz Mag Ang dB Mag Ang dB Mag Ang Mag Ang

0.5 .97 -20 15.1 5.68 162 -32.8 .023 76 .47 -111.0 .93 -41 14.9 5.58 143 -26.0 .050 71 .45 -232.0 .77 -81 13.6 4.76 107 -21.3 .086 51 .36 -38

Noise Parameters

Frequency Noise Figure Gamma Optimum Rn/50GHz dB Mag Ang normalized

0.5 0.45 0.93 18 0.751.0 0.5 0.87 36 0.632.0 0.6 0.73 74 0.33

Figure 1. ATF-10236 Γo vs. Frequency at Vds = 2 volts and Ids = 25 mA.

0.2

0.5

1.0

2.0

1.0

0.5

0.5

0.2

0.2

1.0

2.0

2.0

2000 MHz

1000MHz

500MHz

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optimum performance withunconditional stability up to 12GHz. Decreasing source leadlength improves stability at 12 GHzwhile making k<1 at 400 MHz.Adding series resistance in theoutput matching circuit increasesstability over a wide frequencyrange. The effect of adding a seriesresistor R10 on amplifier perfor-mance is also shown in Table 2.

Achieving the associated gain ofwhich the device is capable isdifficult since the device is notinherently stable. It is not enoughthat the amplifier be stable at theoperating frequency – it must bestable at all frequencies. Any out-of-band oscillation will make theamplifier unusable.

The simplest technique to ensurebroad-band stability is to resis-tively load the drain. Resistiveloading produces a constantimpedance on the device over awide frequency range. In the case

of the ATF-10236, a 50 Ω resistorand a small amount of seriesinductance is used to load theoutput of the device. The seriesinductance provides some imped-ance matching. This producesacceptable gain while ensuring agood output match and retainingstability over as wide a bandwidthas possible.

The amplifier circuit is shown inFigure 2. For simplicity, theoriginal LNA used the self- biasingtechnique to set the bias point.The loss in noise figure associatedwith the bypassed source resistortopology is no greater than 0.1 dBat these frequencies. The 33 Ωresistor between the source andground sets the drain currentwhile the 100 Ω resistor sets thedrain voltage. There is someinteraction, however, between thetwo resistors. The supply voltageis regulated by the TL05 5 voltregulator U2. The disadvantage ofthe self biasing technique is thatvariations in Pinchoff Voltage (Vp)and Saturated Drain Current (Idss)

that may occur from one produc-tion run to another may dictate theneed to change the value of thesource resistor. The calculation todetermine the source resistor RS isas follows.

Table 2. Simulated 900 MHz amplifier performance vs. source lead length , R10 = 0 Ω.

Lead Noise K at 900 K at 6.6 K at 10Length LL1 Figure Gain |S11|2 |S22|2 MHz GHz GHz

0 inch 0.54 dB 20.1 dB -4.9 dB -8.8 dB 0.58 2.94 1.650.06 inch 0.56 dB 18.0 dB -12.2 dB -12.7dB 1.011 2.39 1.090.2 inch 0.62 dB 14.4 dB -12.7 dB -18.0 dB 1.57 0.35 1.47

Simulated 900 MHz amplifier performance vs. source lead length , R10 = 16 Ω.

Lead Noise K at 900 K at 6.6 K at 7.5 K at 10Length LL1 Figure Gain |S11|2 |S22|2 MHz GHz GHz GHz

0 inch 0.56 dB 18.7 dB -4.6 dB -12.0 dB 0.66 4.05 3.05 2.150.06 inch 0.58 dB 16.6 dB -11.7 dB -13.6 dB 1.22 3.36 2.50 1.150.2 inch 0.65 dB 13.2 dB -14.7 dB -13.3 dB 1.96 0.35 -0.14 1.98

This agrees favorably with the33 Ω resistor used in the actualamplifier.

The preferred alternative in aproduction environment is the useof an active bias network asdescribed in Figure 3. Althoughactive biasing does add cost byrequiring extra components,including a way of generating a

RS =(1 –[ Vp

Id

Id/ Idss ) ]

Assuming typical device parameters of:

yields:

Idss = 130 mAVp = -1.3 voltsId = 25 mA

Rs = 29.2 ohms

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negative voltage, the advantagesgenerally outweigh the disadvan-tages. Active biasing offers theadvantage that variations in VP andIdss won’t necessitate a change ineither the source or drain resistorvalue for a given bias condition.The active bias network automati-cally sets Vgs for the desired drainvoltage and drain current.

The active biasing scheme forFETs requires that the source leadsbe grounded and an additionalsupply be used to generate thenegative voltage required at thegate for typical operation. Directgrounding the FET source leadshas the additional advantage of notrequiring bypass capacitors tobypass a source resistor that wouldtypically be used for self biasing in

a single supply circuit. When thesource bypass capacitors areremoved , the source lead lengthmust be increased as suggested inFigure 3 to offset the decrease inseries inductance. When activebiasing is used, the cold end of the100 ohm resistor, R1, in the gatenetwork must be bypassed toground with a 1000 pF capacitorinstead of being dc grounded. Thisallows gate voltage to be applied.

Referring to Figure 3, resistors R6and R8 provide a regulated voltageat the base of Q2. The voltage isincreased by 0.7 volts by virtue ofthe emitter-base junction of Q2and then applied to the drain of Q1through resistor R3. Since R3 isincluded in the RF matchingcircuit, the voltage drop across R3

must be taken into account whendesigning the bias circuitry.Resistor R9 is connected betweentwo regulated voltage points andtherefore sets the drain current. Q1gate is connected to a voltagedivider consisting of R5 and R7connected between the collector ofQ2 and a negative voltage con-verter. The gate voltage can thenassume a value necessary tosustain the desired drain voltageand current as predetermined byR6, R8, and R9.

Measurements OnAmplifiersThe actual measured performanceof the amplifiers compares favor-ably to that predicted by thecomputer simulation. Table 3summarizes the gain, noise figure

Figure 2. Schematic of the GaAs FET amplifier using passive biassing. The only change required to modify the frequencyoperation is the proper choice of RFC1, L1, and LL1/LL2.

Zo Zo

Zo Zo Zo ZoU3INPUT C1 L1Q1

L2

R1

R2

LL1/2

C2 C3

C7

R3 R4

C9

C12C13

U2R9

OUTPUT

Vcc

C1, C2, C3 = 100 pF CHIP CAPACITOR

C5, C6, C7, C9 = 1000 pF CHIP CAPACITOR

C12, C13 = 0.1 µF CAPACITOR

Q1 = HEWLETT-PACKARD ATF-10236 GaAs FET

R1 = 100 Ω CHIP RESISTOR


R3, R9 = 50 Ω CHIP RESISTOR

R10

COMPONENTS COMMON TO ALL AMPLIFIERS:

FREQUENCY 900 MHz 1575 MHzRFC1 0.33 µH 0.18 µHL1 5T #26 2T #28

0.075" ID 0.050" IDCLOSEWOUND CLOSEWOUND

LL1/LL2 0.060" 0.020"

R4 = ADJUST FOR DESIRED MMIC CURRENT

R10 = 5 TO 25 Ω CHIP RESISTOR USED TO IMPROVE STABILITY AND OUTPUT RETURN

LOSS (SEE TEXT)

U2 = 5 VOLT REGULATOR, TOKO TK11650U

U3 = HEWLETT-PACKARD MSA-SERIES MMIC

PROVIDES ADDITIONAL GAIN - OPTIONAL

Zo = 50 Ω MICROSTRIPLINE

C5 C6

COMPONENTS CHANGED WITH FREQUENCY OF AMPLIFIER

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Figure 3. Schematic of the GaAs FET amplifier using passive biasing. The only change required to modify the frequencyoperation is the proper choice of RFC1, L1, and LL1/LL2.

and VSWR parameters. The noisefigure of the 900 MHz amplifier isactually a little lower than thesimulation would predict while thegain is very comparable. The inputreturn loss measured 5.6 dB versus12.2 dB according to the simula-tion. The difference betweenmeasured and simulated may bedue to the loss of the RF choke inthe input circuit. The input VSWRcould be improved by increasingsource inductance while payingcareful attention to stability. Thelast alternative is to sacrifice somenoise figure for a better input

match. Output return loss mea-sured 13.2 dB which comparesfavorably to the 12.7 dB predictedby the computer simulation.

The noise figure of the 1575 MHzamplifier was measured at 0.73 dBwhich is higher than the 0.45 dB aspredicted. This can be explainedby the dielectric board losses ofthe FR-4. The loss of the FR-4accounts for 0.3 dB of loss at 1575MHz. This was verified by cuttingoff the input section of the board ,attaching two SMA connectors andcarefully measuring the loss.Measured gain with the output

series resistor R10 measured 12.5dB which is within 1.3 dB of thecomputer simulation. Gain in-creases to 14.3 dB without R10.Input return loss measured 6.7 dBwhile the output return lossmeasured 15 dB which is fairlyclose to the computer simulation.Stability is very good with noproblems noted when cascadingstages.

The swept gain plots (included inFigures 4-5) show the wide band-width of these amplifiers. Lownoise figure is also retained overthe bandwidth. The 900 MHz

Zo Zo

Zo Zo Zo ZoU3INPUT C1 L1Q1

L2

R1

R5

LL1/2

C2 C3

C8C7C4

R3 R4

C9

C12

C10

C11

C13

U2R9R7

R6 R8

Q2

OUTPUT

Vcc



C5, C6, = NOT REQUIRED, REPLACE WITH COPPER STRAP

C8, C12, C13 = 0.1 µF CHIP CAPACITOR

C10, C11 = 10 µF CHIP CAPACITOR

Q1 = HEWLETT-PACKARD ATF-10236 GaAs FET

Q2 = SIEMENS SMBT 2907A PNP TRANSISTOR


R2 = NOT REQUIRED


U1

R10

COMPONENTS COMMON TO ALL AMPLIFIERS:

FREQUENCY 900 MHz 1575 MHzRFC1 0.33 µH 0.18 µHL1 5T #26 2T #28

0.075" ID 0.050" IDCLOSEWOUND CLOSEWOUND

LL1/LL2 0.090" 0.050"

NOTE: LENGTH OF LL1 AND LL2 INCLUDES LENGTH OF

COPPER STRAP USED TO REPLACE C5 AND C6.

R4 = ADJUST FOR DESIRED MMIC CURRENT

R5, R7 = 10K Ω CHIP RESISTOR

R6 = 1.1K Ω CHIP RESISTOR

R8 = 1K Ω CHIP RESISTOR


R10 = 5 TO 25 Ω CHIP RESISTOR USED TO IMPROVE STABILITY AND OUTPUT RETURN

LOSS (SEE TEXT)

U1 = LINEAR TECHNOLOGY LTC1044CS8 VOLTAGE CONVERTER

U2 = 5 VOLT REGULATOR, TOKO TK11650U

U3 = HEWLETT-PACKARD MSA-SERIES MMIC

PROVIDES ADDITIONAL GAIN - OPTIONAL

Zo = 50 Ω MICROSTRIPLINE

COMPONENTS CHANGED WITH FREQUENCY OF AMPLIFIER

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Table 3. Measured amplifier performance vs. computer simulation(* indicates R10 = 16 Ω, ** indicates R10 = 0 Ω)

NoiseFrequency Figure Gain |S11|2 |S22|2

900 MHz measured 0.46 dB 16.7 dB -5.6 dB -13.2 dBsimulated 0.56 dB 18.0 dB -12.2 dB -12.7 dB

1575 MHz measured 0.73 dB 12.5 dB * -6.7 dB -15.0 dB14.3 dB **

simulated 0.85 dB 13.8 dB -8.3 dB -12.2 dB

amplifier has a maximum of 0.5 dBnoise figure between 800 and 1000MHz. Similarly, the 1575 MHzamplifier has less than a 1 dB noisefigure from 1200 MHz to 1700 MHz.

At frequencies above 2 GHz, theATF-10236 is rated for minimumnoise figure when operated at Vdsof 2 V, and Id of 25 mA. At fre-quencies below 2 GHz, it wasempirically determined that anadditional 0.1 dB reduction innoise figure is possible if thedevice is rebiased. At 900 MHz, 1 Vgave the lowest noise figure while1.5 volts is optimum for 1575 MHz.

Using The Design AtOther FrequenciesThe basic amplifier design can beadapted for any frequency in the400 to 1600 MHz range. Scaling theinput inductor (L1) will allowoperation on a different frequency.The graph shown in Figure 6 givessome idea of the relationship of L1vs. frequency. Source feedbackshould be adjusted accordingly.The ATF-10236 has been usedsuccessfully in circuits operating atas low as 150 MHz with similarresults.

GaAs FET DemonstrationBoardA demonstration board built on0.031 inch FR-4/G-10 is shown inFigures 7 and 8. The board asshown can be set up with either

passive or active biasing. If desiredan additional gain stage in the formof a silicon MMIC such as theHewlett-Packard MSA series canbe used for additional gain at aslight increase in overall cascadenoise figure. A later improvedversion of the artwork has thelocations of R1 and L2 reversed.The mounting pad for R1 and L2has an adverse effect on noisefigure. It is therefore better to havethe RF choke first then followedby the resistor since the imped-ance of the choke is higher thanthat of the resistor.

An additional resistor R10 can beadded in series with capacitor C2in order to improve stability andoutput return loss of the firststage. As an example, the use of a16 Ω resistor at R10 will decreasegain by about 1 dB while improv-ing output match and stability. Itmay also help when cascading theFET stage with the MMIC oranother device.

The measured loss of the inputmicrostripline excluding theblocking capacitor C1 is 0.3 dB at1600 MHz and 0.15 dB at 900 MHz.Keep this loss in mind whenevaluating the devices as mostcomputer simulations appear to beoptimistic about dielectric boardlosses and therefore give anoptimistic (lower) prediction ofnoise figure.

30

20

10

0

-10

-20

-30

100 500 900 1300

+16.45 dB+895.00 MHz

1700 2100FREQUENCY (MHz)

GA

IN (

dB

)

Figure 4. Typical swept gainperformance of 900 MHz amplifier.

Figure 6. Inductance L1 vs. optimumfrequency.

150

100

50

00.50.1 1.0 1.5 2.0

FREQUENCY, GHz

IND

UC

TA

NC

E, n

H

30

20

10

0

-10

-20

-30

100 500 900 1300

+14.31 dB+1.5700 MHz

1700 2100FREQUENCY (MHz)

GA

IN (

dB

)

Figure 5. Typical swept gainperformance of 1575 MHz amplifier.

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Figure 7. GaAs FET Demo board showing component placement. For bestpeformance reverse the location of R1 and L2 (see text).

Figure 8. GaAs FET Demo board 1X artwork.

In Case Of DifficultyGenerally the noise figure shouldbe within a couple or three tenthsof a dB of that indicated in theperformance table and gain withina few dB. If the noise figure orgain is not as expected then checkthe bias conditions. Is the Vdsbetween 1 and 2 volts and draincurrent 20 to 25 mA? If the biasvoltage and current is adjustable,does performance maximize at therated bias conditions or at someother set of values? If so, then theproblem may be excessive sourceinductance which is causing a highfrequency oscillation. If a device isoscillating at any frequency, evenan out-of-the-band frequency, it

may be difficult to obtain ratedperformance. Shorten the sourcelead length or find a source bypasscapacitor with lower parasiticinductance. The design assumed0.4 nH of associated inductance forthe 1000 pF source bypass capaci-tors. If the problem is inbandstability, then the solution is to addseries resistance between the drainand output connector. This inseries with blocking capacitor C2.Additional gain reduction can alsobe had by decreasing the length ofthe shunt output inductor betweenR3 and C7. In some cases, higherthan expected noise figure can beattributed to noise from the dc todc converter or the PNP transistor

used for active biasing. Additionaluse of 0.1 µF bypass capacitors atC4 and C8 may be necessary. Theuse of a lossy or low Q RF chokefor L2 can contribute to an in-crease in noise figure. The use ofsmall molded RF chokes for L2works well in prototypes. Forsurface mount applications be sureto choose a high Q wire-woundchoke such as those made byCoilCraft.

In some instances, the enclosurecan cause undesirable feedbackacross the circuit board which cancause instabilities. This phenom-ena is true of any amplifier design.A cross sectional view of thehousing can be viewed as a pieceof waveguide whose dimensions,both width and height, determinethe band of frequencies that it maypass with minimal attenuation. Acombination of the amplifierresponse along with the housingresponse could contribute toinstabilities if not controlled. Theuse of low profile surface mountcomponents will minimize thiseffect. Making sure that the coveris no closer to the printed circuitboard than is necesary will mini-mize coupling from the cover. It ispreferred to have the cover at least0.3 to 0.5 inches above the circuitboard. RF Absorptive materialsuch as ECOSORB™ can be usedon the cover to minimize reflec-tions if the cover has to be in closeproximity to the board. The use ofa metal divider hanging down fromthe cover is also another method ofbreaking up enclosure effects.

.031 FR-4 GaAsFET DEMO BD

INPUT C1

C4

C6 R3

C7C8R2L2

R1

C5C2 C3

C9

R9R7

U2

R8R6

C12

R5

U1C10

C13

C11

R4

U3 OUTPUT

Vdd

2.1"

.031 FR-4 GaAsFET DEMO BD

INPUT C1C5

L1

2.1"

C2 OUTPUT

R3

C9

R4

U3 C3

C8C7

U1

U2

R7

R6

C12

C13

R8

R9

L2

C4

R5

C10

C11

R1 C6R2 Vdd

5-13

Amplifier TuningDue to the inherent broad band-width of these amplifiers, theyshould require no RF tuning inproduction. With the use of activebiasing, the ampifiers shouldpower up at the proper bias pointwithout any further adjustmentrequired. The frequency at whichminimum input VSWR will occurcorresponds automatically withminimum noise figure and maxi-mum gain. As shown in theforegoing data, the noise figurevaries very little over a widebandwidth, so it might be advanta-geous to tune for minimum inputVSWR as opposed to noise figure.Without the source inductance, theinput VSWR will be considerablyhigher.

Since the noise matching networkis low Q it does offer broadbandwidth and insensitivity totolerance on the series inductor.According to the computer simula-tion, varying the input inductor L1from its nominal value of 7 nH to alow of 5 nH to a high of 10 nHincreases the noise figure less than0.2 dB. To minimize cost, thelumped inductor can be replacedby a microstripline .010 incheswide by 0.28 inches long at theexpense of 0.2 to 0.3 dB increase innoise figure. See computersimulation in the Appendix.

The simple series L/R matchingnetwork in the output circuitforces a good broadband lowVSWR output match. Due to thefinite amount of reverse isolationof the device, the output match isaffected by the input match andvice versa. Therefore the fre-quency of best output VSWR issomewhat dependent on where theinput network is optimum.

Power OutputThe use of heavy resistive loadingin the output circuit to obtainbroadband stability can be ex-pected to limit the power outputcapability of the amplifier. Despitethe resistive loading the 900 MHzamplifier has a measured 1 dB gaincompression point of 5 mW whenbiased at a Vds of 1 volt and Ids of20 mA.

Operation At ReducedCurrentFor applications that are morecurrent conscious, the 900 MHzamplifier was tested at various

drain currents from 1.5 mA to 25mA with a constant Vds of 1 volt.The graph shown in Figure 9reflects typical performance atreduced drain current withoutreadjusting either the input oroutput impedance match. The SParameters shown in Tables 4, 5,and 6 can be used to help design anamplifier for a specific low currentapplication. The noise parametersat the nominal bias (Table 1) willprovide a good starting point for adesign at lower current. The finaldesign is best optimized on thebench.

Freq S11 S21 S12 S22

GHz Mag Ang Mag Ang Mag Ang Mag Ang

0.1 .99 -4 3.66 177 .006 83 .35 -20.5 .99 -17 3.62 164 .031 80 .35 -131.0 .98 -36 3.52 147 .060 66 .34 -282.0 .92 -69 3.25 120 .117 44 .30 -60

Table 4. Scattering and Noise Parameters for the Hewlett-PackardATF-10236 GaAs FET, Vds = 1 volt and Ids = 10 mA



0.1 .99 -3 2.67 178 .004 98 .50 -20.5 .99 -15 2.66 165 .033 76 .50 -121.0 .99 -32 2.63 149 .068 68 .49 -252.0 .95 -62 2.49 123 .128 44 .44 -50




0.1 .999 -3 1.60 178 .009 76 .68 -30.5 .999 -13 1.60 167 .036 78 .67 -111.0 .99 -28 1.61 151 .075 69 .66 -222.0 .97 -55 1.56 126 .143 48 .62 -43


5-14

Other ApplicationsThe basic ATF-10236 amplifiercircuit can also be successfullyused as a mixer for eitherupconverting or downconvertingan input signal. As adownconverter, use the RF inputas the input port and use the RFoutput port as the local oscillatorinput port. Only a few milliwattsof LO is required at this port. Thisis termed a drain pumped mixer.The IF can be taken off of the drainbias decoupling line. The drainbias decoupling line should bebypassed with a fairly low value ofcapacitance such that the normallylow frequency of the IF can stillpass through and be coupled out tothe IF port through a low passmatching network. Mixer design iscovered in more detail in applica-tion note AN-G005.

ConclusionThe results show that high-frequency GaAs FETs can be usedvery successfully in the VHFthrough 1700 MHz frequency rangewith very simple circuit tech-niques. Noise figures of 0.5 dB at900 MHz and 0.73 dB at 1575 MHzare possible using the inexpensiveATF-10236 GaAs FET device on aninexpensive epoxy glass dielectricprinted circuit board.

The single-element input matchingnetwork provides very goodperformance in this frequencyrange and offers the greatestbandwidth and least sensitivity tomanufacturing tolerances. Theresistive loading in the outputnetwork provides the best broad-band stable performance bysacrificing some in-band gain. Thecomputer simulation programs areinstrumental in analyzing overallamplifier performance.

References1. L. Besser, “Stability Consider-ations of Low Noise TransistorAmplifiers With SimultaneousNoise and Power Match,” LowNoise Microwave Transistors andAmplifiers, H. Fukui, Editor, IEEEPress, 1981, pp. 272-274.2. G.D. Vendelin, “FeedbackEffects on the Noise Performanceof GaAs MESFETs,” Low NoiseMicrowave Transistors and Ampli-fiers, H. Fukui, Editor, IEEE Press,1981, pp.. 294-296.

3. D. Williams, W. Lum, S. Weinreb,“L-Band Cryogenically CooledGaAs FET Amplifiers,” MicrowaveJournal, October 1980, p. 73.

4. G. Gonzalez, Microwave Transis-tor Amplifiers, Prentice- Hall, 1984,pp. 131-133.

5. G.D. Vendelin, Design of Amplifi-ers and Oscillators by the S-Parameter Method, Wiley, 1982, pp.53-59.

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Figure 9. Noise figure and gainperformance of the 900 MHz LNA at Vds= 1 volt and various drain currents.

Using the ATF-10236 in Low Noise Amplifier Applications inÊthe UHF through 1.7 GHz Frequency...

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Transcript of Using the ATF-10236 in Low Noise Amplifier Applications inÊthe UHF through 1.7 GHz Frequency...