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TELEVISIONSERVICING

VOLUME 3

Vision Interference Limiters

Vision Automatic Gain Control

Flywheel Synchronizing Circuits

Television Camera & Studio Equipment

G. N. PATCHETTB.Sc. (Eng.), Ph.D., C. Eng., F.I.E.E., F.I.E.R.E., M.I.E.E.E.

Fellow of the Royal Television Society

The theory of television and the servicing of receivers (monochrome)

are covered by this series of four volumes. The series is concerned

mainly with modern receivers but also deals with television systems

and studio equipment as used in this country.

This new edition of the third volume has been enlarged to include

information on the expanding use of transistors in receivers.

Students of television servicing and those requiring a general

knowledge of television will find this a valuable series.

LONDON: NORMAN PRICE (PUBLISHERS) LTD

55p net

TELEVISIONSERVICING

Volume 3

1

G. N. PATCHETTB.Sc. (Eng.), Ph.D., C. Eng., F.I.E.E., F.I.E.R.E., M.I.E.E.E.

Fellow of the Royal Television Society

LONDONNORMAN PRICE (PUBLISHERS) LTD

NORMAN PRICE (PUBLISHERS) LTD.17 TOTTENHAM COURT ROAD, LONDON, W.l.

© NORMAN PRICE PUBLISHERS LTD., 1958

2nd edition 1964

3rd edition 1970

2nd impression 1971

Printed in Great Britain byA. Brown & Sons, Ltd., Hull

CONTENTS

1 VISION INTERFERENCE LIMITERS page 1

2 VISION AUTOMATIC GAIN CONTROL 11

3 FLYWHEEL SYNCHRONIZING CIRCUITS 32

4 TELEVISION CAMERAS AND STUDIO EQUIP-

MENT 47

INDEX 65

ILLUSTRATIONS

FIG. PAGE

1.1 Effect of Interference on Carrier 1

1.2(a) Manually Adjusted Vision Interference Limiter 2

(b) Manually Adjusted Vision Interference Limiter 3

1.3 Manually Adjusted Vision Interference Limiter when GridModulation is used 4

1.4 Manually Adjusted Vision Interference Limiter using Negative

Feedback 41 .5 Effect of Changing Contrast of Picture on Operation of Limiter 41.6(a) Circuit of Self-adjusting Vision Interference Limiter 5

(b) Operation of Circuit with no Interference

(c) Operation of Circuit with Interference Pulses

1.7 Self-adjusting Vision Interference Limiter for CathodeModulation 6

1.8 Self-adjusting Vision Interference Limiter using Negative

Feedback 6

1 .9 Characteristic of Pentode with various Suppressor Grid Voltages 71.10 Circuit of Black Spotter 8

1.11 Circuit of Black Spotter using Amplifier 9

1.12 Circuit of Black Spotter using two Load Resistors in AnodeCircuit of Video Amplifier 9

2.1 Sound Modulation of Carrier 11

2.2 Picture Modulation of Carrier 12

2.3(a) D.C. Component of Signal on Grid of Synchronizing Separator 13

(b) Circuit of a.g.c. System using Synchronizing Separator

2.4 Use of Delay Diode 14

2.5 Synchronizing Pulse-cancelled a.g.c. Circuit 15

2.6 Improved Synchronizing Pulse cancelled a.g.c. Circuit (Mullard) 16

2.7(a) Field Pulses during Field Flyback Time 18

(b) Line Pulses showing Back Porch2.8 Gated a.g.c. System 18

2.9 Gated a.g.c. System 19

2.10 Waveforms 202.11 Gated a.g.c. System 21

2.12 Effect of Changing Contrast on Voltage of Back Porch 21

2.13 Gated a.g.c. System 22

2.14 Waveforms of Circuit shown in Fig. 2.13 23

2.15 Negative Modulation 24

2.16 Simple Peak Detector 24

2.17 Limiter Circuit 25

2.18 Use of Demodulator for a.g.c. during Warming-up Time 25

2.19 Synchronizing Pulse-cancelled Circuit for Negative Modulation 26

2.20 Effect of Signal Amplitude on Black Level Voltage (NegativeModulation) 27

2.21 Transistor Characteristic for a.g.c. Control 28

2.22 High Level Contrast Control 29

2.23 Transistor a.g.c. Circuit 30

2.24 Automatic Contrast Control Circuits 31

2.25 Effect of Increase of Video Signal on Brilliance 32

3.1 Video Waveform showing Noise 33

3.2 Effect of Interference on Vertical Line 33

ILLUSTRATIONS

no. PAGE

3.3 Principle of Flywheel Synchronizing Circuit 343.4 Flywheel Synchronizing Circuit, Oscillator section (Ferguson

991T) 353.5 Flywheel Synchronizing Circuit, Reactance valve section

(Ferguson 991T) 35

3.6 Flywheel Synchronizing Circuit, Phase detector section

(Ferguson 991T) 363.7 Operation of Phase Detector 373.8 Operation of Phase Detector when there is a Phase Shift 373.9(a) Principle of Phase Detector 38

(b) Flywheel Synchronizing Circuit (Mullard) 393.10 Operation of Circuit shown in Fig. 3.9(b) 393.11 Phase Detector using Diodes 403.12 Operation of Circuit shown in Fig. 3.11 403.13 Flywheel Synchronizing Circuit using Diodes 41

3.14 Diode Phase Comparator 423.15 Operation of Phase Comparator 43

(a) Initial Charging Current

(b) Condition just before Line Pulse

(c) Condition during Pulse

(d) Condition during Pulse with point C at +5V 44(e) Condition during Pulse with point C at —5V

3.16 Modified Diode Comparator Circuit 453.17 Operation of Circuit of Figure 3.16 464.1 Flying Spot Scanning System 474.2(a) Iconoscope Camera Tube 49

(b) Mosaic 49(c) Connection of Mosaic and Signal Plate 50

4.3 Image Iconoscope 524.4 Strip Electrodes to Reduce Shading 53

4.5(a) Secondary Emission Ratio 54(b) Bombardment of Insulated Plate P

4.6 Orthicon Camera Tube 564.7 C.P.S. Emitron Camera Tube 574.8 Principle of Electron Multiplier 584.9 Image Orthicon Camera Tube 584.10 Photo-Conductive Type of Camera Tube 604.11 Block Diagram of Camera Chain 62

CHAPTER 1

VISION INTERFERENCE LIMITERS

The effect of interference on a television picture depends on the system of

modulation used. In the 405-line system where positive modulation is

used white spots are produced on the screen. In the 625-line system

(and the 525-line American system) where negative modulation is used, black

spots are produced on the screen. The reason for this is shown in figure 1.1.

CARRIERAMPLITUDE «nuIKI it ®

CARRIERAMPLITUDE Q

INTERFERENCEPULSES

FIG. 1.1. EFFECT OF INTERFERENCE ON CARRIER(a) Positive Modulation (b) Negative Modulation

In positive modulation [figure 1.1(a) ] the effect of interference is to increase

the magnitude of the carrier. It is possible for the interference to be of such afrequency and magnitude that it will cancel the carrier or reduce it, but the

chances of this happening are remote and can be ignored. Usually, the

interference is much larger than the magnitude of the carrier corresponding to

peak white, and results in a large peak as shown. Since this is in the direction

of white the interference produces white spots. In negative modulation as

shown in figure 1.1(b), again the effect of interference is to increase the

magnitude of the carrier but now in the black direction with the result that

black dots are produced. It will be seen that this interference causes pulses in

the synchronizing pulse region which may cause faulty synchronizing. Theeffect of the white spots is much more objectionable than the black spots.

This is an advantage of the negative modulation system; but with negative

modulation synchronizing is much more likely to be affected by interference

than with positive modulation.

It might at first be thought that the white spots would be no moreobjectionable than black spots (since the picture is commonly about 50 per cent

2 TELEVISION SERVICING

white and 50 per cent black) and this would be true if the spots were the samesize in each case. Unfortunately, the effect of interference with positive

modulation is to produce large white blobs instead of small bright spots.

This is due to the fact that the interference pulses heavily overload the cathoderay tube with the result that it goes out of focus resulting in large whitedefocused spots. With negative modulation the "overloading" is in thereverse direction and results in the cathode ray tube voltage going into theblacker than black region, i.e. synchronizing pulse region, but as regards thepicture modulation this can only result in a small black spot.

We will first consider the 405-line system using positive modulation.There are two general methods of dealing with interference:

1. To use a limiter which prevents the grid or the cathode of the cathoderay tube from being driven appreciably beyond a voltage corresponding to

peak white.

2. To use a device which inverts the pulses of interference and hencecauses black spots in place of white ones. This is generally termed a blackspotter.

(1) LIMITER SYSTEMSThese can be divided into two classes:

(0 Those requiring manual adjustment.

(h) Those with automatic adjustment.

(Q Manually Adjusted Limiters

One arrangement is shown in figure 1.2(a) where cathode modulation ofthe cathode ray tube is shown, which is the normal method used.

NEGATIVE -GOINGVIDEO SIGNALFROM ANODE OFVIDEO AMPLIFIER

FIG. 1.2(a). MANUALLY ADJUSTED VISION INTERFERENCE LIMITER

The anode of the limiter valve Vx is returned to the potential divider Rt

which supplies a voltage V, that is slightly less than the voltage correspondingto peak white. Thus, on a normal signal, the cathode of Vx is positive withrespect to the anode and Vx is therefore non-conducting and has no effect.

When an interference pulse arrives the cathode will tend to be driven to avoltage less than V. Hence Vx conducts and, due to its low forward resistance,prevents the cathode potential of the cathode ray tube dropping appreciablybelow the voltage V. Thus, the interference pulses are clipped. The capacitorCi (about 0-1 fiF) is used to produce a low impedance path from the anodeof Vi to chassis. By increasing the voltage V clipping will take place earlierand eventually the peak white portion of the signal will be cupped and detailsin the high-lights of the picture will be missing. If, on the other hand, V is

made too small the effect of the limiter is reduced and the interference pulseswill drive the cathode ray tube beyond peak white. The action of the circuit


is not perfect owing to the finite resistance of Vx in the conducting direction,

i.e. it does not art as a perfect short.

A variation of this type of limiter is shown in figure 1.2(b). The valve

Vi is made normally non-conducting by the voltage V fed through R2 from

NEGATIVE-GOINGVIDEO SIGNALFROM ANODE OFVIDEO AMPLIFIER

VALVE VI CONDUCTS WHENCATHODE BECOMES NEGATIVE

FIG. 1.2(b). MANUALLY ADJUSTED VISION INTERFERENCE LIMITER

the potential divider Ru If valve Vx does not conduct, the voltage across Vt

will be the voltage V from the potential divider Rx together with the video

signal. Since capacitor Ct will remove any d.c. component of the signal, the

waveform will be as shown in figure 1.2(b) with the mean value of the video

waveform at a voltage V. When the negative value of the signal (i.e. in the

white direction and below the mean value) is greater than Kthen valve Vx will

conduct and act as a short circuit (through Cx which has a value of about

0-1/iF) which prevents the cathode voltage of the c.r.t. going more negative.

Thus, by suitable adjustment of the voltage V the diode V% can be made to

conduct when the interference signal exceeds the peak white value.

When grid modulation is used a limiter circuit as shown overleaf in figure

1.3 may be used. In this case the diode Vx is reversed and the voltage Vmust be slightly greater than the voltage corresponding to peak white. Theaction is essentially similar to that of figure 1.2(a) Decreasing voltage Knowincreases the limiting and the high-lights of the picture will be clipped if it

is reduced too far.

Figure 1.4 shows an arrangement using negative feedback. The circuit

shown is for cathode modulation but can easily be altered for grid modulation.

As previously, the potential divider Rx is set to a voltage V which is slightly

less than that corresponding to peak white. Hence, Vx is non-conducting on

a normal signal. When an interference pulse arrives at the anode of the video

amplifier V2, the cathode of Vx drops below its anode potential and the pulse

is transmitted through Vx and Cx to the grid of V2 . Thus, 100 per cent, negative

feedback is applied and the pulse is greatly reduced in value due to the low

gain of the stage when negative feedback occurs.

The disadvantage of the manually adjusted limiter is that the adjustment

depends on the contrast setting. This is shown in figure 1.5. With a low

TELEVISION SERVICINGPOSITIVE-GOING VIDEOSIGNAL FROM ANODEOF VIDEO AMPLIFIER

CRT. HT.-t-

R VI

Rl

CI

INCREASELIMITING

FIG. 1.3. MANUALLY ADJUSTED VISION INTERFERENCE LIMITER WHEN GRIDMODULATION IS USED

Rl

*">

- H.T.+

R2VI

CI

©-4

POSITIVE -GOINGVIDEO SIGNALFROM DEMODULATOR

TO CATHODE OF CRT.

FIG. 1.4. MANUALLY ADJUSTED VISION INTERFERENCE LIMITER USINGNEGATIVE FEEDBACK

VOLTAGEWITH

NO SIGNAL(CONSTANT) 7 LOW CONTRAST

©VOLTAGEWITH

NO SIGNAL(CONSTANT)

7

. LIMITERSETTING

LARGERCONTRAST

These portions will be clipped.

£. LIMITER

SETTING ©FIG. 1.5. EFFECT OF CHANGING CONTRAST OF PICTURE ON OPERATION OF LIMITER

contrast setting the limiter may be set correctly as at (a). If the contrast is

now increased (by adjustment of the contrast control of the receiver) and no

VISION INTERFERENCE LIMITERS 5

alteration is made to the setting of the limiter it will cut off the peak whites

of the picture as shown at (b). If, on the other hand, it is set correctly for (b)

and the contrast is then reduced the limiter will not be effective. For correct

operation, therefore, both limiter and contrast controls require simultaneous

adjustment. In practice this probably means that the limiter is not adjusted

to give the best results.

(;°i) Automatic Limiters

A circuit of this type of limiter is shown in figure 1.6 which is suitable

where grid modulation of the cathode ray tube is used. This circuit is

POSITIVE -COINS.VIDEO SIGNALFROM ANODE OFVIDEO AMPLIFIER

VOLTAGE ACROSS CI

©These portions

removedVOLTAGE ACROSS CI

©FIG. 1.6 (a) CIRCUIT OF SELF-ADJUSTING VISION INTERFERENCE LIMITER

(b) OPERATION OF CIRCUIT WITH NO INTERFERENCE(c) OPERATION OF CIRCUIT WITH INTERFERENCE PULSES

described first as it is rather easier to understand than that for cathode

modulation. In the absence of Ru Ct would charge through Vx to the maxi-

mum voltage appearing on the grid of the cathode ray tube. That is, in the

absence of interference, to peak white and would remain charged to this

voltage since CV cannot discharge through Vx . If a resistor R x of high value is

added across valve Vx then Ct will discharge slightly between peak white

signals. This is shown at (b) but, if the time constant Cx .Ri. is suitable, the

drop of voltage between peak white signals may be neglected. If an

interference pulse occurs it will tend to charge Ct as shown at (c) but, since

the pulse is of short duration, it will not increase its voltage appreciably and

the pulse will be largely removed. In other words, for sudden changes of

voltage above peak white Cx acts more or less as a short across the grid and

cathode of the cathode ray tube. If a large number of pulses occurred there

would be a tendency for d to charge, particularly if Rx was of high value;

this effect can be reduced by decreasing its value. By suitable choice of Ri.Ct

the circuit effectively prevents pulses from overloading the cathode ray tube.

The disadvantage of the circuit is that, in order to keep Cj charged, a certain

amount of clipping must take place on peak white since C, must be kept

charged to the peak white value as shown in figure 1.6(b). This is particularly

noticeable if the amount of peak white in the picture is small (this is then

similar to the interference pulses) and results in loss of detail in the high-lights.

TELEVISION SERVICING

The seriousness of this effect depends on the value of Rt . Thus, a low valueof Rx results in improved interference suppression but a greater tendency tolose detail in the high-lights of the picture. In many cases Rt is made variable

(sometimes in steps) and can then be set to suit the reception conditions in

which the set is being used. In other words, under conditions of severe

interference a certain amount of loss of high-light detail will be tolerated if theinterference is well suppressed.

The corresponding circuit for cathode modulation is shown in figure 1.7.

NEGATIVE-GOING.VIDEO SIGNALFROM ANODE OF

Rl

(IMO)

V *_C '(CMpF)

FIG. 1.7. SELF-ADJUSTING VISION INTERFERENCE L1MITER FORCATHODE MODULATION

The operation is now similar but really reversed. Q tends to charge throughRi (in the absence of Vt it would charge to the mean value of the voltage fedto the cathode of the cathode ray tube) but is discharged by Vx on peak whitesignals so that the voltage V across Cx is approximately the voltage corre-sponding to peak white. As before, if an interference pulse (now negative-going) arrives it tends to discharge Cu but the effect is mainly to remove thepulse. Of course, the circuit as that of figure 1.6 tends to degrade thehigh-lights of the picture. The position of Vu Ri and C, may be interchangedin this circuit without affecting its operation.

The same general idea may be used with the negative feedback circuit andthis is shown in figure 1.8. The action is as figure 1.7 except that, whencurrent flows in Vi the current flows in R2 placed in the grid circuit of the video

NEGATIVE -GOINGVIDEO SIGNAL

POSITIVE -GOINGVIDEO SIGNALFROM DEMODULATOR

FIG. 1.8. SELF-ADJUSTING VISION INTERFERENCE LIMITER USING NEGATIVEFEEDBACK

amplifier. This results in a large amount of negative feedback and the gainof V2 is greatly reduced.

All the foregoing circuits, with the exception of those using negativefeedback, may be placed in the grid circuit of the video amplifier instead of

VISION INTERFERENCE LIMITERS 7

in the feed to the cathode ray tube. This arrangement, however, is notcommon.

Another method of limiting the video signal is by applying a negative

d.c. voltage to the suppressor grid of the video amplifier. A suitable negative

voltage may be obtained from the line output stage. The effect of varying

INCREASING

FIG. 1.9. CHARACTERISTIC OF PENTODE WITHVARIOUS SUPPRESSOR GRID VOLTAGES

the suppressor grid voltage on the I<r-Vgi characteristic is shown in figure 1.9

where it will be seen that when a large negative voltage is applied to G3 thecharacteristic bends over. Hence, if peak white is made to correspond to

grid voltage A any pulse making the grid less negative than this will onlyincrease the anode current by a small amount. The increase in anode current

caused by the pulse can be decreased by the application of a larger negative

voltage to G3. Thus the voltage applied to G3 is made variable so that it canbe adjusted to suit local conditions.

(2) BLACK SPOTTERSThe idea of the black spotter is that if an interference pulse occurs which is

larger than peak white it will be reversed in phase and so appear as a smallblack dot on the screen in place of a white blob. The dot will be small as, ofcourse, no defocusing can occur even though the pulse goes beyond the blacklevel (i.e. blacker than black). The effect of the interference is greatly reduced,

partly due to the fact that a black dot is less noticeable than a white one, andbecause no defocusing can occur. The device must be set correctly as in the

case of the manually controlled limiters or otherwise, if the setting is too low,

the high-lights of the picture will also be reversed and appear black instead

of white. The effect of this is much more objectionable than the clipping ofthe high-lights in a limiter.

There are two general types of black spotter:

(0 The interference pulse is reversed and fed with greater amplitude to

the same electrode as that used for the signal.

(h) The pulse is kept at the same polarity but fed with larger amplitude to

the electrode of the cathode ray tube other than that to which the signal

voltage is applied, i.e. the grid when cathode modulation is used.

Method (i).—A circuit using this principle is shown in figure 1.10. The signal

from the demodulator is fed to the video amplifier through Rx and also to the

valve V\ where it is amplified and reversed in polarity. The diode V2 is set

by R3 so that it does not conduct until the voltage on the anode of Kx falls

below that corresponding to peak white. When there is no interference V2

V2 fs a GermaniumDiode /O

TELEVISION SERVICING

FROMDEMODULATOR

H.T.+

VIDEO AMPLIFIER

AA

FIG. 1.10. CIRCUIT OF BLACK SPOTTER

does not conduct and the circuit operates normally, the signal being fed tothe video amplifier through Ru When an interference pulse occurs V2 conductsand feeds the pulse (of opposite polarity owing to the phase reversal in Kt)

to the grid of the video amplifier so neutralizing the positive pulse fed throughRt . The amount of neutralizing will depend on the gain of Vx and the circuit

may be arranged to produce grey or black spots.

Method (ii).—This arrangement appears to be the more common and onecircuit is given in figure 1.11. The cathode ray tube is cathode modulated fromthe video amplifier valve Vx . The grid is returned through R2 to the brilliance

control Rt . The output of the video amplifier is also fed to the cathode ofVz and the grid of this valve is connected to the limiter control Rt . If Rt is

set correctly V2 will not conduct on a normal signal as its cathode will bepositive with respect to the grid. On interference pulses the cathode of V2 is

driven negative with respect to the grid and the valve conducts, producinglarge negative pulses at the anode. The valve acts like a grounded gridamplifier and there is no phase reversal between input and output. This is

due to the fact that a negative-going voltage on the cathode causes an increasein anode current and so a drop in anode potential. The negative pulses at

the anode are fed to the grid of the cathode ray tube through Ct . Since thesepulses are greater than those applied to the cathode (owing to the amplifica-tion of valve V2), the grid is driven negative with respect to the cathode andblack spots will result. It is, of course, important that the limiter controlJ?i be set correctly. If the grid voltage is too large the tube will conductaround peak white and produces a reversed picture (i.e. a negative picture)on the high-light portions which is most objectionable. If the voltage fed tothe grid is too small then the effectiveness of the circuit will be reduced.

Another arrangement is shown in figure 1.12. The normal signal is

developed across the resistor R2 by the video amplifier valve Vt and this is

fed to the cathode of the cathode ray tube. The grid is returned to thebrilliance control R5 through resistor R6 . A similar but larger signal voltage


- H.T.+

BRILLIANCECONTROL

IncreaseBrilliance

FIG. 1.11. CIRCUIT OF BLACK SPOTTER USING AMPLIFIER

»(R2

Rl

6-8K w•H.T.+

Kb: IOOK

*-IH

(O

CRT.

V2

:R320 K

TIT

CIO-luF BRiaiANCE

CONTROL

R7SPOTTER CONTROL

Increase

R5

Increase

FIG. 1.12. CIRCUIT OF BLACK SPOTTER USING TWO LOAD RESISTORS IN ANODECIRCUIT OF VIDEO AMPLIFIER

is developed on the anode of the video amplifier. This is the voltage acrossboth resistor Rt and resistor R2 . The valve V2 is biased by the spotter controlR7 so that the valve only conducts when the anode of K, goes below thatcorresponding to peak white, as it does on interference pulses. Thus, V2 onlyconducts on interference pulses and these are fed to the grid of the cathode


ray tube through d. Since the magnitude of the pulses fed to the grid is

larger than that fed to the cathode, black spots will occur instead of white.

As in the other circuit it is important that the resistor R7 is set correctly.

As in the manually controlled limiters the setting of the black spotter control

will vary with the contrast setting of the receiver.

It should be noted that both the limiters and the black spotters only

operate on interference pulses which are larger than the signal: this is not so

in the case of the sound interference suppressor described in Volume 1.

Black spotters are no longer used, and even limiters do not seem to be

used much. This may be due to the fact that there are fewer fringe areas

now than there used to be.

When considering the 625-line system it has been seen that the effect of

interference is to produce black spots and hence no type of limiter is required.

On switchable sets the limiter used on 405 lines may be left in circuit or maybe switched out of circuit. If it is of the automatic type, switching out of

circuit is preferable so that no degradation of the picture occurs on peak

whites when on 625 lines.

CHAPTER 2

VISION AUTOMATIC GAIN CONTROL

Automatic gain control applied to the sound section of the receiver wasconsidered in Volume 1 when the basic arrangements are the same as in

ka broadcast receiver. The application of automatic gain control to the

vision section of the receiver is more complex, but first let us see what factors

make it desirable to introduce automatic gain control. In the case of receivers

designed for Band I only, a.g.c. was added to reduce the effect of fading in

fringe areas, and to some extent to reduce the effect of aeroplane flutter. In

areas of good signal strength, and no aeroplane flutter, there is little point in

complicating the receiver with an a.g.c. circuit, as the signal will not normally

vary in strength, only one programme being available. With the introduction

of Band III, which means the reception of two stations with quite different

signal levels, a.g.c. has become important. Although manual gain control

could be fitted, serious overloading would result when changing from one

station to another, until suitable adjustments to the gain control had been

made. This may be overcome by using preset sensitivity controls, one for

Band I and one for Band III, but this becomes difficult when more than twostations are to be received. Accordingly, a.g.c. of some type is always fitted

to modern receivers.

Automatic gain control circuits vary considerably, and basically there are

two types : mean level; and gated systems. Both systems are described although

mean level is used in most sets. Complications occur on a dual-standard

receiver dealing with both positive and negative modulation. With the intro-

duction of transistors in the tuner units and i.f. amplifiers considerable changes

and complications have occurred in a.g.c. circuits although the principles

remain the same. Valve circuits will be described first, starting with a valve

circuit for operation of a 405-line system using, of course, positive modulation.

VALVE CIRCUITSIf we examine an unmodulated carrier and one modulated with a sound

signal as in figure 2.1, we will see that the mean value of the carrier is the same,

independent of the percentage modulation or the waveform of the modulation.

This is because the average increase during one half-cycle is equal to the

VOLTAGE

UNMODULATED MODULATED

FIG. 2.1.

SOUND MODULATIONOF CARRIER

average decrease over the other half-cycle, i.e. the modulating signal does not

contain a d.c. component. Hence, a measurement of the mean value of

the carrier is a measure of the signal level fed to the demodulator, quite

independent of the modulation.

Let us now consider a carrier modulated with the vision signal using

positive modulation. In figure 2.2 is shown a carrier modulated with typical

11 b


VOLTAGE

VOLTAGE

MEAN VALUE

MEAN VALUE

BRIGHT PICTURE

FIG. 2.2. PICTURE MODULATION OF CARRIER

vision waveforms, that at (a) being for a dark picture and that at (b) being fora picture which is mainly white. It will be seen that the mean value of thecarrier is different in the two cases. The two diagrams are, of course, for thesame signal value at the aerial, since the magnitude of the synchronizing pulsesis the same in both cases. For this reason it is not correct to use the meanvalue of the carrier as this depends not only on the magnitude of the r.f.

signal fed to the demodulator, but also on the nature of the signal, i.e. on themodulation of the r.f. signal. Although the use of the mean value of thecarrier or, what is approximately the same thing, the mean value of the visionsignal from the demodulator, is not correct the method is often used becauseof its simplicity.

The effect of mean level a.g.c. is to increase the gain with a picture whichis dark compared with one containing a lot of white. This is equivalent toincreasing the contrast when the picture is dark. Since we maintain thebottom of the synchronizing pulses at a constant value, increase of contrastalso increases brightness. Thus, when a dark picture occurs both the contrastand brightness are increased so that any black portions become grey. Thisis perhaps most obvious with captions on a black background. The effectis to make the background grey instead of black and a tendency to overloadon the whites. The effect is similar to the removal of the d.c. component ofthe signal. (See Volume 1). The system has the advantage of simplicity andtends to reduce aeroplane flutter.

For correct operation we should measure the magnitude of some portionof the signal which does not depend on the nature of the picture. Such aportion is the synchronizing pulse section and later we will see how a.g.c. maybe obtained by measurement of the magnitude of the synchronizing pulses.

First, let us consider the type of a.g.c. which operates on the mean valueof the carrier or vision signal. Although it would appear to be possible toobtain a suitable voltage from the normal diode demodulator for a.g.c, asused in sound, there are two difficulties:

VISION AUTOMATIC GAIN CONTROL 13

(0 The magnitude of the voltage is not large and the a.g.c. would not be

very effective unless the voltage were amplified in some way, and amplification

is difficult with valves.

(h) With cathode modulation of the cathode ray tube the output of the

demodulator is a positive video signal and hence a negative a.g.c. voltage

cannot be obtained from the diode. The diode cannot, of course, be reversed

as in sound: doing so would result in a negative picture. A separate diode

would, therefore, have to be used.

It will be seen from the outline of the carrier in figure 2.2 that it is identical

with the video or vision signal, hence we can use the mean value of the video

signal instead of the mean value of the r.f. carrier. The video output voltage

of the demodulator is of the same order as the carrier input and too small

for direct use in the a.g.c. circuit. But we have an identical signal of muchlarger amplitude on the anode of the video amplifier and it is this signal

which is commonly used. What we require is the mean, or d.c. component, of

the signal output from the video amplifier. This could be obtained by feeding

the signal from the video amplifier to a d.c. restorer but it is not necessary

to use a separate d.c. restorer as we have one available in the form of the

synchronizing separator. In Volume 2 the pentode type of synchronizing

separator was described and it was shown that the grid and cathode of this

valve act as a d.c. restorer. This means that the waveform now hangs from

the zero line as shown in figure 2.3(a). If this waveform is smoothed we

VOLTAGE

AAO-IjjF

-II-

SYNCHRONIZINGSEPARATOR

4-7M<liRl i'Mfl

"™ CI io^F ©A.G.C. LINE

FIG. 2.3(a) D.C. COMPONENT OF SIGNAL ON GRID OF SYNCHRONIZING SEPARATOR(b) CIRCUIT OF A.G.C. SYSTEM USING SYNCHRONIZING SEPARATOR

obtain the mean value, or the d.c. component, which will vary with the

magnitude of the signal (and, as already stated, with the nature of the picture).

Thus, all we need to do in order to obtain the a.g.c. voltage is to feed the

signal from the grid of the synchronizing separator through a filter circuit to

remove the vision signal, as in figure 2.3(b).

Having obtained the a.g.c. voltage we must now consider in what way it

should be applied to the valves of the receiver. It is common practice not to

apply a.g.c. to the final i.f. stage as this is most likely to result in overloading.


This is due to the fact that the final i.f. stage is required to give a relatively

large output to the demodulator and that it is easily overloaded owing to thelow anode impedance which has to be used in order to obtain the requiredbandwidth. This is different to the final i.f. stage in a broadcast receiver

where the anode load is much higher and it is easy to get the requireddemodulator voltage. Ideally, the operation of the a.g.c. should be as follows:

With a very small input signal the gain of the r.f. and i.f. amplifiers should beat a maximum and, as the signal input increases, the gain of the i.f. amplifier

should first be reduced and not that of the r.f. amplifier. The reason for this

is that a large signal must be maintained at the frequency-changer so as tomaintain a high signal-noise ratio and so prevent a "noisy" picture. Whenthe signal becomes large the gain of the r.f. amplifier stage must be reducedmore than that of the i.f. amplifier, or the frequency-changer stage will beoverloaded. Also, it is desirable that a delay be introduced into the a.g.c.

system, otherwise the sensitivity will be reduced, even by weak signals.

In order to get the best control between the r.f. and i.f. amplifiers alocal/distance switch or control may be fitted which commonly controls themagnitude of a.g.c. voltage fed to the tuner unit.

Until recently the contrast of the picture was controlled by varying thegain of the r.f./i.f. stages and therefore some means of varying the a.g.c.

voltage is required. This is usually obtained by varying the magnitude of thedelay voltage as described later.

In the a.g.c. system using the synchronizing separator it is obvious that

the delay cannot be obtained by alteration of the operation of the syn-chronizing separator, as this would prevent it from performing its mainfunction. Hence, the delay is normally obtained by the use of a diode, asexplained in connection with sound a.g.c. in Volume 1. The basic arrangementis shown in figure 2.4. Rt and d form the filter circuit from the synchronizingseparator (or any other source of a.g.c. voltage may be used in this circuit).

TO GRID OFSYNCHRONIZING

SEPARATOR

A.t,.(_. LINt 4.7 Mn

FIG. 2.4. USE OF DELAY DIODE

The a.g.c. line is also connected to a variable positive voltage from potentio-meter Ply through resistor R2 . Px varies the delay voltage and so forms thecontrast control. A diode Ki is connected between the a.g.c. line and chassis.

In the absence of a signal a current will flow through R2 from Plt but valve Kt

will prevent the a.g.c. line going positive, the current completing the circuit

through this valve. If the upper end of Rt is now at a negative potential apart of the current flowing in R2 will flow in Rt and the remainder in valve Vt .

So long as the current in Ri is less than that in R2 a current will flow in Vt

and the valve holds the a.g.c. line at chassis potential. When the upper endof Ri becomes sufficiently negative the current in JRt will tend to exceed thatin R2, so no current will flow in Vu and the a.g.c. line will now go negative byan amount depending on the negative voltage from the synchronizing separator


(applied to the upper end of RJ and the values of jRt and R2 . The delay

voltage is settled by the values of Rx and R2 and the positive voltage from the

potentiometer Pu If Ri and Rz are equal then the delay voltage is equal to

the voltage obtained from the potentiometer P^Instead of varying the delay voltage by using a potential divider Pu the

delay voltage may be fixed and the signal voltage from the grid of the

synchronizing separator varied. This is done by returning jRj to the slider of

a potential divider forming the grid resistor instead of direct to the grid of the

synchronizing separator.

Instead of obtaining the voltage from the grid of the synchronizing

separator, a separate diode d.c. restoring circuit has been used. In this case

the delay can be altered by varying the voltage fed to the cathode of the d.c

.

restoring diode.

Suitable control of the r.f. and i.f. stages may be obtained by feeding the

full voltage to the i.f. stage and a fraction of the a.g.c. voltage to the r.f.

stages. Alternatively, two delay circuits may be used, allowing the a.g.c.

voltage to act on the i.f. amplifier first and then on the r.f. amplifier whenthe signal is large.

Although this mean level type of a.g.c. is so common it is technically not

correct and a viewer does not see what is intended by the producer. However,it appears to be acceptable to the average viewer but a number of circuits

will be described which overcome the fault of mean level a.g.c.

One method of obtaining an a.g.c. voltage, which does not depend on the

content of video waveform, is known as the synchronizing pulse cancelled

circuit. In the simple basic circuit shown in figure 2.5, the video signal fromthe grid of the synchronizing separator valve is fed to the anode of the diode

Grid ofSynchronizing

Separator

C2

HI-Anode ofSynchronizing ^

Separator

RI-VWV-

R2-*Hfr-

VI

-Ok

"ir~rFIG. Z5. SYNCHRONIZING PULSE-CANCELLED A.G.C. CIRCUIT

valve Vi. The anode of this diode is also fed with line synchronizing pulses

of suitable amplitude, which are negative-going as shown. These pulses are

obtained from the anode of the synchronizing separator valve and, of course,

they are of constant amplitude independent of the signal input. When thetwo waveforms are added together the resultant waveform on the anode of Vx

is as shown. The waveform shown assumes that the line synchronizing pulses

from the synchronizing separator are greater than the synchronizing pulses

contained in the composite waveform on the grid of the synchronizing separ-

ator. If the synchronizing separator is operating normally this will, of course,

be the case. The magnitude of the synchronizing pulses from the synchronizing

separator is not important so long as under all conditions they are larger than


those on the grid of the synchronizing separator, so that in the resulting

waveform on the anode of K, the synchronizing pulses are inverted as shownin figure 2.5, i.e. the sychronizing pulses are cancelled and reversed.

It will be seen from this waveform that the minimum voltage (i.e. minimumnegative voltage) corresponds to black level and this voltage therefore is

proportional to the magnitude of the synchronizing pulses in the video signal,

i.e. to the signal being fed to the demodulator and video amplifier. If the

diode is connected as shown, with a suitable negative voltage supply to the

resistor R, then C will charge through R, but, when the voltage across C tends

to rise (in a negative direction) above the black level voltage the diode will

conduct and discharge C to this voltage. Thus, provided that the C-R product(time constant) is large compared with the time of a line, capacitor C will

become charged to an approximately steady voltage equal to the magnitudeof the synchronizing pulses and hence proportional to the signal level at the

demodulator. Thus the voltage across C can be used as the a.g.c. voltage.

Unfortunately, the magnitude of this voltage is not large because the

synchronizing pulses are only 30 per cent of the signal amplitude and someloss occurs in the resistor Rlm For this reason this a.g.c. system is not good in

this simple form and an amplifier is desirable between capacitor C and the

a.g.c. line. It should be noted that the operation of this circuit does not

depend on the line timebase. It will be seen later that some other systems

depend on a suitable pulse from the line timebase and hence will not workif the timebase is not correctly synchronized.

An elaboration of this principle (due to Milliard Ltd.) is shown in figure 2.6.

Valve Vi is fed with a video signal direct from the anode of the video amplifier,

irr~ir

>de of CI

Synchronizing

Separator

-WW--r—WVv—I

—

C2

Anode ofVideoAmplifier

R2-WW-

_n n_

C4S

FROMLINE

TIMEBASE

R5 A&C

BLACK LEVEL

R3

P.4-1

:C3

C5

R6

LACK LEVEL Tt^;--Positive pip

FIG. 2.6. IMPROVED SYNCHRONIZING PULSE-CANCELLED A.G.C. CIRCUIT (Milliard)

through resistor R2 . Line pulses from the synchronizing separator are fed

through Cx and Rx also to the grid so that the synchronizing pulse cancelled

waveform appears on the grid of Vt . Current will flow through valve Vx

charging up capacitor C3 to approximately the black level voltage. CapacitorC3 will actually charge to the black level voltage (which is now positive due to

the positive d.c. voltage on the anode of the video amplifier) plus the grid baseof the valve. Imagine that the grid is at zero potential, then C3 would chargeup until the grid-cathode voltage was equal to the grid base (i.e. cut-off


voltage) of the valve. If the black level was +25 volts then the voltage across

C3 would rise to the cut-off voltage plus 25 volts. In order that C3 does not

remain fully charged should the black level decrease, R3 is added so that C3

discharges slightly during the period of a line.

Since the voltage across C3 decreases in this way during the line period (the

time constant must be kept fairly small so that the circuit is rapid acting),

valve Vi would conduct during any black portion of the signal (since the

signal is then at black level) during the line. Thus the final voltage across C3

at the end of each line would depend on the amount of black in the picture

(particularly towards the end of the line) which is undesirable. Capacitor C2

is added to overcome this. This capacitor produces a differentiated line pulse

which is superimposed on the main line pulse, resulting in a small positive-

going pip on the back of each line pulse. This causes C3 to charge to a slightly

higher voltage at the start of each line so that C3 always remains slightly above

black level voltage during the whole line time. Since the pip is of constant

amplitude it does not upset the operation of the circuit. Up to this point the

result is similar to that obtained by the simple circuit.

Valve V2 is now added as a d.c. amplifier so as to amplify the voltage

across C3 and to convert it into a suitable negative voltage. Valve V2 is fed

with large positive-going line pulses from the line timebase and therefore

conducts on these pulses. This charges up the capacitor C« with the polarity

shown. The action of the circuit is similar to that of a d.c. restorer. With nobias on the grid of V2 (i.e. the valve acts as a diode), C4 will charge up to the

full pulse voltage and the anode of V2 will be at a negative potential (between

pulses) equal to the magnitude of the line pulses fed to V2 . In this circuit the

grid-cathode voltage is variable and the voltage to which C4 will charge will

depend on the potential at which current will flow in V2 . If the pulse voltage

is 150 volts and V2 does not conduct until the anode reaches a potential of

50 volts, then C4 will only charge to 100 volts. Thus the anode of V2 will only

be 100 volts negative between pulses instead of 150 volts when no bias is

applied to V2 . Therefore, the negative voltage between pulses depends on the

voltage on the cathode of V2 (i.e. the voltage across C3) which depends onthe signal strength and the grid voltage provided by the potential divider R4,

which forms the contrast control. The voltage on the anode of V2 is smoothed

by Rs and C5 resulting in a negative a.g.c. voltage.

A reduction in the signal input results in a larger black level voltage on

the grid of Vi, and so C3 will charge to a higher positive voltage. This tends

to cut off V2 (since the cathode is going positive with respect to the grid)

resulting in a smaller negative voltage at the anode and lower a.g.c. voltage.

This increases the gain of the r.f. and i.f. stages so bringing the demodulator

voltage back to normal. The a.g.c. voltage can be fed to the various stages as

already described. Although line pulses are used from the line timebase in

this circuit, the circuit will operate quite satisfactorily even if the timebase is

not synchronized.

Other circuits are used to measure the magnitude of the synchronizing

pulses and these are commonly called gating systems for the reason that wegate out part of the composite waveform.

We may: (0 Measure the amplitude of the field pulses; or (if) measure

the amplitude of the line pulses.

In figure 2.7(a) is shown the waveform during the time of the field pulses

and field flyback. If we measure the magnitude of the signal (which consists

of synchronizing pulses only, the vision signal being suppressed for these

lines) during the period A-B we shall get a measure of the amplitude of the

pulses which, of course, are independent of the type of picture being trans-

mitted. At (b) is shown the waveform over a period of about two lines. If wemeasure the voltage during the period of the back porch (i.e. X-Y) again we


A

« 14 LINES

©FIG. 27(a) FIELD PULSES DURING FIELD FLYBACK TIME

(b) LINE PULSES SHOWING BACK PORCH(405 lines)

shall obtain a measure of the pulse amplitude. In both cases it will be seen

that it is necessary to measure the voltage during a specific time only. This is

often referred to as a gated system as.the device measuring the voltage mustbe "gated" so that it will only operate during the required period, the "gate"being opened by suitable pulses. It is the back porch of the line pulse which is

almost universally used. This method makes the a.g.c. more rapidly acting

than when the field pulses are used, as the sampling rate is 10,125Hz or

15,625Hz instead of 50Hz. Circuits using field pulses will not be described,

but the principles are exactly similar.

A circuit using line gating pulses is shown in figure 2.8. Valve Vi is the

gated valve which is fed with a negative video signal on its grid and the

cathode is returned to a variable positive voltage from the potential divider Pt .

The waveform at the grid of the line blocking oscillator is fed to the shaping

H.T.+

FIG. 2.8. GATED A.G.C. SYSTEM

valve V2 , so that the valve conducts during the period of line pulses and is

then cut off by the flyback. The pulse of current in Lx results in a negative

pulse followed by a positive pulse (due to ringing in Lj) and this positive pulse

is designed to correspond to the back porch of the line synchronizing pulse.

The positive pulse is applied to the anode of Pi. (Note that there is no steady

h.t. applied to the anode of Vt). This positive voltage will cause the valve to

conduct but the magnitude of the current which flows will depend on the

cathode voltage and the magnitude of the voltage fed to the grid from the


video amplifier (through a cathode follower) at the instant of the back porch.

The voltage on the cathode depends on the setting of Px (which forms the

contrast control). If the signal fed to the grid of Vx is large the bias will be

large (it being a negative video signal) and little current will flow in Vx . Hence,

a large pulse will be produced. If the bias is small, Vx will tend to short out

the pulse, which is amplified by V3 . The output pulses of V3 are then rectified

by Ret and smoothed by d. Thus, a large input signal results in a large bias

on Vu large pulses to V3 and out of V3 which, in turn, results in a large negative

a.g.c. voltage. This reduces the gain and hence the signal fed to Vt . Variation

of the potential from i^ also varies the magnitude of the pulses—hence the

a.g.c. voltage and thus the gain and contrast. It is important to note that

Vi can only conduct when it is gated by the pulses from V2 and so the a.g.c.

voltage depends on the magnitude of the back porch, i.e. on the black level

and not on the type of picture.

Another circuit is shown in figure 2.9 where Vt is the gated valve. Vx

is fed with a positive video signal from a resistor in the cathode of the

video amplifier, but the cathode voltage from Pt is such that the valve would

H.T.+


not conduct. The grid is also fed from the anode of the synchronizing

separator with negative synchronizing pulses through a small capacitor Cx

which differentiates the pulses producing a negative pulse corresponding with

the start of the line pulse followed by a positive pulse corresponding to the

back porch. This is shown in figure 2.10 when it will be seen that the latter

pulses sit on the back porch, and cause Vx to conduct. Now, the pulses are

constant in magnitude since they come from the synchronizing separator and

so the height of the pulse relative to the cut off of Vx will depend only on the

height of the back porch. An increase in signal makes the back porch larger

(i.e. more positive) and the pulses are higher. The pulses out of Vt are

therefore larger. The amplified pulses at the anode of Vx are rectified by

Vz and smoothed by C2 to produce the negative a.g.c. voltage.

Non-delayed a.g.c. voltage is fed to the mixer and a smaller non-delayed

voltage is fed to the first and second i.f. stages. A delayed a.g.c. voltage is

fed to the r.f. amplifier through Rt . Delay is produced by V, due to the

current in R2 and R3 . R2 forms a sensitivity control since this controls the


SYNCHRONIZINGPULSES

DIFFERENTIATEDSYNCHRONIZING PULSES(as on grid of VI

)

VIDEO SIGNAL

\r

CUT-OFF OF VI

VIDEO SIGNALPLUS DIFFERENTIATED

SYNCHRONIZING PULSES

FIG. 2.10. WAVEFORMS

delay voltage. In this way a.g.c. voltage is only fed to the r.f. stage when thesignal (and hence the a.g.c. voltage) is large. When the delay is in operationthe a.g.c. voltage fed to the r.f. amplifier is slightly positive since the cathodeof V3 is taken to a small positive potential. To prevent upsetting the a.g.c.circuit by sudden overloads (such as due to channel changing) another circuitis added. In the anode of the last i.f. valve, K4, is a resistor, and when thevalve is overloaded, it acts as an anode bend detector, and a video signal is

produced across the resistor Rt. This is fed to V3 through R5 and C3 . Thesignal is rectified by V3 and produces a negative voltage which is mainly fedto the r.f. stage, so reducing the gain until the normal circuit takes control.The normal r.f. voltage that would be developed across Rt is by-passed tochassis by capacitor C4 .

There are a number of circuits using a sampling diode, one type beinggiven in figure 2.11. The anode of valve V^ is fed with a negative video signalfrom the video amplifier. Positive line pulses are fed to C2 from the lineoutput transformer which are differentiated by C2 and Rt to produce apositive pulse corresponding to the start of the line flyback, and a negativepulse corresponding to the back porch of the line pulse, which is the onethat is used. This pulse is fed to the lower end of Q through R2. In theabsence of pulses, Vt is non-conducting because its cathode is returnedthrough R3 to a potential higher than the voltage on the anode. d willcharge to the voltage difference between the positive voltage fed to R3 andthe positive voltage fed from the potential divider Plm Assume that Vr is

disconnected and that a negative pulse of 40V is fed to point X. Since the


V2

AjC£.UNEr€r T .Vr,

C3J.

R2 539KRl

-Wr pi

FIG. 2.11 GATED A.G.C. SYSTEM

voltage across Ci cannot change instantly the pulse voltage will be divided

between R2 and R3 and, if they are equal, there will be 20V across each.

This means that points Y and Z would each decrease by 20V. Assume nowthat Vx is connected and that its anode is 10V less positive than the voltage

fed to R„ at the instant when the pulse occurs, i.e. at the instant when the

back porch occurs. When the pulse is applied, point Z will tend to go negative

but it will only decrease by 10V and then K, will conduct (or clamp the

potential at Z to the anode potential of Vd and prevent its going more

negative. Thus, the voltage drop across R2 will be 30V. Hence, point Ydrops by only 10V. The lower the potential of the anode of Vx the larger

will be the drop in potential at points Y and Z each time a pulse occurs.

The pulse appearing at Y is rectified by V2, smoothed by C3 and is fed as

a.g.c. voltage.

Thus, if the signal on the anode of Vi is larger, the level of the back porch

(black level) will be lower. This is shown in figure 2.12 where with a small

© ©HG. 2.12. EFFECT OF CHANGING CONTRAST ON VOLTAGE OF BACK PORCH

signal at (a) the voltage corresponding to the back porch is 100V, while with

the larger signal at (b) the level is only 80V. Hence, larger pulses will be

produced at Y with a large signal (because points Y and Z have farther to

drop before diode Vx conducts and clamps or catches the voltage) which in

turn will produce a larger a.g.c. voltage and decrease the gain to reduce the

signal fed to the anode of Vx .

Valve V2 will only conduct when V2 cathode goes negative. The cathode

is fed with a positive voltage from Pt which acts as a delay voltage because

V2 will not conduct until the pulses produced at Y are larger than the delay

voltage from iV Pi therefore varies the gain and forms the contrast control.

Positive feedback may be used to improve the performance of this circuit.

Instead of taking R3 to a source of fixed positive voltage it is taken to a

22 TELEVISION SERVICINGresistor in the anode circuit of the first i.f. valve, the resistor being by-passedto chassis for r.f. When the a.g.c. voltage is applied to the first i.f. valve, theanode current is reduced, the drop across the resistor is reduced and so thevoltage applied to R3 is increased. Thus, for the period between pulsespoint Z is raised in potential. This means that point Z and point Y will dropin potential by a larger amount when the pulses are applied, before thepotential is clamped by diode Vx . Larger pulses are therefore produced at YThis results in a larger a.g.c. voltage and still further reduction of the anodecurrent in the i.f. amplifier, and a still higher potential fed to R3 . Thus wehave positive feedback and the performance is improved since the effectivegain of the circuit is increased. The circuit does not oscillate because of thenegative feedback produced by the a.g.c. action and owing to the fact that thevalue of the resistor in the anode circuit of the first i.f. stage is carefullyadjusted. The circuit results in an almost constant output from the de-modulator.

In circuits of this type the last i.f. valve and the video amplifier are likelyto be overloaded during the warming-up period as no a.g.c. voltage can bedeveloped until the line output stage is operating. This is the last portion ofthe circuit to operate, owing to the long heating time of the efficiency diode.To obviate this overloading, the screen of one of the i.f. valves may be fedfrom the boosted h.t. supply through a suitable dropping resistor.

Another type of sampling circuit is shown in figure 2.13. Positive pulses


are fed to Ct from the line output transformer and are differentiated byCi and R t to produce a negative pulse corresponding to the back porch ofthe line pulses. This is fed to the cathode of V1 through C2 . The anode ofVx is fed with a negative video signal from the cathode follower V2 , whichalso feeds the cathode of the cathode ray tube. If the anode of Pi wereconnected to chassis the circuit C2, R2 and Vt would form a d.c. restorercircuit as described in Part Two. The pulse would sit on the zero lineas shown at (a) of figure 2.14. The anode of Vx is not at chassis potentialbut is at the potential of the cathode of V2 . Since K, is non-conducting (as ina d.c. restorer) except when the pulse occurs, it does not matter what thepotential of Vx anode is between pulses. The potential to which the waveform


is restored is settled only by the potential on the anode of V^ when it conductsat the instant of the pulse. Thus, the pulse now sits on the back porch asshown in figure 2.14(b). As the amplitude of the video signal varies (owingto change of signal strength), the level of the back porch moves up and downas in figure 2.12 and the pulse waveform at X rides up and down with it.

V3 conducts during the period of the negative pulse by an amount whichdepends on the height of the pulse above the zero line (figure 2.14) and thesetting (say Y in figure 2.14) of the potential divider iV The pulses in theanode circuit of V3 are rectified by /tej and fed as a.g.c. voltage. Variationof Pt causes variation of the magnitude of the pulses out of V3—hence thegain—and forms the contrast control. If the signal level at K2 increases, the

PULSES AT X(with VI to chossis)

©

SETTING

OF PI

PULSES AT X

PULSES WHICH CAUSE

V3 TO CONDUCT

WAVEFORM ONCATHODE OF V2

©FIG. 2.14. WAVEFORMS OF CIRCUIT SHOWN IN FIG. 2.13

black level decreases; the pulses move down and the cathode of V3 is drivenmore negative, producing larger pulses in the anode circuit. This producesa larger a.g.c. voltage and reduces the gain to decrease the signal from V2 .

In this receiver overloading of the video amplifier is prevented by inter-connecting the vision and sound a.g.c. lines so that, during the warming-upperiod, the sound a.g.c. voltage prevents excessive gain until the vision a.g.c.voltage is produced. A mean of the two a.g.c. voltages is fed (through adelay circuit) to the r.f. stage and the first two i.f. stages in normal operation.

Gated a.g.c. circuits do not appear to be common in modern receivers.This may be because they tend to be complicated and viewers appear to besatisfied by the mean level type of a.g.c. Gated a.g.c. circuits also suffer fromthe disadvantage that they will not operate until the line timebase is operatingand also is correctly synchronized. Thus, if the line timebase comes out ofsynchronism this upsets the a.g.c. and the operation of the receiver and it maybe difficult to get the line timebase back into synchronism. Owing to the factthat the line timebase is the last part of the television receiver to operate, somemeans must be used to prevent overloading (due to excessive gain) until theline timebase and the a.g.c. are operating. One method has just been described.Various other methods may be used such as feeding the screen grid of one ofthe i.f. valves from the boosted h.t. supply. The boosted h.t. supply is not,of course, present until the line timebase is operating. Another method is touse the negative voltage on the grid of the line oscillator to supply some a.g.c.voltage and then to cancel it with a positive voltage from the boosted h.t.

supply, by which time the line output stage is operating and the a.g.c. hascome into action.


625-LINE SYSTEMLet us now consider methods used on a 625-line system using negative

modulation. In figure 2.2 is shown the carrier when positive modulation is

used and in figure 2.15 the carrier is shown when negative modulation is used.

FIG. 2.15. NEGATIVE MODULATION(a) Dark Picture (b) Bright Picture

(a) is for a dark picture while (b) is for a bright picture. It will be seen that

there is an important difference in the amplitude of the carrier. With positive

modulation it is variable depending on the picture content, but in negative

modulation the maximum amplitude of the carrier (corresponding to the

synchronizing pulses) is constant quite independent of the picture content.

Thus the magnitude of the carrier during the synchronizing pulses (or the

maximum amplitude of the carrier) is a measure of the signal strength. Hence

it is possible to use a peak detector as shown in figure 2.16. The normal

\A£ C charger to Ifais voHaei.

*» TO VIDEOAMPLIFIER

FIG. 2.16. SIMPLE PEAK DETECTOR

demodulator Pi is connected so that a negative output is obtained, resulting

in a positive-going video signal. The maximum negative voltage of this

waveform is a measure of the signal strength. Valve Vz will conduct and

charge C to the maximum negative voltage. Resistor R is added so that the

voltage across C decreases when the signal input decreases. A delay voltage

may be applied to the diode or to the circuit in the usual way. The circuit

has the disadvantage that any interference which extends beyond the syn-

chronizing pulses will cause V2 to conduct and C will be charged up as a

result, probably to a higher voltage than that corresponding to black level.

Thus interference will cause an increase in the a.g.c. voltage and result in a

lowering of the picture contrast. The effect of interference may be reduced by

making the time constant Cx R long, but then the a.g.c. system becomes slow

acting. The advantage of this circuit lies in its simplicity.

Mean level a.g.c. may be used as on the 405-line system, the voltage being

obtained from the grid of the synchronizing separator and the operation is


exactly similar to that on 405 lines. The system suffers from a rather lowloop gain, and the range of a.g.c. control is limited. The a.g.c. voltage dependson the picture content and the system tends to be slow acting. On the otherhand it is simple to apply. (These advantages and disadvantages apply also toits use on a 405-line system). With negative modulation the system suffersfrom the possibility of "lock out". This is caused by the fact that if a largesignal is applied to the video amplifier (the video amplifier being d.c. coupled)then this may be cut-off. This occurs because with negative modulation anegative voltage output is obtained from the demodulator instead of a positiveone as in the case of a positive modulation system. Thus no a.g.c. voltageis developed at the synchronizing separator (since there is no input to thegrid) and the set is locked out until the large signal is removed. This diflicultymay be overcome by using circuits which will prevent severe overloading ofthe video amplifier when no a.g.c. voltage is present. This may be achievedby the use of a C-R circuit in the i.f. amplifier as in figure 2.17. When the

(a)

FIG. Z17. LIMITER CIRCUIT

signal is of normal amplitude the capacitor and resistor have no effect. Whena large signal is present the valve is driven into grid current and C becomescharged with the polarity shown and self-bias (like an oscillator) is producedso tending to reduce the gain of the valve. The effect can also be overcome byusing a.c. coupling between the demodulator and the video amplifier. Anotheralternative is to connect the a.g.c. line to the demodulator (vision) as infigure 2.18. Should a large signal come from the demodulator and there is

u./

DEMODULATOR

>

A.G.C. "LINE

FIG. X18. USE OF DEMODULATOR FORA.G.C. DURING WARMING-UP TIME

little or no a.g.c. voltage then the rectifier Re will conduct and the negativedemodulator voltage will be fed to the a.g.c. line, so reducing the gain untilthe circuit is restored to normal operation. Under normal conditions C and R


act as a smoothing circuit so that the mean demodulator voltage is applied to

the upper side of the rectifier. So long as the a.g.c. voltage is greater than the

mean signal from the demodulator then Rt will not conduct and will not

interfere with the normal operation of any a.g.c. circuit.

As already explained the contrast is normally controlled by varying the

a.g.c. voltage by varying the delay voltage. This changes the gain of the r.f.

and i.f. amplifiers and so varies the magnitude of signal level at the demodu-lator and hence the contrast. On dual-standard receivers some switching maybe used of the a.g.c. circuits but commonly mean level a.g.c. is used on both

systems. A single contrast control may be used as a front of receiver control

which operates on both systems. An alternative sometime used is to dispense

with a front of receiver contrast control but fit two preset controls at the

back of the receiver, one for 405 lines and the other for 625 lines. A brightness

control is, of course, fitted on the front of the receiver.

Instead of using the synchronizing separator to obtain the mean level a.g.c.

voltage a separate diode may be used. This has the advantage that it is rather

easier to apply a delay voltage since it can be applied directly to the diode.

The synchronizing pulse-cancelled circuit (MuUard) described earlier (see

page 16) may be modified to operate with negative modulation. The modified

circuit is shown in figure 2.19. In this figure the same component numbers

Synchronize pul»s/from Synchronltino,

X Snparotw

ITT

~7tN"TvN"-

"uc

"u™l

o

FIG. 2.19. SYNCHRONIZING PULSE-CANCELLED CIRCUIT FORNEGATIVE MODULATION

have been used as in figure 2.6. The operation of K, is as before so that

C3 charges to the black level of the voltage fed to the grid of Fi from the

video amplifier. One must now remember that the black level now changes

in the opposite direction to that when positive modulation is used. This is

shown in figure 2.20. The waveform at the grid of the video amplifier is

shown at (a) for two values of signal while (b) shows the corresponding

waveforms at the anode of the video amplifier. Thus, on a smaller signal the

black level voltage decreases both on the video amplifier grid and on the

output from the video amplifier. Hence the voltage across C3 decreases as

the signal decreases (opposite to that on positive modulation and shown in

figure 2.7). In the circuit of figure 2.19 this decreases the positive voltage on

the grid of V2 , i.e. it increases the negative bias on the valve (i.e. voltage

between grid and the cathode) so causing V2 to conduct less and so reducing


©Vt'tfeo Amplifier Gr/<f

©

Vidto Ampliflar Anodt

FIG. 2.20. EFFECT OF SIGNAL AMPLITUDE ON BLACK LEVELVOLTAGE (NEGATIVE MODULATION)

the a.g.c. voltage across C5 . This, of course, causes the signal to the videoamplifier to increase until equilibrium is reached.

TRANSISTOR CIRCUITSWhen turning to transistor receivers, or those receivers using transistors

in the r.f. and i.f. stages, a.g.c. operation tends to be more complicated andthere are more variations in the circuits used. To vary the gain of a valve,

the grid bias voltage is varied making the grid more negative in order to

reduce the gain. Since no current flows in the grid circuit only an a.g.c.

voltage is required, i.e. the circuit can be of high impedance. When dealingwith transistors the gain can be varied in two ways. If the collector is feddirectly from the h.t. line, through the tuned circuit, the gain can be reducedby reducing the base bias current, so reducing the collector current. Thecurrent gain is reduced at low values of current and eventually the transistor

is cut off. If the transistor is a p-n-p type the base bias (relative to emitter)

will be a negative voltage and the effect of a.g.c. must be to make the baseless negative. Thus the a.g.c. line must be positive—the opposite of the valve

case. If the transistor is an n-p-n type then the reverse is true and the a.g.c.

line must be negative. This method of gain control is known as reverse bias

and is commonly used in radio receivers but not in television receivers. Thedifficulty of using this method in television receivers is that the i.f. transistors

are working into relatively low impedance circuits (relative to a radio receiver)

and a considerable voltage output is required without distortion. With this

method of control maximum signal is applied to the transistor when it is

operating with minimum collector current and distortion is likely to occur.

The alternative method is to increase the base bias so that a large collector

current flows. To reduce the collector dissipation a decoupling circuit is

placed in the collector circuit so that the collector voltage falls at large collector

currents. Successful operation depends on the use of a suitable type of trans-

istor where the current gain decreases at high currents. Suitable transistors

are manufactured for this purpose, a typical characteristic being shown in

figure 2.21. It should be noted that the characteristics have not been drawnfor equal increments of base current. The change of collector current for achange of base current is much less at high values of collector current. If the

line AB represents the resistance of the de-coupling circuit then at low values


BASE CURRENT2

It

COLLECTOR VOLTAGE

Vce

FIG. 2.21 TRANSISTOR CHARACTERISTIC FOR A.G.C. CONTROL

of collector current the transistor operates about point C with a large gain.

At high collector current (i.e. high values of base bias current) it operates at

point D where the gain is low. Thus when the signal input is large the trans-

istor is operating with a large collector current and the characteristics are

such that a large output is obtainable without serious distortion. With this

method of operation the a.g.c. voltage must be of opposite sign to that for

reverse bias control. For p-n-p transistors the base bias (negative with respect

to the emitter) must be increased and hence the a.g.c. line must be negative.

For n-p-n transistors the a.g.c. line must be positive so as to increase the

positive base bias. This method of control is known as forward bias control.

When the base current and hence collector current are changed there will be

changes in the input resistance and capacitance of the transistor which will

cause changes in damping and tuning. Care must therefore be taken in the

design of the amplifier that these changes do not appreciably affect the circuit.

The effect is reduced by using heavily damped and hence wideband circuits.

As already mentioned no current flows in the a.g.c. line of valve circuits

but this is not so in transistor circuits since the a.g.c. line must supply some

of the base bias current. With forward bias control this current is at maximumwith maximum a.g.c. voltage. For this reason it is normal to place an amp-

lifier between the source of a.g.c. voltage and the a.g.c. line itself. Fortunately,

it is relatively easy to use a transistor for this amplifier.

It was explained in connection with valves that it is important to apply

the a.g.c. voltage to the r.f. and i.f. amplifiers in the correct way. The gain

of the i.f. amplifier must first be reduced followed by reduction in gain of the

r.f. amplifier (i.e. tuner). This is equally necessary in transistor circuits and

calls for suitable circuits. In valve circuits a diode is often used to cause the

a.g.c. voltage to be applied in the correct sequence. Also, such a diode is

commonly used in conjunction with a potential divider circuit to vary the

contrast. The potential divider circuit is placed across the h.t. supply and

essentially opposes the a.g.c. voltage, so controlling the gain and hence the

contrast of the picture. In a transistor circuit using p-n-p transistors the

collector supply is negative and the a.g.c. voltage is also negative. Thus there

is no opposing voltage readily available as in the valve circuit. Similarly,


with n-p-n transistors the collector supply is positive and the a.g.c. line is

positive. Hence some alternative method of contrast control must be used.

Although the contrast may still be varied by varying the gain of the r.f.

and i.f. amplifiers by a suitable circuit, many receivers now use what is termed

a high level contrast control. Instead of the contrast control varying the gain

of the r.f. and i.f. amplifiers it now controls the gain of the video amplifier

(or video amplifier driver stage); or is a potential divider across the video

amplifier load. The potential divider is used where the video amplifier is a

valve: a basic circuit is shown in figure 2.22. The video amplifier is V, with a

load consisting of R and L. A potentiometer P is placed across the load so

FIG. 2.22. HIGH LEVEL CONTRAST CONTROL

that the output can be varied. A capacitor C may be added to improve the

frequency response of the circuit. To prevent poor frequency response at

low setting the value of P should be low but not too low or it will short out

the load. A value of 10-20k£i seems usual. This arrangement is essentially

the same as the volume control of a normal receiver. When a transistor

video amplifier is used, the gain of the transistor circuit may be varied, for

example, by altering the value of resistor in the emitter circuit; this controls

the amount of negative feedback.

The advantages of high level contrast control are that there is no inter-

ference with the a.g.c. line and the demodulator and video stage always

operate at the same signal level. It is also usually possible to feed the synchron-

izing separator with a constant signal independent of the setting of the

contrast control. This is obtained in figure 2.22 by feeding the synchronizing

separator from the anode of the video amplifier.

One method of feeding the a.g.c. voltage in the correct way to the r.f.

and i.f. amplifier is shown in figure 2.23, which is only a basic diagram showing

the principle. (See article by M. C. Gander and D. S. Hobbs in Milliard

Technical Communications, January 1967, p. 119).

Trt is the tuner r.f. amplifier with emitter resistor R2, tuned circuit Lt Cx

and decoupling circuit R x C3 . Standing bias is supplied by R3 and R4 and these

are of such values that a relatively small bias current is supplied to Trx and

hence the gain is at a maximum. Transistor Tr3 is the i.f. amplifier with Rl0

as the emitter resistor, L2 C2 as the tuned circuit and decoupled by R9 and C4 .

Standing bias is produced by R7 and Rs which have such values that a large

bias current is provided and the gain of Tr3 is at a minimum (assuming D2

disconnected). Transistor Tr2 is the a.g.c. amplifier and its base is biased

positively by a circuit (not shown) so that it is normally heavily conducting.


FIG. 2.23 TRANSISTOR A.G.C. CIRCUIT

Thus point X is at low potential. Under these conditions point Z is morepositive than point X and hence Dx is non-conducting. However, Y is morepositive than point X, hence D2 is conducting and acts as a short circuitbetween points X and Y bringing down the potential of point Y. Thus thebias on Tr3 is reduced and the gain increased (by suitable design to a maximumvalue).

A suitable negative-going a.g.c. voltage is fed to the base of transistor Tr2so decreasing its collector current. This causes point X to rise in potentialcausing Y to follow and so decrease the gain of the i.f. amplifier Tr3 . Thisaction continues until point X reaches the potential that point Y would haveby virtue of resistors R 7 and Rs . At this point D2 becomes non-conductingand 7>3 remains at minimum gain. At about the same time it is arranged, bysuitable component values, that X is the same potential as point Z. ThusDi conducts and Z and X are shorted together. Further increase of a.g.c.voltage causes point X to rise still further taking point Z with it, so increasingthe bias current of Trt and reducing its gain. Thus the gain of the i.f. amplifieris first reduced followed by that of the r.f. amplifier.

There are variations of this type of circuit that may be used to control thegain in the appropriate way. Very many circuits are possible (transistors lendthemselves to circuit variations much more than valves) and they cannot all

be detailed, but the following are some ideas used.

(0 The a.g.c. voltage is fed through a transistor amplifier to the tuner unitand this is used as a d.c. amplifier to vary the a.g.c. voltage on the i.f.

amplifier, so reducing its gain. When the a.g.c. voltage exceeds a certainvalue a clamping diode prevents further change of the i.f. amplifier biasand the a.g.c. voltage now causes changes in the gain of the tuner unitr.f. stage.

(h) The a.g.c. voltage is fed to the i.f. amplifier which causes a diode toconduct damping of the mixer stage. The i.f. amplifier also acts as anemitter-follower to feed a diode which damps the input circuit of ther.f. stage. The diodes operate in sequence by choice of suitable potentials.

{Hi) The a.g.c. voltage is fed through a single-stage transistor amplifier tothe i.f. amplifier. This stage acts as an a.g.c. amplifier feeding anothertransistor amplifier which feeds the r.f. amplifier.

(iv) The a.g.c. voltage feeds a two-stage transistor amplifier feeding the i.f.

amplifier and r.f. amplifier through delay diodes.The a.g.c. voltage to drive the a.g.c. amplifier may be obtained in a number

of ways. Since it is relatively easy to amplify this a.g.c. voltage there is not


the same necessity to obtain a large voltage as in the case of valve circuits.

Some arrangements are as follows,

(i) Using the output of the video amplifier or video driver stage. Providedthere is some d.c. coupling between the demodulator and video amplifierthe mean output of the latter will vary with the value of the signal fedto the demodulator (and also with picture content). Thus if the outputis smoothed a suitable a.g.c. voltage can be obtained.

(«) Using the output of the phase-splitter driving the video amplifier. Thisis sometimes used in dual-standard receivers to prevent reversing thedemodulator diode. The operation is as (/).

(jit) Using the output of the demodulator. This has a d.c. component whichdepends on signal strength (and picture content).

O'v) Using a separate demodulator fed from the last i.f. amplifier.

The polarity of the a.g.c. voltage required will depend on whether p-n-por n-p-n transistors are used and also on the arrangement of the a.g.c. amplifierbetween the source of the a.g.c. voltage and the a.g.c. line.

All the circuits described operate as mean level a.g.c. circuits. Gatedcircuits do not appear to be used although there is no reason, other than theadded complications and cost, why they should not be.

AUTOMATIC CONTRAST CONTROLWhen the ambient lighting on a cathode ray tube is changed it is desirable

to change the contrast and brightness of the picture—the greater the ambientlighting the greater should be the contrast and brightness. These changesmay be made manually but the introduction of a new type of photoelectriccell enables this to be done automatically and this feature is now fitted to somereceivers. The photoelectric cells used are of the cadmium sulphide typewhose resistance decreases with increasing light applied to them. The cell is

fitted on the cabinet so that the ambient light falls on it.

When mean level a.g.c. is used the cell is made to form part of the contrastcontrol circuit. Two examples are given in figure 2.24. In (a) an increase inambient lighting decreases the resistance of the cell C and hence increases thedelay voltage from the contrast control, so reducing the a.g.c. voltage andincreasing the contrast. The action of (b) is similar in that the effective

©

Synchronizing

Stparotor valve

A.G.C-<-

HX+

©w

Synchronizing

Separator valve

-T—WW—rfWH-TS I WW '

CONTRASTCONTROL

FIG. 2.24. AUTOMATIC CONTRAST CONTROL CIRCUITS


resistance between the potential divider and the a.g.c. line is decreased with

increas in ambient lighting. This increases the effective delay voltage so that

the a.g.c. voltage is decreased and the contrast is increased.

The arrangement may be applied to the amplified synchronizing pulse

cancelled circuit shown in figure 2.6. In this case the cell must be placed at

the lower end of Rt (opposite to that of figure 2.24(a) ) as the contrast control

operates in the opposite direction to those in figure 2.24.

It should be noted that decreasing the a.g.c. voltage not only results in a

greater video signal and greater contrast but also a more brilliant picture

since it raises the black level. This is because the tips of the synchronizing

pulses are at a constant level. This is shown in figure 2.25 where the waveform

is shown as would be present at the grid of the video stage. Increasing the

video input increases the synchronizing pulse amplitude and so increases the

voltage corresponding to black level which is equivalent to an increase in the

brilliance of the picture.

SMALLVIDEO SIGNAL

- - BLACK LEVEL

QFIG. 2.2S. EFFECT OF INCREASE OF VIDEO SIGNAL ON

BRILLIANCE

CHAPTER 3

FLYWHEEL SYNCHRONIZING CIRCUITS

T'he effect of noise or interference on the vision signal degrades the picture

in two ways: (/) It increases the amount of grain or "noise" on the

picture; and (/i) it affects the accuracy of the line synchronizing causing

the triggering of the line timebase to be irregular.

The first effect can be reduced by suitable interference reducing circuits

which have been described in Chapter 1. The second effect can be reduced

by the use of what is known as flywheel synchronizing or automatic frequency

control circuits. The effect of the interference on the synchronizing may be

the more serious effect, particularly with a 625-line system using negative

modulation. It should be remembered that in the case of a negative modula-

tion system any interference causes pulses in the synchronizing pulse region

FLYWHEEL SYNCHRONIZING CIRCUITS 33

and may cause premature triggering of the timebase. For this reason, flywheel

synchronizing circuits are universally used on 625-line receivers.

The effect of the interference on a typical 405-line vision signal is shown in

figure 3.1, when it will be seen that the sharp pulses due to interference tend

FIG. 3.1. VIDEO WAVEFORM SHOWING NOISE

to alter the leading edges of the synchronizing pulses. A positive interference

pulse just at the start of the line pulse delays the fall in voltage and so delays

the flyback and the scan of the next line which displaces this line relative to

the others. Due to this effect the lines tend to be displaced sideways in a

random manner so that what should be a sharp leading edge to an object as

shown in figure 3.2(b) becomes a ragged edge as shown at (a). The effect is

to reduce the horizontal definition by quite noticeable proportions.

© ©FIG. 3.2.

EFFECT OF INTERFERENCE ON VERTICAL LINE

The effect of a flywheel synchronizing circuit is to almost completely

overcome this. The arrangement has, of course, no advantage (and may have


some disadvantages) in a good reception area where interference is small, butit is extremely valuable in fringe areas on 405-line receivers and on all 625-linereceivers. In the flywheel synchronizing circuit, instead of using each line-synchronizing pulse to directly synchronize each line, the pulses are used moreas a whole so that, if an odd pulse is mutilated by interference, it makes little

difference to the synchronizing. With the normal synchronizing circuit weuse a timebase which runs at approximately the correct frequency and this is

synchronized, or kept in step, by the synchronizing pulses, so that the flybackstarts at the leading edge of each pulse. The self-running property of thetimebase is not really used, except when the signal fails. In the flywheelsynchronizing circuit a device is used which acts more like a flywheel (hencethe name) which runs at nearly the correct frequency even if the pulses areabsent for a few lines. The flywheel device is a form of oscillator which is

maintained at exactly the correct frequency by the line synchronizing pulses.The basic idea of one type of flywheel synchronizing circuit is shown in

figure 3.3. The flywheel device is the oscillator O which is adjusted to oscillate

PHASEDETECTOR

P

REACTANCEVALVER

OSCILLATOR

SynchronizeO

To Tim<

Pulses

FIG. 3.3. PRINCIPLE OF FLYWHEEL SYNCHRONIZING CIRCUIT

at the nominal line frequency of 10,125Hz (405-line system), and the outputfrom this oscillator is used to synchronize the timebase. (In some otherarrangements the oscillator may actually be the timebase). The frequency ofthis oscillator will, of course, never be quite correct and it is necessary tomaintain it at the correct frequency and phase by the line synchronizing pulses,

which are obtained from the synchronizing separator in the normal way. Inorder to detect the error in frequency, or phase, of the oscillator a phasedetector P or comparator is used. Into this detector or comparator are fed theline synchronizing pulses and the output from the oscillator. This comparatoris so designed that it will give an output proportional to the error in phasebetween the synchronizing pulses and the output of the oscillator. The outputis then fed into the reactance valve R which operates on the oscillator so thatit corrects the phase error of the Oscillator. The circuit is arranged so that

should the oscillator tend to increase in frequency, and so cause the outputof the oscillator to lead the synchronizing pulses, an output is obtained fromthe comparator of such a polarity that the reactance valve will decrease theoscillator frequency, thus bringing the oscillator and line pulses into stepagain. The circuit is really a type of servomechanism and has some similarity

to the a.g.c. circuits used in radio and television.

When there is no interference the oscillator will run at the correctfrequency and phase, relative to the line synchronizing pulses. In the presenceof interference an odd pulse may be delayed but, due to the inertia of theoscillator (and to the smoothing in the phase detector or comparator stage),

it will make a negligible difference to the frequency of the oscillator andhence to the timebase. Thus, the flyback is not delayed by a mutilated pulseas in the case of the direct synchronizing circuit. The function of the flywheelcircuit is therefore to smooth out all the variations in the position or phaseof the line synchronizing pulses so that the average phase or timing of thepulses is used. In this way the noise or interference will have little effect onthe synchronizing of the individual lines and the ragged edges will be smoothedout. Even severe interference, such as from car ignition, will have only a smalleffect and will only cause slight movement of a portion of the picture, instead


of causing line tearing which takes place with normal direct synchronizing

methods.We will now consider in detail an older circuit of this type and each block

will be considered, starting first with section O shown in figure 3.4. Valve Vx

is the oscillator in which the screen grid, control grid and cathode are used

FIG. 3.4. FLYWHEEL SYNCHRONIZING CIRCUIT, OSCILLATOR SECTION(Ferguson 991 T)

to form a triode oscillator. Llt Cx form the resonant circuit in the grid circuit

which is coupled to the cathode circuit by the coil L2 . This valve operates as

an L-C oscillator at a nominal frequency of 10,125Hz and will produce asinusoidal voltage across the tuned circuit Lu Cx . By suitable choice of

component values the waveform at the anode can be made to consist of

square pulses having a mark space ratio of about 3/2. This is due to the

fact that the valve will run class-C, the bias being provided by the self-bias

circuit composed of the 0-001-^uF capacitor and 470-kO grid resistor. Thesepulses are fed to the line output valve Vz which operates as a conventional

line output stage. Feedback is provided from the screen grid of valve V2 to

the grid of Vx so as to sharpen the pulses obtained from the anode of Vx .

The frequency of the oscillator is controlled by the reactance valve circuit

which is shown in figure 3.5. The valve V3 is connected (from an a.c. point of

view) across the coil Lx and is fed with d.c. through the coil Lx . The reactance

valve is made to act as a variable capacitor and, since it is across the tunedcircuit, it will alter the frequency when its effective capacitance is varied.

The valve circuit is similar to that used in the automatic frequency control

circuits used in some pre-war radio receivers. The type used for this purpose,

which is designed to act as a variable inductance, is described in Volume 3

FIG. 3.5. FLYWHEEL SYNCHRONIZING CIRCUIT,REACTANCE VALVE SECTION (F«nuson 991 T)


of Radio Servicing. The operation of the circuit of figure 3.5 is as

follows: The anode is fed with h.t. through L, but the anode also varies

sinusoidally in potential due to the voltage developed in Lt from the oscillator.

The grid of the valve is fed through the potential divider formed by C3 , JJ,

and C4 . The capacitance of C4 is large in value and may be regarded, as far

as a.c. is concerned, as a short circuit. The reactance of C3 is made large

(about 200kQ) compared with Ri (12kfi) and hence the current in the circuit

will lead the applied voltage by nearly 90°. (The current flowing through a

capacitor leads the applied voltage by 90°). The voltage across iJt is in phase

with the current and so leads the applied voltage (which is also the anodevoltage) by approximately 90°. The voltage across R t is fed to the grid of V3

and this will cause the anode to take a corresponding alternating componentof current. This current will be in phase with the grid voltage, hence,

almost 90° leading the anode voltage. In other words the valve acts as a

capacitance, taking a leading current in the same way as a capacitor. Theeffective capacitance of the valve can be varied by varying the alternating

component of anode current by variation of the mutual conductance of the

valve. This can be done by varying the bias on the valve and this bias, or

control voltage, is fed through Rx . Thus, by controlling the voltage on the

grid of V3, its effective capacitance is changed and so the frequency of the

oscillator. Making the grid more negative will reduce the alternating com-ponent of anode current. This will cause the valve to be equivalent to a

smaller capacitance and hence increase the frequency of the oscillator.

The control voltage for valve V3 is obtained from the phase detector,

comparator or discriminator, the circuit of which is shown in figure 3.6. The

W II i p^i

FIG. 3.(5. FLYWHEELSYNCHRONIZING CIRCUIT.PHASE DETECTOR SECTION

(F«r«inon 991 T)

two diodes V, and V, are fed in antiphase (i.e. push-pull) from the centre-

tapped winding L3 which is coupled to the oscillator shown in figure 3.4.

The output from the normal synchronizing separator, which is composed ofnegative synchronizing pulses, is connected to the diodes through capacitor

Cj but, in this case, the pulses are fed with the same polarity to each diode.

If the pulses are absent the sinusoidal voltage across L3 will cause half cycles

of current to flow in V4,R2 and V5,R3 in the directions shown. Since these

currents will be equal (the circuit being symmetrical) there will be equal but

opposite voltages across R2 and R3 on alternate half cycles, and no steady

voltage (i.e. voltage having a d.c. or mean component) will be developedacross point A-B. In a similar way if the pulses are present and the sinusoidal

voltage is absent equal currents will pass through R2 and R3 in opposite

directions and no voltage will be developed across A-B. Now, suppose that

the line synchronizing pulses are applied so that they arrive at the instant


when the sine wave is zero as in figure 3.7. At (a) are shown the voltages fed

to the diode cathodes from L3, the two voltages being in antiphase. At (b)

FIG. 3.7.

OPERATION OF PHASE DETECTOR

are shown the synchronizing pulses fed in the same phase to the cathodes of

both K, and V5 from the synchronizing separator. The resulting voltages are

shown at (c) which are simply the addition of the voltages shown at (a) and (b).

The diodes will only conduct when the cathodes are negative with respect

to the anodes {i.e. the anodes are positive with respect to the cathodes) and

the shaded area shows the current flowing through the diodes. Since the

shaded areas are equal, then equal currents flow in R2 and R3 and the voltages

cancel out, producing no steady or mean voltage across points A and B.

Suppose that the oscillator frequency is too low and the sine wave then

tends to lag behind the pulses as shown in figure 3.8. As before, the resulting

voltages applied to the cathodes of the diodes are as shown at (c). It will

FIG. 3.8.

OPERATION OF PHASE DETECTORWHEN THERE IS A PHASE SHIFT

now be seen that the waveforms are not the same (as was the case in figure 3.7).

The pulse on valve Vs does not cause it to conduct but the pulse on valve V,

causes a large increase of voltage and a correspondingly larger current than

that through valve V5 . The larger current in valve Vt causes a larger voltage

across R2 than across R3 and so point A becomes negative with respect to

point B. It should be noted that the pulse on valve V5 may be large enough

to cause it to conduct; but even if it does the current will still be smaller than

that through valve Vt and the result will be the same. The negative voltage

at A is fed through the smoothing circuit consisting of R*,C6 and capacitance

C4 , the latter being shown in figure 3.5. A negative voltage fed to the reactance

valve V3 has already been shown to result in an increase in oscillator frequency

and thus the control voltage from point A will cause the oscillator frequency

to increase so that the sine waveform catches up with the line synchronizing

pulses, to a condition near to that shown in figure 3.7. The circuit R*,C6 and

C4 acts as a smoothing circuit to smooth out the voltage across R2 and R3,

38 TELEVISION SERVICINGwhich consists of mainly half cycles and also to give some of the flywheeleffect to the circuit. Thus, any sudden variations of voltage across A and B,which might be caused by interference, will not have any appreciable effecton the reactance valve. A negative voltage is obtained from the grid of theline output valve and fed to R5 from which a variable negative voltage maybe fed to point B and hence to the reactance valve. Resistor R5 acts as a linehold control to bring the oscillator to approximately the correct frequency,so that the phase detector circuit can take over control.

In a later version of the circuit the pentode oscillator valve Vx is replacedby a triode, the tuned circuit being in the anode and the coupling winding inthe grid circuit. A separate pulse-shaping valve is then used to convert thesine wave voltage at the anode of the oscillator to a sawtooth waveform tofeed the line output stage.

There are variations of this general arrangement:(0 The phase detector or comparator may be of a different type and two

other types will be described.

(w) Instead of using an L-C oscillator, the timebase itself may be used andthe control voltage from the phase detector being used to vary the frequencyof the timebase. Circuits of this type will also be described and are now muchmore common.

Another type of phase detector is shown in principle in figure 3.9(a).Positive line synchronizing pulses are fed to grid Gx of the pentode andpositive pulses derived from the line timebase or line output stage, are fed to

FIG. 3.9(a) PRINCIPLE OF PHASE DETECTOR

grid G3 . The bias on the valve is such that both grids are normally biasedbeyond cut-off and therefore no anode current will flow unless the two pulsescoincide. The anode current therefore flows for a period corresponding to theoverlap of the pulses. Thus, as one pulse varies in phase with respect to theother, the length of the pulse in the anode circuit changes and, therefore, sodoes the mean voltage developed across C and R in the anode circuit. Inmany cases the shape of the pulses are modified so that a suitable change ofmean anode current takes places as the phase of the pulses varies.

A circuit using this principle is shown in figure 3.9(b). The circuit is fedwith negative synchronizing pulses which are differentiated by Cu Rx and Rtto produce a negative pulse corresponding to the leading edge of the linesynchronizing pulses. The negative pulse is followed by a positive pulse(corresponding to the trailing edge of the synchronizing pulse) which islargely removed as the grid of valve Vx is biased positively by Rx and R2 .

Hence, only the negative pulses have any appreciable effect on the anodecurrent and so positive pulses are produced at the anode. These are fed tothe grid G2 of the pentode section of Vx and act as sampling pulses. In thiscase the pulses are fed to G2 instead of to G3 as in figure 3.9(a), but the action is

similar. V2 is the line timebase valve and is connected as a multivibrator.


FIG. 3.9(b). FLYWHEEL SYNCHRONIZING CIRCUIT (Mullard)

C2 is the discharge capacitor being discharged by the pentode section of the

valve. During scan, therefore, the pentode is cut off due to the charge onthe 120pF capacitor and C2 charges through the anode resistor of the pentodesection. During flyback the pentode section conducts and C2 is discharged.

During this time the triode section is cut off and positive pulses corresponding

to the line flyback are produced at the anode of the triode which are fed to

the grid Gj of the pentode section of Vt . These pulses are shown in figure 3.10

and the shaded area indicates the corresponding anode current. It will beseen that as one pulse is moved relative to the other the degree of coincidence

changes and hence the value of the mean anode current, as indicated by the

shaded area. The output of the pentode section of Kt is smoothed by C3 andC4 and then fed to the pentode section of V2 through the charging resistor

R3 .

Suppose that the frequency of the timebase increases so that the pulses

fed to Gi of the pentode section of Kt tend to move more in coincidence

with the pulses fed to the screen grid (i.e. the line synchronizing pulses). Themean anode current will increase (see figure 3.10) and the mean anode voltage

will therefore fall. This means that R3 is returned to a point of lower positive

potential and so it will take longer for the 120pF capacitor in the grid circuit

to discharge. Hence the frequency of the timebase will be decreased, whichwill tend to move the pulses to the normal position as in figure 3.10.

There are a number of circuits based on the use of diodes in a bridge

circuit to form the comparator. The basic method of operation of these is

the same although they differ in detail. The basic circuit is shown in figure 3.11

Pulse fed to screen grid of VI

Pulse fed to CIof pentodesection of VI

Anode current of pentodesection of VI

FIG. 3.10 OPERATION OF CIRCUIT SHOWN IN FIG. 3.9 (b)


FROM SYNCHRONIZINGSEPARATOR

^Mb V2

R2i

-Il-

ea

-»-D

JLJL.

FROMSYNCHRONIZINGSEPARATOR

FIG. 3.11. PHASE DETECTOR USING DIODES

which consists of a bridge composed of Vu V2, Ri and R2 (Ri and R2 are the

same value). The bridge is fed at points A and B with pulses of equal

magnitude in antiphase. These pulses may be obtained from a phase splitter

stage, with equal anode and cathode resistors, or from a suitable centre tapped

transformer. These pulses are generally obtained from the synchronizing

separator but they may be obtained from the line timebase or output stage.

If the pulses at A and B are from the synchronizing separator the waveform

at C is from the timebase and, if the pulses at A and B are derived from the

timebase or output stage, then the waveform at C is obtained from the

synchronizing separator. The waveform at C may be sawtooth or a modified

pulse waveform. Let us assume a sawtooth waveform (as shown in the

figure) as this is easier to understand. This may be obtained from the line

output stage by a suitable shaping circuit, and it should be noted that it is of

reversed polarity to the sawtooth voltage available at the timebase.

The circuit of figure 3.11 is really a clamping circuit. The easiest way to

understand it is to consider that V1 and V2 are switches, which are open

between the pulses fed to A and B but are closed during the time of the pulses.

Thus, point D is connected (through Rt and R2) to point C during the period

of the pulses but disconnected for the remainder of the time. Vx and Vz are

made to conduct and not conduct by the pulses fed to A and B. During the

period of the pulse a current flows in Vt and V2 as shown, charging upcapacitors d and C2 (Ct being equal to C2). Between pulses the charges on

d and C2 bias valves Vx and V2 in such a direction that they are non-

conducting. Ct and C2 will discharge to some extent during this period but,

if the time constants C^.R^ and C2.R2 are large then the loss of charge between

pulses will be small. The exact operation of the circuit is quite involved but

the above is the general principle of its operation.

Assume that the waveform at point C is as shown in figure 3.12. If the

P ^XFIG. 3.12. OPERATION OF CIRCUIT SHOWN IN FIG. 3.11


pulses (commonly known as gating pulses) fed to A and B occur during the

period when the potential at C is zero or almost so {i.e. as indicated by AT in

figure 3.12) no voltage will be fed to point D. If the timebase tends to run

slow so that the sawtooth waveform tends to move to the right the pulses

would occur during the period Y (figure 3.12). During this period the voltage

at C would, therefore, be negative and this negative voltage is fed through

the diodes (and Rt and RJ to the output at D. On the other hand, if the

timebase tends to run fast, the sawtooth waveform will move to the left and

the pulses will occur during the period marked Z. During this period the

voltage C is positive and so a positive voltage is fed to point D. The control

voltage at Z> may then be fed to a reactance valve to control the frequency

of an L-C oscillator or to the charging resistor of a multivibrator timebase

[as in figure 3.9(b) ] or, to the charging resistor ofa blocking oscillator timebase.

In the case where the control voltage is fed to the timebase an amplifier and

phase reversing valve will be required. A circuit that uses this is shown

below in figure 3.13. Valve Vt forms the phase splitting valve being fed with

-H.T.+ H.T.+

470K

FIG. 3.13. FLYWHEEL SYNCHRONIZING CIRCUIT USING DIODES

negative synchronizing pulses. Ct and C2 correspond to C\ and C2 offigure 3.11 and the two 150-kO resistors correspond to Ri and R2 of the samefigure. The output voltage across Ri (figure 3.13) is smoothed by C5 and fed

to the amplifier valve K4 . As the current in VA is varied, the voltage across

R( also varies. This voltage is used to feed the charging resistor of a blocking

oscillator type of timebase.

It has been shown that if the timebase tends to run slow the output is

negative. This will {a) reduce the current in K4 and increase the voltage

across i?4 ; (6) increase the voltage fed to the charging resistor, hence increasing

the frequency of the timebase, and bringing it into step again.

This type of circuit is very commonly used but has many variations whichall operate in the same way:

(0 The diode valves are usually replaced by semiconductor diodes in

modern receivers.

(h) Instead of using a phase splitter valve Vx this is replaced by a centre

tapped transformer.


(hi) In place of the blocking oscillator timebase a multivibrator timebasemay be used.

(iv) The amplifier valve may be omitted.

(v) Instead of the sawtooth waveform being fed to point C (figure 3.11)

a modified pulse waveform may be used and, so long as there is a sloping

edge to the pulse such as P-Q (figure 3.12), the circuit will operate in a similar

manner.Another diode circuit commonly used is shown in figure 3.14. This has

the advantage that a push-pull input is not required nor does it use a centre-

tapped transformer. However, it is not easy to explain how the circuit operates.

SW e

<—R«i

> '

AII

B;

1

1

1

-^

II

IT c'

R«2-L

<2

< —

—

-*

c2 "OUTPUT

D

FIG. 3.14. DIODE PHASE COMPARATOR

Line pulses (and, of course, field pulses since the composite pulse waveformis used) are fed to point A, these being negative-going and obtained from the

anode of the synchronizing separator. A line sawtooth waveform is normally

fed to point C this being obtained from the line output stage although other

waveforms may be used. The output voltage to control the timebase is taken

from across capacitor C2 . Capacitor C2 is much larger than Ct—about 50

times. The time-constants C\ Rx and d R2 (Rx normally being equal to R2)

are of the order of 30ju.S. The actual values of the components may vary

(particularly between valve and transistor circuits) but the relationship between

Ci and C2 and the time-constants are maintained approximately the same.

Suppose that the normal voltage at A is + 100V and that this reduces to zero

voltage during the period of the line pulse. The actual values of voltage are

not important but the explanation is simplified if some values are assumed.

First, suppose that there is no voltage between C and D and that there is

a low resistance circuit between them. Immediately after a pulse, point Arises to + 100V and hence Ct charges, the current flowing in Rx and R2 since

the flow of current is in the non-conducting direction of the two rectifiers

Rel and Re2 . The direction of the current flow is shown in figure 3.15(a).

Since the time-constant Cx Rx and Cx R2 is only 30/aS, capacitor Cx will almost

fully charge during the periods between line pulses (90ju.S for 405 lines and60^iS for 625 lines). Some current will flow into capacitor C2 through R2

but the time-constant is much greater and hence the voltage across it will be

small. Suppose that Cx has charged to 90 volts then the voltages would beapproximate as in figure 3.15(b).

At the start of the line pulse, point A is reduced to zero voltage and the

initial condition is shown in figure 3.15(c) since the voltages across the


«0

i

R«,

I-— Bl

—

»

c

:+r,

~~•" II 1

AC| <

R«2T

1

I

C2 B

;J"2

>UT

D^ Q

—

°c

R«,

Z. I R|

+CI -B

-+IOVA II

R«a'3~

| "2

+ 2V^-^''i J

+2V Jc2

D

0>>

R«,

HK*«2.

E

C2J—>r

't I"2

/

-OUTPUT

/D

FIG. 3.1 S. OPERATION OF PHASE COMPARATOR(a) initial charging current

(b) Condition just before line pulse

(c) Condition during pulse


capacitors cannot change suddenly. A. current will not flow in Rel and Re2as shown, and will discharge the capacitor Ct . Since Rel and Re2 are con-ducting during this time, points C, B and E are effectively joined togetherand hence the voltage across C2 rapidly becomes that of point C. Due tothe low resistance of Rel and R&, Ci is rapidly discharged until the voltageacross R^ tends to reverse. Thus in this case the voltage at B and thereforethat across C2 becomes the same as point C, i.e. zero voltage. Between pulsesthe action is as in figure 3.15(a) and C2 is charged slightly positive.

Suppose that point C is at a voltage of +5V during the period of theline pulses. Again, points B and C can be considered as connected togetherby Rel and, therefore, Ct discharges not to zero but to +5V as shown infigure 3.15(d) [it really discharges to zero and then charges in the other direc-

tion to 5 volts]. C2 will also have a voltage of + 5V across it, this rise from+2V of figure 3.15(b) being obtained by charging during the period betweenpulses. The voltage output is again the same as that at point C. If point Cis at —5V then Q only discharges to — 5V as in figure 3.15(e). C2 will nowbe discharged first to zero and then charged to —5V by the current in Rt2 .

Again, the output voltage is the same as that at point C.

-•

—

+ 5V C

*"2

'-I1

Rl

~ll* >•*.—+SV

A II b'

R«2_ft

1R2

1

C2 B

+ SY

+

5 5VD

(<0

Hih B

R«2_

:

Ri

5v+

(«)

FIG. 3.15. OPERATION OF PHASE COMPARATOR(d) Condition during pulse with point C at +5V(e) Condition during puise with point C at —5V


Suppose now that a line frequency sawtooth waveform is fed to point C(this having no d.c. component, i.e. equal positive and negative portions).

The voltage output during the line pulse period will depend on the voltage

at C during this period, which depends on the phase of the sawtooth voltage

relative to the line pulse. (This is shown in figure 3.12 and has already been

explained).

The operation is complicated by the fact that between pulses the voltage

at C changes but since there is no d.c. component to this voltage there is noresultant voltage across C2 . The mean voltage across C2 between line pulses

depends on the voltage on C2 during the period of the pulse and it has been

shown that this voltage depends on the phase of the line timebase waveform.Thus, if the voltage across C2 is smoothed by a C-R network, the resultant

voltage will depend on the phase of the line timebase. Hence this voltage can

be used to vary the frequency of the line timebase, which may be a blocking

oscillator or multivibrator timebase, and the voltage may be fed directly or

through an amplifier.

Variations of this circuit occur and one is shown in figure 3.16. In this

circuit point B is always discharged to zero during the period of line pulses

by the current in Re2 - Similarly, point E is brought to zero voltage by the

/-V1

FIG. 3.16. MODIFIED DIODE COMPARATOR CIRCUIT

current in Rel . This means that the waveform is clamped to zero at the instant

of the line pulse. Thus the position of the waveform relative to the zero line

depends on the phase of the pulse relative to the sawtooth waveform. Thisis shown overleaf in figure 3.17. After smoothing by R3 and C3 the resultant

voltage will depend on the voltage across C2.

Some means must be used to vary the line timebase frequency in the formof a line hold control. It may be done in this circuit by returning point Fto a source of variable voltage (potential divider across the h.t.) instead ofto h.t. negative.

Although flywheel synchronizing has advantages from the viewpoint ofpicture quality it has its disadvantages and there are certain difficulties thathavenot been mentioned. It obviously complicates the receiver and has little if anyadvantage when used in an area of good signal strength free from serious inter-

ference. Some older receivers have arrangements to use either flywheel or di-

rect synchronizing. One disadvantage of flywheel synchronizing is that when the

set is switched on and ifthe frequency of the oscillator or timebase (whichever is

used in the flywheel synchronizing circuit) is far from that of the synchronizing


LINEPULSES

POSITIVEOUTPUT

FIG. 3.17. OPERATION OF CIRCUIT OF FIGURE 3.16

pulses, it will often be found that the circuit will not pull into step, necessitating

adjustment of the frequency or hold control to get the line timebase tosynchronize. In some receivers a press button is fitted which shorts theoutput of the phase detector to facilitate adjustment of the L-C oscillator ortimebase. The reason why the circuit will not pull into step (if the frequency

is not nearly correct) is that all the comparators described are phase detectors

and do not respond directly to a frequency difference. Many circuits (those

using two sets of pulses: one from the synchronizing separator and the otherfrom the timebase) will only operate over a very limited phase difference,

i.e. when there is some overlap between the two sets of pulses. If the phasedifference exceeds this then there is no output from the phase detector and nocorrecting signal. If the frequency of the two sets of pulses is not the samethen the two sets of pulses will, of course, come into coincidence at someinstant but, unless the frequency correction is rapid, they will come out ofcoincidence again before the frequency has been corrected and the correcting

signal will then fall to zero. In this way a short correcting signal is obtainedas the pulses are in coincidence but this never has time to correct the frequencyof the oscillator or timebase and, therefore, the circuit never locks itself into

step, it being necessary to make a manual adjustment to bring it into step.

Once it is in step it is usually possible to make quite large changes in the

setting of the manual control before it pulls out of step. In other words, it

will hold over quite a large range but should it once come out of step for anyreason will not pull itself into step again. This difficulty is most importantwith multichannel receivers as there is the possibility of it coming out ofstep at each change of channel. There are two possible ways of reducing this

trouble: (0 by arranging that the oscillator or timebase drifts slowly throughthe required frequency range when the receiver is switched on, or when achange of channel is made; and (it) by arranging that the comparator operatesas a frequency comparator until the frequency is nearly correct and thenchanges over to a phase comparator.

Both arrangements are complicated and do not appear to have been usedin commercial receivers.


Another difficulty arises during the field pulse period when the pulses

are at double line frequency (i.e. five or eight half line pulses). The phase com-parator does not operate correctly under these conditions and the correcting

signal is not of correct magnitude. The exact effect depends on the type ofphase comparator. This means that the circuit must be designed either

with a short pull-in time so that it will pull back into step after the field

pulses have occurred and be in correct phase by the time the picture signal

starts. Alternatively, it must be made with sufficient inertia to run the lines

during which the field pulses occur without coming out of phase by anappreciable amount. From the point of view of interference the inertia shouldbe large but this causes difficulties with the pulling into step when first

switched on. The pull-in range, as it is called, is decreased if the inertia andsmoothing of the circuit are increased. Owing to this the inertia given to the

circuit must be a compromise between these two factors.

CHAPTER 4

TELEVISION CAMERAS ANDSTUDIO EQUIPMENT

InVolume 1 mention was made of mechanical scanning arrangements

at the transmitter. Mechanical arrangements are not used for highdefinition television so the various mechanical scanning arrangements

will not be considered and only electronic scanning systems and cameras will

be described.

FLYING SPOT SCANNINGThe basic arrangement of this method of scanning is shown in figure 4.1.

On the screen of the cathode ray tube T is produced a raster that is un-modulated. This raster is focused on to the film, or slide, F by the lens L x .

FIG. 4.1. FLYING SPOTSCANNING SYSTEM

Thus, as the spot traces out the raster on the screen of the tube T, the

corresponding image of the spot scans the picture on the film F. The light

passing through the film will vary as the density of the film varies over the

picture. This light is collected by lens L2 and falls on to a photoelectric

multiplier P, which produces a signal output proportional to the light falling

on it. Thus, the electrical output of P will depend on the density of the

particular element of the picture being scanned at any instant. In this waythe electrical output of P is a signal corresponding to the picture on the film.

To obtain sufficient light to operate the photomultiplier P (for details of

electron multipliers see page 58), the light output of the cathode ray tube Tshould be high, which means operating the tube at a high e.h.t. voltage. It is


necessary that the focus of this tube is extremely good if high definition is to

be obtained. Also it is necessary that the afterglow time {i.e. the time the

screen glows after the beam has been cut off, or has passed from one position

to another on the screen) is extremely short—say a fraction of a microsecond.If this is not so, then instead of obtaining the light from a single element of the

picture, we also obtain light from other elements which have just been scannedwhich upsets the definition. The effect is similar to a poor high frequency

response in a video amplifier. Even with special tubes that have an extremely

short afterglow a correction has to be made for it in the amplifier circuits.

This system may be used for stationary slides or for moving film. Withfilm it is necessary to move the film from one frame of the film (i.e. one picture)

to the next in the time of the flyback of the spot between television fields

(about 14 lines or 1400/jS on 405 lines). This is difficult as the time is so

short and a special fast pull-down mechanism must be used. This fast pull-

down is only practicable with 16mm film. Other methods must be usedfor scanning 35mm film, which cannot be considered here.

The arrangement has the advantages of simplicity and not requiring anexpensive camera tube. Owing to the limited light output from the cathoderay tube the system is normally limited to transparent pictures of small size.

It is possible to use opaque objects in which case photoelectric multipliers are

placed so that they pick up the reflected light from the object. Owing to the

fact that the reflected light is spread out in almost all directions several

photoelectric multipliers are used but, even so, they can only pick up a small

fraction of the already small amount of light. This arrangement therefore

can only be used for small scenes; it has many disadvantages and has never

been used other than experimentally.

ICONOSCOPE OR EMITRON CAMERA TUBEOne disadvantage of the mechanical system of scanning which was

mentioned in Volume 1 was the lack of light on the photoelectric cell. This

was largely due to the fact that the Nipkow disc only allows the light to pass

to the photoelectric cell from one element at a time. If there are 100,000

elements then the amount of light passing to the photoelectric cell cannot begreater than l/100,000th part of the light picked up by the lens. In other

words, we are only using the light from an element of the scene for

l/100,000th part of the total time. The ideal solution would be to use the

light for all the time and store up the effect of the light over the wholescanning period. This could be done by using (say) 100,000 photoelectric cells

spread over the image of the picture, using the output of the cells to charge

100,000 capacitors and then connecting to these capacitors in turn. In this

way we have utilized the whole of the light energy and converted it into acorresponding charge on the capacitor. By doing this we should be able to

make the device 100,000 times more sensitive. Although this may soundquite impracticable, the iconoscope and modern camera tubes effectively act

in this way. It is this storage effect which has enabled modern camera tubesto be so sensitive.

A camera tube is an extremely complex device and the descriptions that

follow are simplified as are many of the diagrams in order that the principles

may be made clearer. This should, therefore, be borne in mind when reading

the sections dealing with these camera tubes.

The iconoscope or Emitron camera tube was developed during the early

1930s and was used for the television cameras in this country between 1936and 1939, and for a short time after the war. It is now no longer used as better

camera tubes are available, but the basic principles are the same. The general

arrangement is shown in figure 4.2(a). The tube consists of a spherical glass

TELEVISION CAMERAS 49

FIG. 4.2(»)

ICONOSCOPE CAMERA TUBE

bulb B about 7* diameter with a side tube which contains an electron gun G.

Deflecting coils D fit on this side tube, as shown. The gun G is electrostatically

focused and operates with a final anode voltage of about 1500V. In principle

it is similar to the cathode ray tube used in oscilloscopes but differs in detail.

The deflecting arrangements are basically the same as those of a television

cathode ray tube. The electron beam falls on the mosaic M which it scans

by suitable currents in the deflecting coils D. The spot is deflected horizontally

by the line timebase and vertically by the field timebase. The arrangement is

more complicated than that of a cathode ray tube. If constant amplitude line

scanning is used, the area scanned on the mosaic will be trapezium shaped, dueto the fact that the gun is at an angle to the mosaic. For the scanning area to

be rectangular the amplitude of the line scanning must vary with the field

scanning current: this can be arranged by a suitable electronic circuit. Anoptical image of the scene to be televised is focused by the lens L on to the

mosaic M. To avoid the image being distorted by the spherical glass bulb apolished glass window W forms part of the bulb B.

The mosaic M consists of a thin mica sheet about 5* x 4", which is coated

on the right-hand side with a conducting metallic coating, and is known as

the signal plate. A connection is brought out from this plate. The other side

of the mica sheet is coated with a very thin layer of silver. The mica is then

heated so that the silver layer breaks up into an extremely large number of

"islands", which are electrically insulated from each other. The silver is then

oxidized and coated (by evaporation in vacuum) with a minute layer of

caesium. This makes the silver islands photosensitive, i.e. they will emit

electrons when light falls on them. An enlarged cross-sectional view is shownin figure 4.2(b). Each island is like the cathode of a photoelectric cell andthere is a capacitance between each island and the signal plate. Thus

MICA SHEET

FIG. 4.2(b) MOSAIC

©ONNECTION

essentially we have a large number of photoelectric cells connected to

capacitors. The anode of these little photoelectric cells is formed by a coating

on the inside of the bulb B, which is connected to the final anode of the gun G


The mechanism of operation is complicated but will first be considered

in a simplified way. When light falls on the islands they will emit electrons

which will be collected by the final anode of the gun G [see figure 4.2(c) ].

Thus, as the islands lose electrons they charge up the capacitance formed

between themselves and the signal plate, the islands becoming charged

MOSAIC

SIGNAL PLATE

FIG. 4.2(c) CONNECTION OFMOSAIC AND SIGNAL PLATE

©OUTPUT

positively. The amount that each capacitance is charged will depend uponthe amount of light falling on that particular island because the more light

falling on the island the larger will be the number of electrons emitted. This

action goes on continuously. At the same time the electron beam is made to

scan the mosaic and as it passes over the islands it discharges them, i.e. it

supplies them with electrons to make up for those lost by photoelectric

emission between each scan. This discharging current must also flow in the

other plate of the capacitance, in this case the signal plate, and the current

is made to pass through the load resistor R, as shown in figure 4.2(c). Thus,

a voltage is produced across R as the beam moves across the mosaic. This

will be proportional to the charge on the islands which is approximately

proportional to the light falling on the particular portion of the mosaic. Thus,

a signal is produced across R corresponding to the image on the mosaic. Since

the "islands" are losing electrons all the time the charge on the "capacitors"

is relatively large and the device should therefore be much more sensitive

than the mechanical arrangements, i.e. it uses the storage effect which is so

important.

However, the mechanism of operation is much more complex and the

camera tube is less efficient than might appear from this simple explanation.

The exact operation is so involved that we cannot go fully into the details.

It is, however, important to say a little more about its operation to understand

the reason for the method of operation of the more modern camera tubes.

The simple explanation of the tube is upset by secondary emission. When a

high velocity beam (the velocity is high because the final anode voltage of the

gun G is high and the signal plate is connected through R to the anode) of

electrons strikes an electrode (such as the islands on the mosaic) secondary

electrons are emitted in a larger number than the primary electrons, i.e. those

of the beam. In a typical case seven secondary electrons may be emitted for

each primary electron. The effect of this is that, when the beam falls on the

islands it does not charge them in a negative direction (or cancel out the

positive charge) by collection of electrons, but actually charges them positively

because secondary electrons leaving an island exceed the number arriving onit. After the island has been scanned it may have a potential of a few volts

positive with respect to the final anode. Suppose that the first line has been

scanned and the beam is travelling along the second line causing the liberation

of secondary electrons. Since the islands of the first line are more positive

than the anode, electrons are attracted to these islands rather than to the


anode. Thus, the islands that have just been scanned tend to collect electrons

and so become negatively charged. The action of light as as before: those

islands with light falling on them will lose electrons and are, therefore, less

negatively charged than those with no light. It can be shown that the effect

of these large numbers of secondary electrons is to decrease the sensitivity;

the storage time is not a whole field period but actually only about l/10th

of the field time.

Probably the most troublesome result of this secondary emission is the

production of spurious signals. It is found that an output is obtained from

the tube even when there is no picture focused on to the mosaic. This is due

to the fact that the conditions just after the start of a field are different

from those at the end. At the start there are no islands which have just been

scanned near to these being scanned; as the beam reaches the end of a field,

however, there are a lot of islands which have just been scanned. Thus, the

distribution and collection of electrons by the islands will be different in the

two cases. A similar result applies to the line scan. These spurious signals

occur both at line and field frequencies and the effect on the picture is to

darken the top left-hand portion, as if it had been shaded from the light.

In practice, correcting signals have to be produced (called bend and tilt

signals) which can be fed into the video amplifier to cancel out the spurious

signals. The exact nature of the spurious signals depends on the picture

content and so continual adjustment of these correcting, or shading signals

as they are called, is necessary.

The iconoscope has the following disadvantages:

(0 The sensitivity, although much higher than that of mechanical systems,

is low compared with later camera tubes.

(if) The spurious signals cause serious trouble and require continual

correction. (They are particularly bad if the scene being televised is poorly lit).

(iii-

) Due to the spurious signals there is no definite black level from the

camera, i.e. there is no portion of the output waveform which can definitely

be related to black.

(iv) Owing to the long distance between the mosaic and the bulb on the

lens side of the tube, the lens must have a long focal length. This means an

expensive lens with a small depth of focus and renders the use of a lens turret

almost impossible.

(v) The tube is large and of an inconvenient shape to fit into a camera,

(vi) Trapezium distortion correction (or keystone correction as it is some-

times called) is required in the deflecting circuits.

The advantages are:

(0 It is stable at all light values.

00 Owing to the fact that the storage time is only a portion of the field

period it produces sharp images of rapidly moving objects. (Similar to a

high speed shutter on a camera).

The next development was the image iconoscope or Super Emitron. Oneof the difficulties in the manufacture of the iconoscope is that, if sufficient

caesium is evaporated on to the mosaic to get maximum photosensitivity, the

insulation between the islands is poor and the image becomes blurred. The

amount of caesium must therefore be a compromise between these two

conflicting requirements. The idea of the image iconoscope is to separate the

action of photoemission from the action of the mosaic.

The general principle is shown in figure 4.3. An optical image is focused

on to the photocathode P. This is now a transparent photocathode, so that

electrons are emitted from the inner surface corresponding to the illumination

on the other side. The optical image is produced on the photocathode by

the lens L. The electrons from the photocathode are focused by the magnetic

focus coil (or electrostatic focusing may be used) and accelerated towards the

TELEVISION SERVICING52

mosaic. At the mosaic the electrons from each section of the photocathodecome to a focus and we have an electron image of the optical image on the

,-PHOTO CATHODEP

MOSAIC

OUTPUT

FIG. 4.3.

' GUN §-I500V

IMAGE ICONOSCOPE

photocathode. Thus, the electrons reaching a particular portion of the

mosaic correspond to those leaving a similar point on the photocathode. Toprevent distortion of the image it is necessary to carefully screen this section

of the camera tube from the deflecting coils D on the neck of the tube. Themosaic is not now photoemissive and may consist of individual conducting

islands all insulated from each other, or it may be just a mica sheet with a

signal plate on the right-hand side. A voltage of 500-1000V is maintained

between the photocathode and the mosaic (which is positive) so that the

electrons arrive at the mosaic with a high velocity. As a result of this they will

produce secondary electrons and the islands (or portions of the surface of

the mica card) will charge up in a positive direction by an amount depending

on the number of primary electrons. They will charge in a positive direction

because the number of secondary electrons will be greater than the numberof primary electrons. The result is that the islands are charged positively bythe action of the primary electrons in a similar way to the action of light in

a normal iconoscope. The action of the scanning beam is similar to that of

the iconoscope and a signal output is obtained from the signal plate.

Since the photocathode is separate from the mosaic a more efficient

photocathode can be produced (an improvement of three to four times) since

it will be remembered that the photosensitivity of the iconoscope is limited

by the reduction of the insulation resistance between the islands. For the

same reason the insulation of the islands on the mosaic will be high, as they

are not caesium coated. Also, since each primary electron from the photo-

cathode will liberate (say) five electrons from the mosaic (by secondaryemission) the sensitivity will be improved by (say) five times. These factors

increase the sensitivity by about 20-25 times for a given lens.

In practice the same lens is not used but one of a shorter focal length as

this is more convenient. On the other hand the amount of light collected bya lens of given "f" number decreases as the focal length is reduced. Thus the

overall sensitivity of the tube is reduced. In practice the amount of light is

reduced to about a quarter by the change of lens to one of shorter focal length

so that the overall sensitivity over an iconoscope is increased by a factor 25/4

or 6 times.

Owing to the higher initial velocity of the secondary electrons emitted

from the mosaic (compared with the very low velocity of photoemission in

the iconoscope) the shading signals are less but are still present and important.


Due to the different arrangement of the image iconoscope the lens can have a

smaller focal length (the photocathode is also smaller than the mosaic, the

electron image on the mosaic being greater than the image on the photo-

cathode) which results in a greater depth of focus (for a given aperture) and it

is easier to use lenses of different focal lengths in a lens turret.

The image iconoscope has all the advantages of the iconoscope and fewer

disadvantages.

The advantages compared with an iconoscope are:

(0 The sensitivity is greater.

(if) The optical system is easier to arrange.

(iii) It can be made smaller than the iconoscope.

THE PHOTTCON (OR PHOTOELECTRON STABILIZED IMAGEICONOSCOPE)

The main disadvantage of the image iconoscope is the presence of spurious

signals. These are overcome in a Photicon, which is a modified form of

image iconoscope. In this tube the construction is similar to that of the

image iconoscope but two strips are placed near to the mosaic along the twoedges which cause the shading or spurious signals. They are shown in

figure 4.4. Each strip is connected to a source of positive voltage (a few volts

only) and this is variable. The inner surface of the tube around the mosaic

MOSAIC

FIG. 4.4. STRIP ELECTRODES TO REDUCE SHADING

is coated to form a photocathode and this is illuminated by small lampsround the camera tube. This light causes electrons to be emitted by the

photocathode coating which fall on the mosaic. The distribution of these

electrons can be controlled by varying the voltages applied to the two strips.

Increasing the potential will cause more electrons to be collected by the strips.

Spurious signals are caused by uneven distribution of electrons due to the

actions of the scanning beam, and these other electrons are used to even upthis distribution and so eliminate the spurious signals.

The use of this principle of photoelectron stabilization results in a great

improvement on the picture and largely removes the spurious signals. Forthis reason there is now a definite black level voltage in the output.

This type of tube is no longer used.

THE MONOSCOPEAlthough this device is not a camera tube it is convenient to mention it

here. It consists essentially of a cathode ray tube except that the normal

fluorescent screen is replaced by an aluminium plate. On the plate is printed,

in a pigment with a low secondary emission, the pattern or photograph whichit is required to reproduce. The plate is maintained at about 30V negative

with respect to the anode so that all the secondary electrons are collected by

the anode. As the plate is scanned the current in the plate lead (which is

rather similar to the signal plate of the iconoscope) varies as the secondary

emission varies and a signal output is produced.

Obviously, only one pattern or photograph can be produced from a single

monoscope but the device is useful for generating test signals (such as Test


Cards) and certain captions. It is relatively simple to operate and it producesa reasonably large output voltage.

SECONDARY EMISSIONBefore dealing with other types of camera tubes it is necessary to say a

little about secondary emission. If we bombard a plate with primary electrons

and measure the ratio of secondary to primary electrons, we should obtain acurve similar to that shown in figure 4.5(a). (The exact shape and scales will

depend on the material but the general conclusions will be the same). At lowenergies of the primary electrons (i.e. low velocities due to low accelerating

voltages) the secondary emission ratio is less than unity. This is to be expected

as the primary electrons have little energy which can be given to other

PRIMARY ELECTRON ENERGY(or accelerating voltage) ©

CATHODE

ANODEX

^ELECTRONBEAM/

| |

cA P

(©FIG. 4.5(a). SECONDARY EMISSION RATIO

(b). BOMBARDMENT OF INSULATED PLATE P

electrons to cause secondary emission. As the accelerating voltage is increased

the secondary emission ratio increases until it is unity at point X. Furtherincrease in accelerating voltage increases the secondary emission ratio still

further as would be expected, but it reaches a maximum value at point Z.As we now increase the accelerating voltage the secondary emission ratio

decreases, due to the fact that the electrons penetrate too far into the metaland so give up the energy to electrons well below the surface. These electrons

lose their energy by collision with the atoms of the material before theyreach the surface, and they therefore do not have sufficient energy left for

emission to take place. At point Y the secondary emission ratio is againunity and at higher accelerating voltages the ratio becomes less than unity.

Let us now consider an insulated plate P (figure 4.5(b) ) which is

bombarded with primary electrons from a simple gun formed by the cathodeC and the anode A. Suppose that the voltage between the anode A and thecathode C (i.«. the accelerating voltage) is greater than that corresponding to


point X and less than that at Y. Suppose, also, that the plate P starts at a

potential somewhat less than that of the anode but above that of point X.

The action of the beam is to produce more secondary electrons than primary

electrons (since the ratio is greater than unity) and the secondary electrons

will be collected by the positive anode A. Thus, P loses electrons and so

charges up in a positive direction. This action continues but the plate P will

never charge up appreciably more positive than the potential of the anode A.

If it did, secondary electrons would not be attracted to A but would return

to the plate P (which is the more positive) and plate P would now collect

electrons and therefore charge up in a negative direction. Thus, P mustcharge to an equilibrium potential such that the same number of secondary

electrons leave P and are collected by the anode A, as there are primary

electrons falling on P. This means that P will be only slightly more positive

than that of the anode A. This is what occurs in a normal (non-aluminized)

cathode ray tube where P is the fluorescent screen. This is the action that also

takes place in the iconoscope and image iconoscope, the potential of the

mosaic being approximately that of the anode of the scanning gun.

Assume now that the plate starts at a potential less than that corresponding

to point X. Although the anode voltage may be higher than that corresponding

to point X, so long as the potential of the plate is less than that of point X,

then electrons will arrive at the plate with a velocity corresponding to the

potential of the plate. They will be accelerated by the anode (which may have

a potential well above X and hence the electrons will have a high velocity bythe time they reach the anode) but they will then be decelerated by the field

between the anode and the plate P (note that this field is in the direction of

plate to anode and hence against the direction of electron travel). When the

electrons fall on the plate since the potential is below that of point X, there are

fewer secondary electrons than primary electrons and the plate therefore

collects electrons. Thus, the plate charges in a negative direction and, as it

does so, the velocity of the electrons reaching the plate is reduced until, whenit charges to the potential of the cathode, the electrons will be slowed downand only just reach it. This is because the deceleration produced between the

anode and the plate P is equal to the acceleration between the cathode C andthe anode A. If the plate were charged to a lower potential than the cathode,

the electrons would not reach it and they would return to the anode. There-

fore, the plate cannot go below the potential of the cathode. On the other

hand it cannot be charged positively by the beam. Should it acquire a positive

charge (less than that corresponding to X) then the beam will charge it back to

cathode potential—in other words the plate is stabilized at cathode potential.

This is the principle used in modern camera tubes which are referred to as

low velocity or cathode stabilized camera tubes. It is the initial potential of the

plate which is important, rather than the accelerating voltage. Even if the

accelerating voltage is greater than that corresponding to point X, so long as

the initial potential is below X it will be charged to cathode potential. It maybe interesting to note that if a cathode ray tube is operated at a low final

anode potential, the fluorescent screen may charge in this way to cathode

potential and no spot will be visible. When tubes are used at low voltages this

sometimes occurs over a portion of the screen. It should be noted that if the

accelerating voltage is greater than that corresponding to point Y the plate

will never exceed the potential corresponding to Y. If it did it would collect

electrons, since the secondary emission ratio above Y is less than unity andthe plate would charge in a negative direction until it reached the potential Y.

ORTHICON CAMERA TUBEThe first low velocity tube was the orthicon which is shown in simple form

in figure 4.6. The electron gun now has a final potential of only about 100V.

The electron beam is focused by a solenoid S which extends the whole length

56 TELEVISION SERVICINGELECTRON BEAM

I ^ P .— ELECTRONGUN

MFIG. 4.6. ORTH1CON CAMERA TUBE

of the tube. The beam is deflected horizontally (line scan) by electrostatic

deflecting plates P and vertically (field scan) by deflecting coils C. Althoughit might at first be thought that there is nothing unusual about the method offocus and deflection as they might appear similar to those used in a cathode

ray tube, this is not the case. During the travel between the plates P the beamis influenced by the electrostatic field between the plates and by the magnetic

field produced by the solenoid 5. This causes the beam to be deflected in the

direction of the plates (at right angles to that which occurs in a normalcathode ray tube). Further, it is only deflected during its travel between the

plates and emerges in a direction parallel to the original direction of travel

(i.e. along the axis of the tube). Similarly, the effect of the deflecting coils Cis to cause deflection in a direction at right angles to that obtained in a normalcathode ray tube and, again, the beam leaves the coils travelling in a direction

parallel to the axis of the tube. Thus, the beam reaches the mosaic M at

right angles to it whatever the position of the beam—which is essential.

The target now consists of a transparent conducting signal plate S on the oneside, and a transparent photosensitive mosaic M on the other side.

The tube now operates in a similar manner to the simple explanation of

the iconoscope given earlier. With no light on the mosaic it is charged to

cathode potential, so that the electron beam just does not reach the mosaic

and the electrons are collected by the anode which is extended as a conducting

coating down the sides of the tube. When light falls on a portion of the

mosaic, photoelectrons are emitted and are attracted to the anode (which is

now about 100V positive with respect to the mosaic) and so this portion ofthe mosaic becomes charged positively. When the beam falls on this section

it charges it back to cathode potential and the sudden change of potential

results in a signal output from the signal plate, the signal plate being connected

to the cathode of the gun through a load resistor.

The focusing of the beam in this type of tube is much more complex thanmight be imagined. Most of the electrons in the beam travel in a helical path

but they all come to a focus at a number of points along the length of the

tube, and one of these points is made to correspond with the target (i.e.

the mosaic) by adjustment of the focusing field. Details of focus and the

deflection systems cannot be discussed in this book.Since no secondary emission now occurs there are no spurious signals

such as those produced in the iconoscope and image iconoscope. Owing to

the fact that there is a strong electrostatic field away from the mosaic(towards the anode) all the photoelectrons are collected by the anode, the

storage effect and efficiency being high. This is not the case with the

iconoscope, since the mosaic is at a similar potential to that of the final anode.

The tube has a definite output corresponding to black since there are noappreciable spurious signals.

The sensitivity of the tube is between that of the iconoscope and the

image iconoscope. Loss of sensitivity occurs owing to the difficulty of making


the transparent photomosaic. This early form of orthicon tube is not nowused.

C.P.S. EMITRON CAMERA TUBE (Cathode Potential Stabilized Emitron)This tube is the modern form of the orthicon and operates in a similar

manner to that just described. The basic construction is shown in figure 4.7.

MOSAICELECTRON BEAM

ELECTRON GUN C(cathode at earth

potential)

FIG. 4.7. C.P.S. EMITRON CAMERA TUBE

As before, the beam is focused by a long solenoid S but it is now deflectedmagnetically in both directions, by deflecting coils C, and Q which are atright angles to each other. The target consists of a transparent signal platewhich only absorbs about 10 to 20 per cent, of the light, and the mosaic iscomposed of antimony islands which are made photosensitive with caesiumThis mosaic must, ofcourse, be transparent. Now, antimony will not divide intoislands when heated as in the case of silver and the mosaic is produced by evapo-rating the antimony (in the evacuated tube) on to the glass or mica, which formsthe target, through a mesh with about 1000 meshes per linear inch [i.e. 1,000 000holes per square inch). This mesh is removed after coating the target. 'Allthis has to be done in a high vacuum and ingenious ways have been developedto keep the mesh in place and to remove it without breaking the vacuumThe sensitivity of the photocathode is about 10 to 15 times greater than thatof the orthicon and 2 to 3 times greater than that of the iconoscope. Toprevent an ion burn being produced due to positive ions travelling to themosaic, a fine screen St is placed in front of the mosaic at a suitable distancefrom it (actually at a point where the beam is not in focus). This mesh hasabout 200 holes per linear inch.

This tube is unstable (as is a normal orthicon) if excessive light falls onthe mosaic because the mosaic charges up sufiiciently in a positive directionto accelerate the beam sufiiciently to cause the secondary emission ratio toexceed unity, i.e. beyond point .Tin figure 4.5(a), this voltage being about 13 V.Once this occurs the mosaic charges up to anode potential and the wholeaction fails. This may be prevented by placing a fine mesh (actually that usedto produce the mosaic) about 1mm from the mosaic and maintaining thisat + 13V. In this way the mosaic is prevented from exceeding this voltage(corresponding to point X) and the instability cannot occur.

This type of tube has the following advantages:

(i) Definite black level, (fl) No spurious signals.


(fit) Convenient shape and causes no difficulty with the design of the

optical system.

(iv) Sensitivity and resolution much better than the iconoscope and image

iconoscope.

IMAGE ORTHICON CAMERA TUBEThe image orthicon uses an electron multiplier to increase its sensitivity.

The principle of an electron multiplier is shown in figure 4.8. Electrodes

shaped in the form of Venetian blinds as shown, are connected to increasing

positive voltages. Assume that a primary electron falls on the first electrode.

This may produce (say) three secondary electrons which will be attracted to

the second electrode. Here, each of the three electrons may produce three

more electrons and thus nine electrons are collected by the anode A. A fine

mesh is placed on the left-hand side of the Venetian blind structure so that

there is an accelerating field to accelerate the secondary electrons to the next

electrode. Secondary multipliers of this type are commonly used to increase

the sensitivity of photoelectric cells, the primary electrons being produced

V \1 /%----—

1

/V \

FIG. 4.8.1 2 A

PRINCIPLE OF ELECTRON MULTIPLIER + IOOV + 20OV +3QOV

from a photocathode. An amplification of 1,000,000 times or more may be

obtained in this way, the device being known as a photomultiplier.

In this type of camera tube a gain of 100 to 1000 times is obtained from the

secondary emission multiplier section of the tube which consists of five stages

of secondary emission. A simplified diagram of the image orthicon camera

tube is shown in figure 4.9. The beam is produced by the gun G and, as

previously, this is a low velocity beam, the wall coating W which forms the

IMAGE ACCELERATOR•200V

ELECTRON GUN Gcathode at earth

potential)(«'

FIG. 4.9. IMAGE ORTHICON CAMERA TUBE

anode being at + 300V approximately. The beam is focused by solenoid S


and deflected by coils Cx and C2 as in the C.P.S. Emitron. Similar to the

image iconoscope the photocathode and the target are separated. P is a

plain transparent photocathode and an electron image is focused on to the

target T by a suitable electrostatic field between the photocathode and the

target. The target T consists of a thin glass plate about 0-0002' thick with

a fine mesh M (500 meshes per linear inch) about 0001* in front of it. Theelectrons from the photocathode bombard the target and cause secondary

emission, since they are accelerated by a potential of about 500V. These

secondary electrons are collected by the mesh M which is maintained at

about + IV with respect to the cathode of the gun G, and hence the

equilibrium potential of the target. Thus, a portion of the photocathode Pwhich is brightly illuminated will cause a large number of electrons to be

emitted which will bombard the target Tproducing large numbers of secondary

electrons, causing the corresponding portion of the target to be charged in a

positive direction (due to the loss of electrons). Now, the glass used for the

target is very thin and has a rather critical resistivity. A charge on the front

of the target travels through the glass and hence the back of the target

becomes positively charged. Thus, there is a positive charge distribution on

the right-hand side of the target, corresponding to the picture on the photo-

cathode.

When there is no light on the photocathode the target T is charged to

cathode potential by the action of the beam and so no electrons fall on it.

Instead, all the electrons travel back along the same path but are persuaded

(by an electrode near to the gun called a persuader) to enter the electron

multiplier which is arranged around the electron gun. Hence, a compara-

tively large current is obtained from the multiplier. When a portion of the

target becomes charged positively (as explained above) due to the action of

the light, some of the electrons in the beam are used to recharge the target

back to cathode potential and hence fewer electrons return to the electron

multiplier. Thus, the return beam, and hence the output current, varies

according to the charge on the target which is controlled by the light on the

photocathode.

The advantages of this tube are:

(0 Very high sensitivity.

(it) Stable when correctly operated.

(iit) Definite black level.

(iv) Convenient shape and causes no difficulty with the optical system.

(v) No spurious signals.

A disadvantage is that the target should be maintained at a constant

temperature so that its resistivity is of the correct value. The tube should

therefore be switched on some time before it is required for use or a heater

may be fitted round the tube.

This type of camera tube is used throughout the world, both in studios and

for outside broadcasts where its high sensitivity is a great advantage. This

sensitivity is of the same order as the human eye, and it will give an output

under lighting conditions that would be impossible with other types of tube.

If excessive light falls on the photocathode the number of electrons falling

on the target is so great that the potential of this portion of the target reaches

approximately that of the mesh (about + IV); when this occurs the mesh no

longer collects all the secondary electrons. Instead, some of the secondary

electrons fall back on to the surrounding area of the target. They tend to

charge this area in a negative direction corresponding to little light. Theeffect is a dark halo around brightly-lit objects, which may sometimes be seen

when this type of camera tube is being used. Normally, it is not serious and


has the effect of increasing the contrast of the picture and giving the appearance

of sharp definition and clearness.

VIDICON AND STATICON CAMERA TUBESThese types of tube operate on a different principle from those described.

Instead of using a photo-emissive cathode, a photo-conductive material is used

which changes its resistance with the amount of light falling on it. The tube is

much smaller, being about 1* in diameter and about 6* long with an electron

gun G at one end, as shown in figure 4.10. The beam is focused by the

ELECTRON BEAM

ANODE+ 300V

CATHODE(earth pot«ntial)

•30V

FIG. 4.10. PHOTO-CONDUCTIVE TYPE OF CAMERA TUBE

solenoid S and deflected by the coils Cx and C2 as in the C.P.S. Emitron and

image orthicon. On the face of the tube is a transparent conducting coating

which forms the signal plate, behind which is a very thin layer of photo-

conducting material such as Antimony Trisulphide. To prevent ion bombard-

ment a fine mesh M is placed about 2-3mm from the target.

With no light, the side of the target scanned by the beam is stabilized at

cathode potential. The signal plate is biased at about +30V, hence an

electric field is produced across opposite faces of the target. The magnitude

of the current which flows through each section of the target material, from

signal plate to scanned side, will depend upon the light on that particular

portion of the target. The current which flows charges positively the side of

the target which is scanned and, when the scanning beam passes over this

section, it is recharged to cathode potential by the beam. Hence, a current

flows in the signal plate which depends on the change of potential which

occurs when each section of the target is charged back to cathode potential,

this being the capacitance current between the two sides of the target over

the portion being scanned (similarly to the signal current produced in an

iconoscope).

The main disadvantage of this type of tube is its poor sensitivity and Us

long time-lag. The change of resistance with light is not instantaneous and

this results in smearing of moving objects, which is particularly bad if the

light is poor. On the other hand, it is simple to operate and it is small.

For these reasons it is mainly used for closed circuit television where cost

and small size are important. Such systems are used extensively in industry,

in education and for many other purposes. Many closed circuit television

systems are available and they all use this type of tube. The resultant picture

quality can be excellent but the quality of the picture depends very much on

the cost of the equipment. This type of camera has been used in certain


studios where good lighting can be provided and where little rapid movement

takes place, such as news studios.

PLUMBICON CAMERA TUBERecently a new photo-conductive camera tube known as a Plumbicon tube

has been developed, which has a number of advantages over the Vidicon tube.

In basic principle the tube operates like the Vidicon and is similarly constructed,

but the photo-conductive material is lead monoxide. The target is "doped" so

that it forms an n-i-p sandwich (i is an intrinsic semiconductor without added

impurity) and this acts like a reverse biased diode. The tube has a higher

sensitivity than the Vidicon, has a faster response and extremely small dark

current. It is being used mainly in colour television cameras.

STUDIO EQUIPMENTHaving described the camera tubes we will now consider briefly the

equipment connected with the camera tube, usually called the camera chain.

This is complicated and only an elementary idea of the equipment and circuits

can be considered.

The first piece of equipment essential to any camera chain (or for any

TV system) is a pulse or waveform generator. This is rather like a pattern

generator, of the higher class type, but rather more complex and contains

many valves or transistors. The purpose of the waveform generator is to

produce the pulses necessary to operate the camera tube and associated equip-

ment. It must provide a composite synchronizing pulse signal, consisting of

both line and field pulses correctly positioned relative to each other, and with

the pulses of correct width. This signal is used for synchronizing the timebases

of the camera and monitoring equipment and is also used to add to the vision

signal from the camera, for transmission. Also, the waveform generator must

provide what are termed blanking pulses. These are required at both line and

field frequencies and may be used to suppress the beam of the camera tube

during both line and field flyback periods, or suppress the picture signal

during these periods.

The generator consists of a master oscillator operating at twice line

frequency (i.e. 20,250Hz for 405 lines) which is divided down by frequency

divider stages, or counter stages, by a factor of 405 (say, by dividing by 9 twice

and then by 5) which results in 50Hz—the field frequency. By dividing by a

factor of 2 from this master oscillator frequency the line frequency is obtained.

An oscillator at twice line frequency is used for the following reasons:

(/) Pulses of twice line frequency are required during the field pulse period

(i.e. the eight field pulses are at twice line frequency) and these are derived

from this oscillator.

(if) We can only divide by a whole number and not by 202J (which is the

number of lines in a field) in order to obtain the field frequency.

Various circuits may be used for this counting down process. At one

time the television signal was locked to the nominal 50Hz mains on 405 lines

and some means had to be used to control the master oscillator (at 20,250Hz)

so that when it was divided by 405 the frequency obtained was exactly that of the

mains. This arrangement has some similarity to flywheel synchronizing. The

50Hz signal obtained from the divider circuits was compared in phase with the

mains in a phase comparator. A signal was obtained from this comparator and

fed to the master oscillator in order to control its frequency.* Many valves or

transistors are, of course, necessary to shape the pulses to the correct shape

and precise width. Arrangements are sometimes made so that a simple

pattern, or a number of patterns, such as a cross is available from the

waveform generator for testing purposes.

* On 625 lines the field frequency is not locked to the mains. Since 405-line pictures are con-

verted from 625-line pictures these also are not locked to the mains frequency.


2.

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The exact arrangement of the other equipment will depend on the type of

camera tube used and on the manufacturer. The following description is a

typical modern camera tube chain. The camera itself contains the following

(see figure 4.11 opposite):

(0 Line timebase scan circuits.

(if) Field timebase scan circuits.

(Hi) E.H.T. generator.

(iv) Protection circuits to cut off the beam of the camera tube if either of

the scans fails.

(v) Blanking amplifier to cut off the beam of the camera tube during both

line and field flyback times or suppress the picture signal.

(v0 Video preamplifier to increase the magnitude of the signal from the

signal plate of the camera tube to a sufficiently large value, say 0-3V peak to

peak, so that it can be fed through a long camera cable without the intro-

duction of noise and interference.

(vii) View-finder, comprising:

(a) A small magnetically focused and deflected cathode ray tube.

(b) Line timebase and scan circuits.

(c) Field timebase and scan circuits.

(d) E.H.T. supply for cathode ray tube.

(e) Video amplifier.

(f) Synchronizing separator.

(viif) Various facilities for talk back, i.e. 'phones and microphone con-

nected to the producer's equipment.

(ix) Indication facilities, i.e. cue lights and "camera on" lights.

(x) Lens turret with means for changing lenses rapidly or zoom lens and

means for adjusting focus and iris setting.

The camera is fed to the remainder of the equipment through a cable

which in outside broadcasts may be 1000ft. long. As the time of travel along

the cable is appreciable in terms of a line pulse-time, corrections have to be

made so that the pulses arrive at the camera at the correct instant relative to

the pulses fed to other parts of the equipment. There is a loss at high

frequencies in the cable, so correction for this is necessary and is madevariable to enable different lengths of cable to be used.

Each camera feeds to a camera control unit and a camera monitor as

shown in figure 4.11. The multicore cable from the camera first enters the

camera control unit where the video signal is amplified in an amplifier. This

amplifier corrects for loss of high frequencies in the camera tube and circuits

and, as explained, for the loss in the camera cable. Means of adjusting the

gain are provided in this amplifier so that the correct amplitude of signal maybe fed out of the control unit. The vision signal is now fed into a clamp

circuit. This is a circuit which operates at line frequency so as to clamp the

waveform during the line flyback period to a fixed reference voltage (but not

in the same way as a d.c. restorer used in television receivers). By using this

circuit the effect of low frequency interference, such as hum and microphony,

can be largely removed. Next, circuits are used to restore the black level and

clip any spurious signals which may occur below black level. These circuits

are too complicated to describe in this book. Blanking signals are now used

to eliminate any signal during the blank period, during the line and field

flyback periods. This is done in the blanking amplifier. Next, the composite

synchronizing pulses are added to the vision waveform in the correct ratio.

The complete signal is now fed back to the view-finder so that the camera

operator can see the picture actually picked up by the camera tube. Thecomplete signal is also fed through cathode followers to the output and also

to the camera monitor.

A section of the control unit is devoted to the production of correct pulses


from the synchronizing pulses fed from the waveform generator, as manypulses are required of definite length and delayed by precise amounts relative

to each other.

Most of the electrical controls of the camera are on the control unit andoperated by the control engineer. The camera operator only controls the

type of lens (focal length), the focus and iris setting of the lens, together withthe local view-finder controls.

The signal now goes to the camera monitor unit which consists of a

monitor tube with associated timebases and scanning circuits, e.h.t. supply andvideo amplifiers. The picture is displayed on the tube so that the control

engineer can adjust the camera controls for best results. The camera monitorunit also contains an oscillograph tube so that the video waveform can beexamined, viewed at either line or field frequencies.

The above is the equipment of one camera chain. More than one camerawill normally be required and a set of the equipment (apart from the waveformgenerator which is common) is required for each camera. When a number of

cameras are in use a mixer unit is required so that the cameras may be selected

as required. Mixing is a more involved operation than at first might appear

:

arrangements are made so that each camera may be faded into circuit as

required or be switched instantly in and out. The mixer unit contains a

monitor tube for viewing the picture being transmitted. There is no great

difficulty in mixing cameras which are all operating from the same waveformgenerator {i.e. in synchronism). It must be remembered, however, that the

operation of the faders and switches must not alter the magnitude of the

synchronizing pulses, only the picture portion being faded or switched. This

means that only the vision signals must be fed through the fader controls andswitches, and the synchronizing pulses must be added after the faders andswitches.

The above is only a brief description of the complicated equipment used

in a studio or for an outside broadcast. In addition, film scanners, video tape

recorders, test equipment and various monitors for viewing the outgoing signal

are required.

INDEX

Automatic Contrast ControlAutomatic Frequency Control CircuitsAutomatic Gain control, 625-line

from synchronizing separatorgatedvision

with transistors

limiter

313224131711

275

Bend and Tilt SignalsBlanking pulsesBlack spotter

5161

2,7

Camerachain

Cathode stabilized camera tubeC.P.S. Emitron camera tubeClamp circuit

Comparator (phase)Contrast Control (automatic)

high level

34,

47,4861555763

36,383129

Delay Diode (A.G.C.)Diode Phase Detector

1436,39,42

Emitron Camera Tube 48

Fast Pull-down MechanismFlying spot scanningFlywheel synchronizing circuits

Forward bias control (a.g.c.)

48473228

Gated A.G.C. SystemGating pulses

1718

High Level Contrast Control 29

Iconoscope Camera TubeImage iconoscope camera tubeImage orthicon camera tubeInterference

effect on synchronizinglimiter

48515833331

Limiter, on a.g.c

Limiters, vision interference

Low velocity camera tube .

.

251

55

Manually Adjusted LimiterMean level a.g.c

MonoscopeMosaic

2125349

65

66 INDEX

Negative Feedback Type of Limiter 3,

6

Orthicon Camera Tube 55

Pentode Interference Limiter 7

Phase Detector 34,38using diode 36,39

discriminator... 34

comparator 34,36,38,61

Photicon camera tube 53

Photo-multiplier 58

Plumbicon camera tube 61

Pulse generator ,°1

Pull-in range (of flywheel synchronizing circuit) 47

Pull-in time (of flywheel synchronizing circuit) 47

Reactance Valve 35

Reverse bias control (a.g.c.) 27

Secondary Emission 54

Shading signals 51

Signal plate £9Staticon camera tube £«Studio equipment °{

Super-Emitron camera tube •«

Synchronizing pulse cancelled a.g.c 13,26

Television Cameras 47

Transistor a.g.c. circuits ^7Trapezium distortion correction -> l

Vidicon Camera Tube 60

View-finder°f

Vision a.g.c X

JVision interference limiter •• *

using negative feedback i,o

Waveform Generator 61

COLOUR TELEVISION WITH PARTICULAR REFERENCETO THE PAL SYSTEM

G. N. Patchett, B.Sc.(Eng.), Ph.D., C.Eng.2nd edition £2.50 ipaprrback)

ELECTRICAL ENGINEERING FOR ORDINARY NATIONALCERTIFICATE

G. N. Patchett, B.Sc.(Eng-), Ph.D.. C.Eng.Vol. I—Current Electricity 42Jp Vol. 3—Altermutnm Cmrrmt Theory 62)

p

Vol. 2—MapuHiim and Etearoitaiici J7Jp Vol. 5—Basic FJectrnmct 45u

RADIO SERVICINGVol. 1—Basic Eiactrotechnolory 42*p Vol. J—Final Radio Theory 75pVol. 1 -imermedtale Radio Theory 62) p Vol. 4—Fault Finding TOp

TELEVISION SERVICINGG. N. Patchett, B.Sc.(Eng.), Ph.D., C.Eng

Vol I TOp Vol. 2 7Jp Vol. 3 J5p Vol. 4 75p

THE OSCILLOSCOPE BOOK J5P

E. N. Bradley

ELECTRONIC NOVELTY DESI< 42»p

I. J. Kampel

HOW TO GET THE BEST OUT OF YOUR TAPE RECORDERPercival i. Guy

PRACTICAL TAPE RECORDINGPercival J. Guy

USING AN OSCILLOSCOPED. W. Eastedmg

RADIO SERVICING PROBLEMSW. A.L.Smith, M.I.Prodi

WELDING SCIENCE. -Vol. I 2ip

J. Gardner, F.R.S.A., M.Inst. Met., A.iivtt.W.

37JP

70,.

4Sp

Vol. 2 J71P

DUAL-STANDARD AND 625-LINE TELEVISION RECEIVERS 42j„

Gordon J. King, Assoc.I.E.R.E., Mis

FAULT LOCATION EXERCISES IN RADIO AND TELEVISIONSERVICING Vol. I Vol. 2 42*,. Vol. 3 »p

K. i. Bohlman, A.M.Inst.E.

THE DESIGNER'S GUIDE TO BRITISH TRANSISTORSI. I. Kampel il 25 {paperback)

PAL-D COLOUR RECEIVER: QUESTIONS AND ANSWERS 67| P

K. J. Bohlman, A.M.Inst.E.

MECHANICS' COURSE IN RADIO, 1 1 I EVISIONAND ELECTRONICS (433)

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Electronic Systems (Pari I: First Year) 7SP

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