IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998 69

Ez = KE

(z � h)x

r21

g1(r1)�(z + h)x

r22

g1(r2) Px

+y(z � h)

r21

g1(r1)�y(z + h)

r22

g1(r2) Py

+ �

x2 + y2

r21

g1(r1) + g2(r1)

�

x2 + y2

r22

g1(r2) + g2(r2) Pz

+y

r1g3(r1) +

y

r2g3(r2) k0Mx

�

x

r1g3(r1) +

x

r2g3(r2) k0My (14)

Px; Py; andPz are the electric dipole moments.Mx;My; andMz

are the magnetic dipole moments.The following definitions are used in the above equations:

f(r) = e�jk r

=r (15)

KE = �jk0�0

4�(16)

g1(r) =3

(k0r)2+ j

3

k0r� 1 f(r) (17)

g2(r) =2

(k0r)2+ j

2

k0rf(r) (18)

g3(r) =1

k0r+ j f(r): (19)

REFERENCES

[1] BS 6527: 1988 and EN 55022: 1987, British Standards Institution.[2] Bulletin OST 55,Characteristics of Open Field Test Sites, Fed. Commun.

Commission Office Sci. Technol., Washington, DC, Aug. 1982.[3] M. L. Crawford, “Generation of standard EM fields using TEM trans-

mission cells,”IEEE Trans. Electromagn. Compat., vol. EMC-16, pp.189–195, Nov. 1974.

[4] I. Sreenivasiah, D. C. Chang, and M. T. Ma, “Emission characteristics ofelectrically small radiating sources from tests inside a TEM cell,”IEEETrans. Electromagn. Compat., vol. EMC-23, pp. 113–121, Aug. 1981.

[5] P. Wilson, D. Hansen, and D. Koenigstein, “Simulation open area testsite emission measurements based on data obtained in a novel broadbandTEM cell,” in Proc. IEEE Int. Symp. Electromagn. Compat., Denver,CO, May 1989, pp. 171–177.

[6] P. Wilson, “On correlating TEM cell and OATS emission measure-ments,” in IEEE Trans. Electromagn. Compat., vol. 37, pp. 1–16, Feb.1995.

[7] L. Turnbull and A. C. Marvin, “Effect of cross-polar coupling on openarea test site measurement correlation and repeatability,”Proc. Inst.Elect. Eng. Sci. Meas. Technol., vol. 143, no. 4, pp. 202–207, July 1996.

Surface Transfer Impedance Measurement:A Comparison Between Current Probe andPull-On Braid Methods for Coaxial Cables

A. Morriello, T. M. Benson, A. P. Duffy, and C. F. Cheng

Abstract— Theoretical models to compute the surface transferimpedance of cables often rely on simplifying assumptions. This, togetherwith the fact that surface transfer impedance can vary considerablybetween cable samples of the same type, means that measurementsbecome necessary. In this way anaverage performance may bedetermined. Many transfer impedance measurement methods have beenproposed over the years and each has its own relative strengths. In thispaper, two frequency-domain measurement methods are compared: thecurrent probe method and the pull-on braid method. Both the methodsare inexpensive and can be set up very quickly without expensive cablepreparation. Moreover, they operate over a broad frequency range withhigh accuracy. This is shown by the good agreement obtained betweenmeasurements carried out with the two methods.

Index Terms—Coaxial cable, measurement method, surface transferimpedance.

I. INTRODUCTION

Both the surface transfer impedance(Zt) and the transfer capac-itive impedance(ZF ) are quantities for expressing numerically themechanism that allows the passage of electromagnetic energy acrossthe cable shield.

The surface transfer impedanceZt is the quotient of the serieselectromotive force (e.m.f.)V induced per unit length inside theshield to the disturbing currentI flowing along the outer surfaceof the shield in an electrically short piece of cable (see Fig. 1)

Zt =V

Il: (1)

The transfer capacitive impedanceZF is the quotient of the currentinduced per unit length flowing inside the shieldI to the outsidedisturbing voltageV multiplied by Zc Zc in the same electricallyshort piece of cable (see Fig. 2)

ZF =I

V lZc Zc = YtZc Zc (2)

whereZc is the characteristic impedance of the cable,Zc is thecharacteristic impedance of the outer disturbing circuit, andYt iscommonly indicated as the transfer admittance.

Usually, the shielding effectiveness of a cable depends on the valueof the transfer impedance alone since the capacitive impedance playsa minor role, provided that the cable is sufficiently remote from aground plane [7]. Notable exceptions are theoptimizedbraid shieldsin which Zt is minimized and may be comparable withZF .

Many methods for measuring transfer impedance have been pro-posed over the years [1]–[3] including the current probe [4], [5] andthe pull-on braid [6] methods compared in this paper. Several models

Manuscript received May 8, 1997; revised October 1, 1997.A. Morriello is with the Department of Electrical Engineering, University

of Rome “La Sapienza,” Rome, 00184 Italy.T. M. Benson and C. F. Cheng are with the Department of Electrical and

Electronic Engineering, University of Nottingham, Nottingham, NG7 2RDU.K.

A. P. Duffy is with the Department of Electronic and Electrical Engineering,De Montfort University, Leicester, LE1 9BH U.K.

Publisher Item Identifier S 0018-9375(98)01719-0.

0018–9375/98$10.00 1998 IEEE

70 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998

Fig. 1. Definition of the surface transfer impedanceZt.

Fig. 2. Definition of the transfer capacitive impedanceZF .

have also been developed in order to calculate this from cable designparameters [8], [9].

As far as braided cables are concerned the transfer impedance canbe regarded as the sum of a surface transfer resistance(Rd) and atransfer mutual inductance. The latter, in a braided shield, is due tocoupling through both holes(Mh) and braid wires(Mb). The transferimpedance can thus be determined as

Zt = Rd + j!(Mh �Mb): (3)

Accurate models to calculateRd have been developed [8] for manycable types. Predictions of the transfer mutual inductance is muchmore difficult, especially when inhomogeneous shields such as braidsare involved [3] due to the complexity of the shield.

Therefore, computation ofZt is often either impossible or unreli-able and it is usual for it to be determined by measurements.

The most obvious way to perform a surface transfer impedancemeasurement is to inject a known current on the shield under testand measure the voltage induced within the cable.

A triaxial test unit is commonly employed [10]. The device is madeof a cylinder, which encloses coaxially the cable under test (CUT),as shown in Fig. 3. The cable (which is usually referred to as thepickup line) is terminated in its nominal characteristic impedance atone end, while the other end forms the output to a detector. Thedriveline (made of the outer cylinder and the cable shield) is fed from oneend and the opposite end is short circuited.

If the cable under test is electrically short (e.g., less than onetenth of wavelength) relationships between measured data and transferimpedance can be derived with a lumped circuit analysis. At higherfrequencies, resonance effects arise and a correction factor must beintroduced. However, for even higher frequencies (such that the cablelength corresponds to say a quarter wavelength) the correction factormay introduce severe errors in the calculations, rendering the analysisunreliable.

Fig. 3. Scheme of a generic triaxial fixture.

Fig. 4. Current probe method setup.

In order to increase the useful frequency range, it is usual todecrease the cable length. The shortcoming is that the influence ofthe connectors then increases, adversely affecting the measurementaccuracy.

One way to overcome this limitation is to have the shield and theouter conductor as close as possible. Thus, the end effects, which areactually proportional to the distance between the shield and the outerconductor, are reduced and a shorter cable sample can be used withincreased confidence.

In order to realize such a triaxial fixture a “milked-on” braid (orpulled-on braid) can be tightly fit over the insulating jacket of thecable sample under test. Themilked-onbraid plays the same role asthe cylinder in the classical triaxial test unit. An example of this testunit is shown in [11], but a more interesting arrangement has beenproposed in [6].

Another method devised to overcome the effects of resonances usestwo current probes (current transformers), which are to be clippedaround the cable under test [4], [5]. This measurement configurationis illustrated in Fig. 4. A current transformer acts as a source injectingcurrent at one end of the shield, while the other monitors the currentthat flows along the shield at the opposite end. There is also the needof a return path that must run outside the transformers.

The circuit model is the same as that of the triaxial device, butin this case, the feed of the drive line is unknown. On the otherhand, the current flowing through the short circuit is measured bythe monitor transformer.

The cable under test is terminated in its characteristic impedanceat one end and connected to the detector at the other.

Two measurements must be carried out (at the CUT end and at themonitor transformer) and the transfer impedance is computed as theirratio. This ratio acts to cancel out the resonance effects so that theuseful frequency range can extend past the first resonant frequency.As a matter of fact, measurements up to the third or fourth resonantfrequency can be achieved without loss of accuracy.

IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998 71

The main reason for difficulties in measuringZt arises becausemost methods neglect the effects of the test jig or do not distin-guish them from measurements of the cables [12]. Comparison ofmeasurements obtained with different methods can be very usefulfor detecting test-jig effects. The main problem is that, ideally,measurements must be performed on the same cable sample or atleast on cable samples from the same reel. In fact,Zt of cables ofthe same type (but supplied by different manufacturers) may varyconsiderably [3].

Substantial changes inZt can take place even in the same sampledue to aging and handling effects. Some cables exhibit a notablechange in shielding when the cable is flexed: both methods describedallow testing for this effect, although this feature has not beeninvestigated in the present work.

Finally, for a consistent comparison between measurement meth-ods, the accuracy of these methods must be carefully evaluated andcompared.

In the present paper,Zt of a number of coaxial cables hasbeen measured with the pull-on braid method and the current probemethod. Experimental results showed good agreement between themethods, thus indicating the reliability of both. Moreover, the pull-on method has been successfully used in assessing the shieldpolarity.This is conventionally definedpositive if Mh > Mb andnegativeifMh < Mb.

In order to assess the shield effectiveness, only the magnitude ofthe transfer mutual inductance is needed. On the other hand, often themeasurement is made to obtain a model of the cable, which is to beused in the full system simulation, for electromagnetic compatibility(EMC) performance evaluation. In this second case, the cable polarityis needed for more accurate simulations.

II. THEORY

With a generic triaxial device (Fig. 3), a few calculations [13]are needed to relate the source signal and the detected signal to thetransfer impedance as

Vl1 = �A(ZtYc + Zc Yt)e�l � 1

�V�

02

�B(ZtYc + Zc Yt)e��l � 1

�V+02

+ A(ZtYc � Zc Yt)e�l � 1

�V+02

+B(ZtYc � Zc Yt)e��l � 1

�V�

02 : (4)

Here

A =1

2

(1 + �l1)e�k l

1� �01�l1e�2k l(5)

B = ��01A (6)

V+02 =

Vg

2

1� �g

1� �g�l2e�2k l(7)

V�

02 = V+02�l2e

�2k l (8)

� = k1 + k2 (9)

� = k1 � k2 (10)

where �l1 is the reflection coefficient of the pickup line at thedetector,�01 is the reflection coefficient of the pickup line at thetermination,�l2 is the reflection coefficient of the drive line at thetermination,�g is the reflection coefficient of the drive line at thesource,k1 = �1 + j�1 is the propagation coefficient of the pickupline, k2 = �2 + j�2 is the propagation coefficient of the driveline, Zc1 is the characteristic impedance of the pickup line,Yc2is the characteristic admittance of the drive line,Zt is the transfer

impedance,Yt is the transfer admittance,Vl1 is the detected voltageat the cable end,Vg is the open circuit generator voltage, andl isthe cable sample length.

If the sample length iselectrically shortor, more precisely, if

f<

�40

l(11)

wheref is the operating frequency in megahertz andl is the cablesample length inm, the following approximations hold [6]:

elx � 1

x� l (12)

elx� 1: (13)

When using the pull-on method, cable samples can be made soshort that (11) applies and approximations (12) and (13) can beused. Since both themilked-onbraid and the CUT are short-circuited�01 = �l2 = �1 and we can rewrite (4) as

Vl1 =lVg

ZgZt: (14)

We note thatYt has been cancelled out and does not interfere inZt measurement any more.

As far as the current probe method is concerned since both endsof the CUT can be matched to the cable characteristic impedance�l1 = �g = 0 and the shield is short-circuited to the current returnpath so�l2 = �1. Furthermore, if we can neglect the transferadmittance, as is the case if the cable is sufficiently remote froma ground plane, we may assume

Zt �> Zc1Zc2Yt: (15)

Then (4) becomes

Vl1 =Il2Zt

4CF (16)

where the correction factorCF is given by

CF =1� e�l

�+1� e�l

�(17)

and Il2, the current flowing in the short circuit at the far end, ismeasured by the monitor current probe. The latter feeds a voltageVm to the detector given by

Vm = ZttIl2 (18)

where Ztt is the transfer function of the probe and is suppliedby the manufacturer. If (15) does not apply, then theoreticallyZtmeasurements are not possible. Nevertheless, where approximations(12) and (13) can be introduced [that is, under restriction (11)],Yt iscancelled out and does not interfere with theZt measurement.

III. EXPERIMENTAL SETUP

For both pull-on braid and current probe methods, a vector net-work analyzer (HP8752A) was employed to measure the scatteringparameterS21. Substituting the source and detector with the networkanalyzer had the advantage of reducing the overall measurement time.The network analyzer was linked to a PC through an IEEE 488General Purpose Interface Bus (GPIB) connection, which allowedautomatic collecting and storing of data on a disk. Furthermore,calibration is simpler than with a discrete source and detector.

As far as the pull-on braid method is concerned, a detailed expla-nation of the cable sample setup is provided in Fig. 5. Experimentshave been carried out with 30-cm cable samples in order to measureZt up to 100 MHz. Both ends of the sample were connected to thenetwork analyzer andS21 measurements were carried out. Taking into

72 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998

(a)

(b)

(c)

(d)

Fig. 5. Sample preparation for pull-on braid method. (a) The pull-on braidis first fitted over the cable insulating jacket. (b) The cable shield is shortcircuited to the pull-on braid at one end and to (c) the central conductor at theother. (d) Finally, a standard plug (N type or BNC, depending on the cablesize) is attached at each end.

account relationships between scattering parameters and voltages [6],we can rewrite (14) as

Zt =25

lS21 (19)

becauseZg = 50 and

S21 =2Vl1

Vg

by definition.From (19) it is evident that using a vector network analyzer, both

magnitude and phase ofZt can be measured. This is importantbecause by means of phase measurements it is possible to assessthe polarity of the shield. For positive braids the argument ofZt

tends to+�=2 as frequency tends to1, whereas for negative braids,the argument ofZt tends to��=2 as frequency tends to1. Such anassessment usually requires specific and rather complicated fixtures[2], [14].

The setup configuration of the current probe method is shown inFig. 4. The current return path is made of a brass board, whereasthe terminating plate at thefar end short circuits the braid to thecurrent return path. To enhance sensitivity a RF power amplifier was

Fig. 6. Current probes:S(m)21 measurement scheme.

Fig. 7. Current probes:S(l1)21 measurement scheme.

employed to feed the source transformer. For the same purpose cablesamples of 1 m were employed, although shorter samples would haveallowed measurements at higher frequencies.

The two measurements required are schematically shown in Figs. 6and 7. All the cables and devices had 50- characteristic impedanceso the relationships between theS21 parameter and voltagesVm andVl1 were obtained straightforwardly as

S(l1)21 =

Vl1

AcVREF(20)

S(m)21 =

Vm

AcVREF(21)

whereVREF is the output voltage level of the network analyzer andAc is the cable attenuation.

The output level(VREF) was the same in the two measurementsand the same cable was used to connect the transmission port (wheredetector is located) to the test fixture. This means thatAc andVREFare the same in (20) and in (21), therefore

Vl1

Vm=

S(l1)21

S(m)21

: (22)

Then (16) and (18) yield

Zt =4ZttS

(l1)21

CFS(m)21

: (23)

Using this method, assessment of the phase ofZt and, therefore,the braid polarity is feasible, provided that both magnitude and phaseof the transfer functionZtt are supplied. Unfortunately, in the presentcase, no information on the phase ofZtt of the transformers used wasavailable and the phase ofZt cannot be assessed.

As far as the correction factorCF is concerned, it is clear thatthe propagation constant of thedrive line (�2) is the same as invacuum(�0). Since the dielectric fill of all the tested cables had a

IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998 73

Fig. 8. Pull-on method accuracy versusZt.

relative permittivity of approximately 2.25, the propagation constantinside cables(�1) was equal to

p2:25�0 � 1:5�0. The attenuation

constants�1 and�2, which are also needed to work outCF from(17), are small and can be safely neglected.

IV. M EASUREMENT ACCURACY

Assessing the measurement accuracy is a rather difficult taskbecause there is no single accurate model against which to validate theexperiments. Notwithstanding this, as far as the pull-on braid methodis concerned,theoreticalaccuracy evaluation is straightforward.

From (19), the following expression for the maximum relativeinaccuracy onZt can be calculated:

�Zt

Zt MAX

=�l

l+

�S21

S21: (24)

Tolerance on sample length can be safely assumed not greater than7% (2 cm over a 30-cm sample). The network analyzer specificationsas supplied by the manufacturer express the inaccuracy�S21 asa function of the measured value ofS21. By means of (19), theinaccuracy can be expressed as a function ofZt. Thus, from (24),the relative maximum inaccuracy�Zt=Zt as a function ofZt itselfcan be worked out, yielding the plot of Fig. 8. This shows a lowaccuracy at lowZt, but tends to an error below 8% at higherZt,which does not vary with frequency.

The theoretical accuracy of the current probe method is moredifficult to evaluate. To simplify the matter we consider in (23)CF � 2l as far as error propagation is concerned

�Zt

Zt MAX

=�l

l+

�S(m)21

S(m)21

+�S

(l1)21

S(l1)21

: (25)

If all cables have the same length (as in the present experiments),S(m)21 is, for accuracy assessment purposes only, virtually independent

on the cable sample and a mean reference curve (i.e., the averagemeasuredS(m)

21 ), which is a function of frequency, can be employedfor all cables of the same length.

Taking into account the network analyzer accuracy specifications,assessment of�S(m)

21 as a function ofS(m)21 and, consequently, of

frequency is straightforward. From (23), an approximation ofS(l1)21

as function ofZt and frequency can be derived using the meanreference curve forS(m)

21 . Since�S(l1)21 is a function ofS(l1)21 it canbe transformed in a function of frequency andZt, using the abovementioned approximation ofS(l1)21 .

Fig. 9. Current probe method accuracy versusZt at several frequencies.

Finally, the relative tolerance on sample length can be assumedequal to 2% (the same 2 cm as in the previous case, but over a 1-msample).

By inserting the three contributions in (25), the relative maximuminaccuracy�Zt=Zt is calculated as a function ofZt and frequency,as shown in Fig. 9. At higherZt, the measurement error does notexceed 4% and is virtually frequency independent, while it increasesconsiderably for lowerZt—especially at higher frequencies.

The lowest measurable transfer impedance is determined in bothmethods by the detector noise floor. For the current probe method itis a function of frequency, whereas for the pull-on braid method isfrequency independent.

For the pull-on braid method the lowest measurableZt can befound by using the lowest measurableS21 in (19), as specified bythe network analyzer’s manufacturer.

As far as the current probe method is concerned, the lowestmeasurableZt is determined by the lowest measurableS(l1)21 and canbe easily worked out from (23) by substituting the lowest measurableS21 for S(l1)21 and using the mean reference curve mentioned earlierin place ofS(m)

21 .In Fig. 10 the lowest measurableZt with both methods is plotted

against frequency. The pull-on method allows measurement ofZtdown to 0.08-m/m. The other method has a better performance atlower frequencies (0.02-m/m at 1 MHz) but it falls dramaticallyas frequency increases (about 1-m/m at 100 MHz).

V. RESULTS

The magnitude ofZt as a function of frequency for a range ofcommon 50- coaxial cables with braided shields has been measuredwith both methods and results are compared in Figs. 11–16.

In each case, a comparison was also made between practical data(dashed and dotted lines) and predicted values of surface transferimpedance (solid lines).

Braid transfer inductance was calculated according to Tyni’s model[9] modified by Cuddet al. [15] where, with the notation of (3)

Mh =2C�0

� cos�

w

�Dm

2

e� �2 (26)

and

Mb =h�0

4�Dm

(1� tan2�): (27)

74 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998

Fig. 10. Minimum measurableZt versus frequency.

Fig. 11. Measured transfer impedance: cable RG 174 U/C MIL.

Here

h = radial spindle separation=2d2

w + d(28)

w = hole width=2�Dm

Ccos��Nd (29)

Dm = mean braid diameter= D0 + 2:5d, D0 is the braid diameterover dielectric,C is the number of spindles,d is the diameter ofbraid wires, and� is the weaving angle. Transfer resistanceRd in(3) has been modeled according to [8] as

Rd = Rdc

d

�(1 + j)

1

sinh d

�(1 + j)

(30)

where

Rdc = DC braid resistance

=4

�NCd2� cos�

� =1p

�f��0(31)

andN is the number of wires per spindle.

Fig. 12. Measured transfer impedance: cable URM43.

Fig. 13. Measured transfer impedance: cable RG 58 standard.

It is worth noting thatRd is a complex function of frequencybecause skin effect has been taken into account. As frequencyincreases, the skin depth decreases and magnitude ofRd decreasesand its phase goes negative. Up to a certain frequency, theRd termpredominates and magnitude ofZt decreases with frequency. Abovethat frequency, the mutual inductance predominates andZt start risinglinearly with frequency. If the mutual inductance is sufficiently low,a sharp dip may be found around the transition frequency.

For a positive shield, theZt phase shows a sudden shift fromdecreasing negative value to the asymptotic value of+�=2. On theother hand,negativeshields have aZt phase, which smoothly reachesthe asymptotic value of��=2.

For the double-braid coaxial cable (RG 223 U/C), the total transferimpedance was worked out from the transfer impedance of singlebraids, as shown in [8].

Due to manufacturing tolerances, the prediction of transferimpedance as a single value is meaningless. Thus, by measuring therange of variability of each mechanical parameter (strand diameter,weaving angle, and diameter over the dielectric) a range of possiblevalues for the transfer impedance can be quoted. The maximum and

IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998 75

Fig. 14. Measured transfer impedance: cable RG 58 U/C MIL.

Fig. 15. Measured transfer impedance: cable RG 213 U/C MIL.

minimum predicted values forZt calculated in this way are shown,along with the measured data, in Figs. 11–16.

The current probe method allowed measurements down to 1 MHz,limited only by the frequency response of the transformers. The upperfrequency limit was determined by the inaccuracy introduced bythe correction factorCF , which allowed measurements up to 600MHz with a 1-m cable sample length. WhenYt is not negligible, theupper frequency limit is lower. The RG 213 U/C cable, which is anoptimizedbraided cable, shows aYt so high thatZt measurements(Fig. 15) were practicable only up to 70–80 MHz.

The pull-on braid method has a lower frequency limit determinedby the network analyzer operating range and which is 30 kHz foran HP8752A. The upper limit is determined by (11). With 30-cmsamples, measurements up to 100 MHz turned out to be repeatableand accurate.

The resulting frequency overlap of the two methods spanned from1 to 100 MHz, which was broad enough to fully compare the twomethods. For all the cables, measurements with the two methodsare very close each other. A little discrepancy was found betweenthe measurements for standard RG 58 (Fig. 13) and RG 213 U/C(Fig. 15) at higher frequency. This discrepancy may be reasonably

Fig. 16. Measured transfer impedance: cable RG 223 U/C MIL.

TABLE IMEASURED AND PREDICTED [9] BRAID POLARITIES

associated with a high value ofYt, which lowered the frequencylimit of the current probe method (see last paragraph of Section II).

Except for the two cables mentioned above, measurements fallwithin the predicted tolerance range (although they cluster around theminimum), which suggests that the theoretical model systematicallyoverestimatesZt, as noted by other authors [3]. For standard RG 58and RG 213 U/C cables, measurements turn out to be just below theminimum predicted values, which further confirms the overestimationhypothesis.

As far as polarity assessment is concerned, in Figs. 17 and 18,measuredZt phase of a positive and a negative braid is shown, forillustrative purposes. In Table I the polarities of the measured braidedcables are compared with predictions [9] and show perfect agreement.

VI. CONCLUSION

Two methods for measuring cable surface transfer impedancehave been analyzed and compared. Both are inexpensive and enablerapid measurement, provided that a network analyzer linked to aPC is used. Either method is able to overcome limitations due toresonances. The current probe method relies on current transformersand resonance effects tend to cancel out when computingZt frommeasured data using ratios. The pull-on braid method employs veryshort cable samples to avoid resonances. Such short samples can beused provided that care is taken with sample preparation.

The upper frequency limit of the current probe method is, generallyspeaking, higher than the pull-on braid method, but this depends onfactors inherent in the measuring system (current probe operational-frequency range, presence of transfer admittance).

Measurements between approximately 1–600 MHz, with the cur-rent probe method and between 30 kHz and 100 MHz with the pull-onbraid method, have been carried out. In the overlapping frequencyrange (1–100 MHz) results show excellent agreement. For a complete

76 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 40, NO. 1, FEBRUARY 1998

Fig. 17. Measured phase ofZt: cable URM 43.

Fig. 18. Measured phase ofZt: cable RG 213 U/C MIL.

comparison, an indicative evaluation of measurement accuracy hasbeen carried out.

Finally, it is noted that the pull-on braid method enables the shieldpolarity to be found. This is very useful when the surface transferimpedance measurements are aimed at getting an accurate model ofthe cable, which is needed for EMC analyses of the whole system.

REFERENCES

[1] K. A. Simons, “A review of measuring techniques for determining theshielding efficiency of coaxial cables,” Jerrold Electron. Corp., May1973.

[2] E. P. Fowler, “Test methods for cable effectiveness—A review,” inProc.Inst. Elect. Eng. Int. Conf. Electromagn. Compat., York, U.K., 1988, pp.259–267.

[3] F. A. Benson, P. A. Cudd, and J. M. Tealby, “Leakage from coaxialcables,” Proc. Inst. Elect. Eng., vol. 139, pt. A, no. 6, pp. 285–303,Nov. 1992.

[4] M. J. Tealby, “Coupling between coaxial cables,” Ph.D. dissertation,Dept. Electron. Elect. Eng., Univ. Sheffield, U.K., Sept. 1987.

[5] T. M. Benson, C. F. Cheng, and P. A. Cudd, “Rapid method formeasurement of cable surface transfer impedance,”Microwave Opt.Technol. Lett., vol. 12, no. 4, pp. 187–189, July 1996.

[6] A. R. Martin and M. Mendenhall, “A fast, accurate and sensitive methodfor measuring surface transfer impedance,”IEEE Trans. Electromagn.Compat., vol. EMC-26, pp. 66–70, May 1984.

[7] E. P. Fowler, “Observations on the use ofZT (c) for comparing thebreakthrough capacitance of cable braids,” Int. Electrotech. CommissionWorking Group I, Document SC46A/WG1 (Fowler)3, Nov. 1973.

[8] E. F. Vance,Coupling to Shielded Cables. New York: Wiley, 1978.[9] M. Tyni, “The transfer impedance of coaxial cables with braided

conductors,” inProc. Electromagn. Compat. Symp., Wroc law, Poland,Sept. 1976, pp. 22–24.

[10] IEC-TC46(SC46A), “Radio-frequency cables 91-1—Part 1: Generalrequirements and measuring methods,” 4th ed., Int. Electrotech. Com-mission Publ., Geneva, Switzerland, 1986.

[11] E. P. Fowler, “Test rigs forZt measurements at high frequency,” IECWorking Group I, Doc. SC46A/WG (Fowler), Nov. 1973.

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