RADIATED EMISSIONS OF ELECTRICALLY LONG PCB BASED ON...

RADIATED EMISSIONS OF ELECTRICALLY

LONG PCB BASED ON IMBALANCE DIFFERENCE

AND DIPOLE ANTENNA MODELS

AHMED MOHAMMED YAHYA SAYEGH

UNIVERSITI TUN HUSSEIN ONN MALAYSIA

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RADIATED EMISSIONS OF ELECTRICALLY LONG PCB BASED ON

IMBALANCE DIFFERENCE AND DIPOLE ANTENNA MODELS

AHMED MOHAMMED YAHYA SAYEGH

A thesis submitted in

fulfillment of the requirement for the award of the

Doctor of Philosophy

Faculty of Electrical and Electronic Engineering

Universiti Tun Hussein Onn Malaysia

JANUARY 2017

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To the memory of my father, who would have been glad to see me at this moment

To my beloved mother for her constant, unconditional love during all my life.

To my wife and beloved daughter, Malak, for their love and support.

To my brothers and my sisters for their support and encouragement

To all my family members and friends for their love and support

To science,

enlightening us.

DEDICATION

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ACKNOWLEDGEMENT

Alhamdulillah, I am so grateful to Allah for giving me enough strength,

inspiration and guidance throughout my Ph.D. study. Many people have redounded

directly or indirectly to the completion of this thesis and their assistances are highly

appreciated.

First and foremost, I would like to express my deepest gratitude to my

supervisor, Professor Dr. Mohd Zarar Mohd Jenu, for his invaluable guidance and

assistance during my Ph.D. journey. Without his patience and motivation, this thesis

would not have been completed successfully. He gave me the opportunity to start

with him a new constructive experience of research work. I have learned from him

many aspects not only in the academic life but also in my living attitude. I am also

thankful to him for spending many hours for reading and commenting to review my

research publication including this thesis.

Thanks to Prof. Dr. Frank Leferink from university of Twente and Prof. Dr.

Todd Hubing from Clemson University for their valuable technical advices. I am also

grateful to all my colleagues from the EMC centre for their generous assistance in

my research-related problems, namely Mr. Encik Mohd Rostam Bin Anuar, Madam

Miskiah bt. Muhamad Ihsan and Mr. Sharifunazri Johadi.

I would like to acknowledge Universiti Tun Hussein Onn Malaysia (UTHM)

for giving me the opportunity to undertake my doctorate program by bestowing upon

me university grant scholarship.

Finally, I would like to extend my deepest gratitude to my mother for her

never ending love, my wife for her kind support and encouragement. I also dedicate

this Ph.D. thesis to my lovely daughter, Malak Ahmed, who always enjoys my time.

At last, I want to thank all my family members and friends who supported me during

my Ph.D. journey.

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ABSTRACT

Radiated emission (RE) compliance is one of the most important requirements for

legal and global marketing of electronic devices. Conventionally, REs are quantified

using full wave numerical simulations or measurement in semi-anechoic chambers

(SAC). However, these methods are expensive and time-consuming. Thus, they are

not suitable for early prediction of RE. This thesis proposes novel analytical methods

for estimating RE of printed circuit board (PCB) attached with cables at the design

stage for time and cost saving. First, a novel analytical method has been introduced

to estimate the RE from electrically long PCB traces based on transmission-line (TL)

theory, imbalance difference model (IDM), travelling wave antenna (TWA) and

dipole antenna model. However, this proposed method is limited to PCB-traces with

Quasi-TEM (QTEM) current distribution. Therefore, artificial neural network (ANN)

model has been constructed and trained using backpropagation algorithm for

estimating the RE from electrically long traces beyond Quasi-TEM operation mode.

Secondly, a novel analytical method was developed to estimate the common mode

RE from cables attached to a PCB-trace by adopting the IDM in conjunction with

asymmetrical dipole antenna model. The effectiveness of the proposed methods was

verified by comparing the predicted results to both 3D HFSS simulation and

measurement results. The results obtained using the proposed methods were in a

good agreements with both HFSS simulation results and measurement results with

accuracy of ±3dB which is acceptable range from design point of view. Finally,

graphical user interface (GUI) was developed which would be useful for early

prediction of RE in the electronics industry. These GUIs can be enhanced in future to

characterize automatically the RE from the entire PCB.

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ABSTRAK

Pancaran radiasi (RE) merupakan satu daripada keperluan penting dalam pemasaran

sah dan global peranti elektronik. Kebiasaannya, RE diukur menggunakan simulasi

berangka gelombang penuh atau pengukuran di dalam kebuk separa gema (SAC).

Walau bagaimanapun, kaedah-kaedah tersebut adalah mahal dan memakan masa.

Oleh itu, kaedah-kaedah tersebut tidak sesuai untuk perkiraan awal RE. Tesis ini

mencadangkan kaedah analisis untuk menganggarkan RE papan litar bercetak yang

dipasangkan kabel pada peringkat reka bentuk untuk menjimatkan masa dan kos.

Pertama, satu kaedah analisis baharu telah diperkenalkan untuk mengganggarkan RE

daripada laluan elektrik PCB yang panjang berdasarkan kepada teori talian

penghantaran (TL), model perbezaan ketakseimbangan (IDM), antena gelombang

kembara (TWA) dan model antena dwikutub. Walaubagaimanapun, kaedah yang

dicadangkan terhad kepada laluan elektrik PCB dengan pengagihan arus Quansi-

TEM (QTEM). Oleh yang demikian, model rangkaian neural buatan telah dibina dan

dilatih menggunakan algoritma rambatan belakang untuk menggangar RE daripada

laluan elektrik panjang melampaui mod operasi Quansi-TEM. Kedua, satu kaedah

analisis baharu telah dibangunkan untuk menganggarkan mod biasa RE daripada

kabel yang dipasang kepada laluan elektrik PCB dengan menggunakan IDM beserta

dengan model antenna dwikutub tak simetri. Keberkesanan kaedah yang

dicadangkan disahkan melalui perbandingan hasil anggaran dari kedua-dua hasil

ukuran dan simulasi 3D HFSS. Keputusan yang diperolehi menggunakan kaedah

yang dicadangkan adalah memuaskan dengan kedua-dua hasil ukuran dan simulasi

memperolehi keputusan ketepatan sekitar ±3dB iaitu julat yang diterima untuk

sesuatu rekabentuk. Akhirnya, antara muka pengguna bergrafik (GUI) dibangunkan,

yang bakal berguna untuk anggaran awal RE dalam industri elektronik. GUI ini

boleh ditambah baik pada masa hadapan untuk menentukan ciri RE secara automatik

daripada keseluruhan PCB.

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CONTENTS

DECLARATION ii

DEDICATION iii

ACKNOWLEDGEMENT iv

ABSTRACT v

ABSTRAK vi

CONTENTS vii

LIST OF PUBLICATIONS xii

LIST OF TABLES xiv

LIST OF FIGURES xv

LIST OF SYMBOLS AND ABBREVIATIONS xxii

LIST OF APPENDICES xxiv

CHAPTER 1 INTRODUCTION 1

Research background 1

Overview of EMC 2

1.2.1 EMC standards 3

Characterization methods of radiated emissions 5

Problem statement 6

Objectives of the research 9

Scope of the research 9

Significance of the research 10

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Organisation of the thesis 10

CHAPTER 2 LITERATURE REVIEW 12

Fundamentals of the radiating systems 12

2.1.1 Infinitesimal dipole 12

2.1.2 Short dipoles 16

2.1.3 Long dipoles 17

Measurement and modelling of EM emissions of PCB 17

2.2.1 Measurement of EM emissions of PCBs 18

2.2.2 Modelling methods for EM emissions of PCB 22

Sources of EM emissions on PCBs 26

2.3.1 Radiated emissions of PCB-traces 27

2.3.2 Radiated emissions of PCB-attached cables 32

Introduction to artificial neural networks 37

2.4.1 Description of multi-layer perceptron network for

EMC Applications 39

Summary of the literature review 40

Research gap 44

Chapter summary 47

CHAPTER 3 RESEARCH METHODOLOGY 48

Overview 48

Estimation of RE from electrically long PCB-traces 50

3.2.1 Analytical model for estimating DM RE of

electrically-long traces 50

3.2.2 Analytical model for estimating CM RE of

electrically long traces 52

3.2.3 Estimation of RE from PCB traces using ANN 53

Estimation of RE from cables attached to PCB-traces 55

Verification process of the proposed models 56

3.4.1 Verification process using HFSS 57

Chapter summary 62

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CHAPTER 4 ANALYTICAL METHODS FOR RADIATED EMISSIONS

OF PCB-TRACES 63

Introduction 63

Novel solutions for estimating RE of electrically long

PCB-traces 65

4.2.1 Analytical modelling for DM-RE of electrically

long PCB-traces 65

4.2.2 Analytical modelling for CM RE of electrically

long traces 78

4.2.3 ANN-MLP model for predicting RE from PCB-

traces 84

Chapter summary 98

CHAPTER 5 ANALYTICAL METHODS FOR RADIATED EMISSIONS

OF PCB-ATTACHED CABLES 99

Impact of attaching cables to PCB-ground plane on

CM RE 99

Analytical method for CM RE of cables attached to

PCB-ground plane 109

5.2.1 Identification and quantification of CM voltage

on board-cable structure 110

5.2.2 Estimation of CM RE from PCB with one

attached cable 114

5.2.3 Novel model for CM RE from two cables

attached to PCB 119

5.2.4 Flow chart for maximum CM RE of cables

attached to PCB 122

Analytical method for CM RE from actual cables

attached to signal trace and ground plane of PCB 123

5.3.1 Calculation of CM RE using IDM and asymmetrical

dipole antenna 123

5.3.2 Application of the proposed model on practical

cables 126

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Chapter summary 128

CHAPTER 6 RESULTS AND DISCUSSIONS 129

Validation results of the proposed models for RE from

PCB-traces 129

6.1.1 Results of the analytical modelling for RE based

on dipole antenna 131

6.1.2 Results of the analytical solution for DM RE using

TWA antenna 146

6.1.3 Results of the proposed ANN-MLP model for RE

of PCB-traces 155

Validation results of the proposed models for RE from

PCB-attached cables 160

6.2.1 Results of the proposed model for RE from cables

attached to PCB-ground plane 160

Results of the proposed model for CM RE from one cable

attached to PCB at load junction 161

Results of the proposed model for CM RE from two cables

attached to PCB at source/load junctions 169

6.4.1 Results of the proposed model for RE from real cables

attached to PCB- trace and PCB-ground plane 174

GUI of the proposed models 178

Summary 182

CHAPTER 7 CONCLUSIONS AND FUTURE WORKS 183

Conclusion 183

Recommendation for future works 184

REFERENCES 186

APPENDIX A- MEASUREMENT EQUIPMENT 200

APPENDIX B- HORN ANTENNA DATASHEET –

CALIBRATION CERTIFICATE 202

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APPENDIX C- DIPOLE ANTENNA DATASHEET –

CALIBRATION CERTIFICATE 203

APPENDIX D- UNCERTAINTY EVALUATION FOR

RADIATED EMISSION (EMC MEASUREMENT) 204

APPENDIX E- MATLAB M-FILE OF THE MAIN GUI OF

THE SOFTWARE TOOL 205

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LIST OF PUBLICATIONS

Journal Papers

[1] Ahmed Mohammed Sayegh, Mohd Zarar Mohd Jenu and Samsul Haimi Bin

Dahlan, “Analytical Solution for Maximum Differential-Mode Radiated

Emissions of Microstrip Trace”, IEEE Transaction on Electromagnetic

Compatibility, Vol. 58(5), OCT, 2016 (ISI indexed, Q1 with IF=1.3).

[2] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu, “Prediction of

common mode radiation from cables attached to PCB using imbalance

difference and asymmetrical dipole antenna models”, IET Electronic Letters,

Vol. 52, No 8, April, 2016 (ISI indexed, Q2 with IF=0.93).

[3] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu "Prediction of

radiated emissions from high-speed printed circuit board traces using dipole

antenna and imbalance difference model", IET Science, Measurement &

Technology Journal, Vol. 10, No 1, Jan.,2016 (ISI indexed, Q3 with

IF=0.954).

[4] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu "Estimation of

common mode radiated emissions from cables attached to high speed PCB

using imbalance difference model", ARPN Journal of Engineering and

Applied Sciences, Vol. 10, No 19, Oct., 2015 (Scopus indexed journal).

Conferences Papers

[1] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu, “Estimation of

Radiated Emissions from Microstrip PCB using Neural Network Model”,

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2016 IEEE Asia-Pacific Conference on Applied Electromagnetics (APACE),

Langkawi, Kedah, Malaysia 11 - 13 Dec, 2016 .

[2] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu, “Estimation of EM

Emissions From Coaxial and Twin-Lead Cables Attached to A PCB Using

Imbalance Difference and Asymmetrical Dipole Models”, 2016 IEEE Asia-

Pacific Conference on Applied Electromagnetics (APACE), Langkawi,

Kedah, Malaysia 11 - 13 Dec, 2016 (Invited Paper).

[3] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu "Estimation of

common mode radiated emissions from cables attached to high speed PCB

using imbalance difference model", International Conference on Electrical

and Electronic Engineering (IC3E), Melaka, Malaysia, August 2015.

[4] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu, “Closed-form

expressions for estimating maximum radiated emissions from the traces of a

Printed Circuit Board”, 2015 Asia-Pacific Symposium on Electromagnetic

Compatibility (APEMC), Taipei, Taiwan,26 – 29 May 2015.

[5] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu, “Prediction of

radiated emissions from high speed PCB traces using travelling wave antenna

model”, 2014 IEEE Asia-Pacific Conference on Applied Electromagnetics

(APACE), Johor, 8 - 10 Dec, 2014.

[6] Ahmed Mohammed Sayegh , Mohd Zarar Mohd Jenu and Syarfa Zahirah

Sapuan “Neural network based model for radiated emissions prediction from

high speed PCB traces”, International Conference on Computer,

Communications, and Control Technology (I4CT), Langkawi, 2 - 4 Sep,2014

[7] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu, “High Power

Electromagnetics (HPEM) as Re-emerging Sources of NIR, Non Ionization

Radiation conference, Kuala Lumpur, Malaysia. 29-30 October, 2013.

[8] Ahmed Mohammed Sayegh and Mohd Zarar Mohd Jenu “Evaluation of the

Radiated Emission of a Printed Circuit Board Attached with Cables”, 3rd

International Conference on Electric and Electronics (EEIC 2013), Hong

Kong, Atlantis Press, pp 195-198, Nov., 2013.

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LIST OF TABLES

2.1 Basic characteristics of the modelling methods of PCB-RE 26

2.2 Summary of the previous works for computing RE of PCB-traces 41

2.3 Summary of related works for computing RE from PCB-attached cables 43

2.4 Summary of literature review for RE from PCB using ANN 44

2.5 Research gap of this thesis for computing the RE from PCB-traces 45

2.6 Research gap for computing the RE from PCB-attached cables 46

2.7 Research gap for quantifying the RE from PCB using ANN 47

4.1 Samples of the PCB-parameters under study 86

4.2 Sample input data set for the proposed MLP model 94

4.3 RMSE of the proposed ANN with different hidden nodes 96

6.1 Accuracy and errors of the proposed model 143

6.2 Measurement and estimation of θmax in configuration A 153

6.3 Various configurations of PCBs to verify the proposed MLP using

HFSS 158

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LIST OF FIGURES

1.1 EMC aspects and their interrelationship 2

1.2 RE limits for Class A and Class B digital device in the FCC standards 4

1.3 RE limits for Class A ITE equipment in CISPR 22 4

1.4 Radiated emissions of PCB attached with cables 6

1.5 Cost of EMC implementation on the electronic product 7

1.6 Basic idea of the proposed models 8

2.1 A z-directed electric point dipole 13

2.2 Definition of near-field and far-field regions 14

2.3 Wave impedance of electric and magnetic sources 15

2.4 A z-directed short dipole 16

2.5 Measurement set-up of EMC-RE using TEM cell 19

2.6 Measurement set-up of RE using NF scanning method 19

2.7 Measurement of RE using radiation pattern method 20

2.8 Measurement of RE using reverberation chamber method 21

2.9 Electromagnetic field components of one cell using FDTD 23

2.10 Sources of EM emissions on PCB 27

2.11 DM-RE of PCB-traces 28

2.12 Maximum electric field due to DM currents on PCB-traces 29

2.13 Maximum electric field due to CM currents 31

2.14 Voltage-driven coupling from the PCB-trace to the attached cable 34

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2.15 (a) 3D structure describing current driven mechanism (b) cross section

of PCB using current driven mechanism 36

2.16 (a) Structure of PCB with two attached cables (b) CM equivalent

structure using imbalance difference concept 37

2.17 3-layer MLP for EMC applications 40

3.1 Research methodology of the proposed models 49

3.2 Flow chart of the proposed model for DM RE of PCB-traces 51

3.3 Flow chart of the proposed model for CM RE from PCB-traces 52

3.4 Estimation of RE from PCB-traces using MLP 53

3.5 Flowchart of the proposed ANN model for RE from PCB-traces 54

3.6 Flowchart for estimating CM RE from cables attached to PCB 55

3.7 Flow chart for verification process of the proposed models 56

3.8 Process of HFSS configuration 58

3.9 The main interface of HFSS 58

3.10 Configuration of solution type in HFSS 59

3.11 Boundary assignment in HFSS 59

3.12 Excitation port assignment in HFSS 60

3.13 Configuration of frequency sweep in HFSS 61

3.14 Reporting results in HFSS 61

3.15 Configuration of far-field radiation setup in HFSS 61

4.1 Two PCBs under test for study the impact of trace length on RE 64

4.2 Impact of short and long trace on RE of PCB 64

4.3 (a) Simple single-sided PCB for illustrating trace segmentation method

(b) equivalent circuit of the PCB under study 66

4.4 Segmentation of long trace into small segments for RE calculations 67

4.5 Flow chart of trace segmentation method 69

4.6 Calculation of the far-fields of two-parallel long traces 71

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4.7 Simple microstrip PCB for maximum DM-RE formulation (b) PCB-

trace above ground plane equivalent to TWA 74

4.8 Simple microstrip PCB structure (b) equivalent CM model based on

asymmetrical dipole antenna 80

4.9 Equivalent asymmetrical dipole model for CM RE 82

4.10 Equivalent symmetrical dipole antenna 83

4.11 Structure of microstrip PCB used for data collection 85

4.12 Impact of board length on the maximum RE 86

4.13 Impact of board width on the maximum RE 87

4.14 Effect of trace length on the maximum RE 88

4.15 Effect of trace width on the maximum RE 89

4.16 Impact of source impedance on the maximum RE 90

4.17 Impact of load impedance on the maximum RE 90

4.18 Impact of signal voltage on the maximum RE 91

4.19 Impact of dielectric permittivity on the maximum RE 92

4.20 Impact of dielectric thickness on the maximum RE 93

4.21 Topology of the proposed ANN model. 95

4.22 Training process of the MLP model with 15 hidden nodes 97

4.23 Training, validation and testing of the MLP model 97

5.1 PCB board under test in SAC 100

5.2 (a) Structure of PCB-circuit with one attached wire (b) equivalent

circuit of PCB under study attached with one wire 101

5.3 (a) Structure of PCB-circuit with two attached wires (b) equivalent

circuit of PCB under study attached with two wires 102

5.4 RE of PCB under test without attached wires 103

5.5 Total emissions of PCB with one attached wire 103

5.6 Total emissions of PCB with two attached wires 104

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5.7 Total RE of PCB loaded with 50 ohm 105

5.8 Microstrip PCB under study with 21cm x 10cm geometry 106

5.9 PCB on wooden table inside SAC 106

5.10 PCB with 0.5 m attached cable in SAC 107

5.11 PCB on table attached with two cables in SAC 107

5.12 RE of PCB with/without attached cables 108

5.13 RE of PCB with one cable of various lengths 109

5.14 Simple microstrip PCB attached with two cables 110

5.15 Exaggerated structure of PCB with cables attached to ground plane 111

5.16 The equivalent CM structure based on IDM 112

5.17 Equivalent TL model of microstrip PCB 113

5.18 PCB attached with one cable 115

5.19 CM equivalent structure of PCB with one cable using IDM 115

5.20 Equivalent CM model based asymmetrical dipole antenna 116

5.21 Equivalent asymmetrical dipole antenna 117

5.22 Microstrip PCB under study attached with two cables 120

5.23 Microstrip PCB under study attached with two cables (a) The

equivalent CM structure using IDM (b) Decomposition process

of CM structure using the proposed method 121

5.24 Flow chart of computing CM RE from one/two cables attached to PCB 122

5.25 PCB with two-wire attached cable 123

5.26 Equivalent CM model of PCB with two-wire attached cable 124

5.27 The equivalent CM structure based on IDM and asymmetrical dipole 126

5.28 Twin-lead cable with 300 ohm characteristic impedance 127

5.29 Coaxial cable with 75 ohm 128

6.1 3D radiation pattern for three configurations (open, short and load) 130

6.2 Schematic layout of PCB#1 under study 131

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6.3 Cross-sectional dimensions of PCB#1 under test 132

6.4 PCB#1 under study (a) top view (b) bottom view of DUT 133

6.5 (a) DUT placement in the shielding box (b) DUT within shielding

box in SAC (c) DUT Measurement set-up in SAC 134

6.6 (a) DM current versus CM current for short circuit and (b) DM current

versus CM current for open circuit (c) DM Current versus CM current

for 100-Ohm load 136

6.7 (a) DM radiations and (b) CM radiations both for short circuit

configuration, by proposed methods versus conventional method 138

6.8 (a) DM radiations and (b) CM radiations both for open circuit

configuration, by proposed methods versus conventional method 139

6.9 (a) DM radiations and (b) CM radiations both for 100-Ohm configuration,

by proposed methods versus conventional method 140

6.10 Predicted and measured total PCB radiated emissions for (a) short-

circuit (b) open circuit (c) 100 Ohm load 142

6.11 Simple microstrip PCB under test (PCB#2) 143

6.12 Estimated (a) DM & (b) CM RE of 150 mm long PCB trace with

75 ohm load 145

6.13 Prediction and measurement results of total RE from 150 mm long

PCB trace with 75 ohm load 145

6.14 Maximum DM- RE @ 3 meters of microstrip PCB with short-circuit

configuration 147

6.15 Maximum DM- RE @ 3 meters of microstrip PCB with open-circuit

configuration 148

6.16 Maximum DM- RE @ 3 meters of microstrip PCB with matched load

configuration 148

6.17 (a) Placement of PCB on the wooden table in SAC (b) Conducted

measurement in SAC 150

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6.18 Maximum RE @ 3 meters of microstrip PCB with (a) short-circuit

(b) open-circuit (c) 82 ohm load 152

6.19 (a) PCB under test installed on metallic box (b) DUT placement on

the wooden table in SAC 154

6.20 Measurement set-up of PCB shielded with metallic box in SAC 155

6.21 Maximum RE @ 3 meters of compact PCB shielded with box 155

6.22 RE of PCB#1 using HFSS simulator and MLP model 156



6.25 Envelope of the trapezoidal signal with 66 MHz, 6ns rise/fall time 159

6.26 RE of PCB under test using measurement in SAC 160

6.27 The estimated CM voltage at the (a) near-end to source (b) far-end near

to load 162

6.28 The estimated CM RE for PCB-cable structure using HFSS simulation

and analytical solution 163

6.29 Estimated and simulated results for RE in (a) 50 Ohm load configuration

(b) open-circuit configuration 164

6.30 RE of PCB under test (PCB#3) using HFSS and the proposed model 165

6.31 (a) Computation of the function )(f for the entire theta values

(b) determination of the angle, (θ), with maximum emission versus

frequency 166

6.32 PCB under test with one attached cable 166

6.33 (a) Measurement set-up of DUT in SAC (b) PCB inside metallic box

(c) placement of DUT above the wooden table 168

6.34 Estimated and measured RE of PCB under test 168

6.35 Simulated and estimated CM RE of PCB attached with two cables (a)

20cm x 4 cm PCB with two (0.5m, 0.5m) attached cables (b) 10cm x

4 cm PCB with two (0.5m, 0.5m) attached cables (c) 10cm x 4 cm

PCB with two (0.3m, 0.3m) attached cables 170

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6.36 Simulated and estimated CM RE of PCB attached with two cables (a)

10cm x 4 cm PCB with two (1m,1m) attached cables (b) 10cm x 4 cm

PCB with two (0.3m, 0.5m) attached cables 171

6.37 PCB under test with two attached cables 172

6.38 Estimation and measurement of CM RE of PCB attached with two cables

(a) Measurement setup of DUT in a SAC (b) Estimated and measured

results of CM RE of DUT 173

6.39 (a) PCB inside metallic box attached with twin-lead cable (b) Measure-

ment of CM of the twin-lead cable in SAC 174

6.40 CM voltage at the load junction between PCB and twin-lead cable 175

6.41 Estimated and measured CM RE of 1m twin-lead cable attached to PCB 176

6.42 Measurement set-up of CM RE from coaxial-cables attached to PCB 177

6.43 CM and DM voltages at the junction between the coaxial cable and PCB 177

6.44 Estimated and measured CM RE from coaxial cables attached to PCB 178

6.45 Main GUI of the developed software tool 179

6.46 GUI for choosing the source type to compute the RE from PCB-traces 179

6.47 Sub-GUI to input the PCB features 180

6.48 GUI for computing RE from one or two cables attached to PCB 181

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LIST OF SYMBOLS AND ABBREVIATIONS

E - Electric Field

H - Magnetic Field

dB - Decibels

V - Volt

θ - Antenna angle

f - Frequency

Wavelength - ߣ

Zin - Input Impedance

Z0 - Characteristic Impedance

I - Current

푍 - Load Impedance

휌 - Reflection Co-efficient

Ω - Ohm

β - Phase constant

푍 - Source Impedance

휂 - Intrinsic Impedance

휀 - Free Space Permittivity

휇 - Free Space Permeability

푘 - Wave Number

퐴 - Vector Potential

AF - Array Factor

ANN - Artificial Neural Network

CE - Conducted Emission

CISPR - Special International Committee on Radio

Perturbations Interference

CM - Common Mode

CS - Conducted Susceptibility

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CST - Computer Simulation Technology

DM - Differential Model

DRC - Design Rule Checker

EMC - Electromagnetic Compatibility

EMI - Electromagnetic interference

EN - European Norm

EUT - Equipment under Test

FCC - Federal Communications Commission

FEM - Finite Element Method

FF - Far-Field

GUI - Graphical User Interface

HFSS - High Frequency Structural Simulator

ICs - Integrated Circuits

IEC - International Electro technical Commission

IEEE - Institute of Electrical and Electronics Engineers

MAPE - Mean Absolute Percentage Error

MoM - Method of Moment

NF - Near Field

OATS - Open Area Test Sites

PCB - Printed Circuit Board

PEC - Perfect Electric Conductor

QTEM - Quasi-Transverse Electromagnetic

RE - Radiated Emission

RF - Radio Frequency

RS - Radiated Susceptibility

SAC - Semi Anechoic Chamber

SIRIM - Standards and Industrial Research Institute of

Malaysia

TEM - Transverse Electromagnetic

TL - Transmission Line

TRP - Total Radiated Power

TSM - Trace Segmentation Method

US - United States

UTHM - Universiti Tun Hussein Onn Malaysia

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LIST OF APPENDICES

APPENDIX TITLE PAGE

A Measurement Equipment 200

B Horn Antenna Datasheet Calibration Certificate 202

C Dipole Antenna Datasheet Calibration Certificate 203

D Uncertainty for Radiated Emission Measurement 204

E M-file of the Main GUI for the Software Tool 205

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1CHAPTER 1

INTRODUCTION

This chapter describes in details the research background, the focus of this research

study, the problem statement, research aim, research objectives, research scopes and

the significance of the study.

Research background

The worldwide proliferation of high-speed digital devices has imposed many

challenges to the circuit-designers of these modern devices. They must take into

account not only the product functionality, but also the electromagnetic compatibility

(EMC) regulatory requirements of their products. Thus, it is important to make those

electronic devices electromagnetically compatible with mandatory governmental

regulatory requirements before they can be sold legally and internationally [1], [2].

Radiated emission (RE) compliance is one of the EMC requirements to

ensure that the electronic product can function satisfactory without introducing any

electromagnetic interference (EMI) to the nearby electronic devices. In the past, the

electronic devices were operating in the low frequency range (less than 1 GHz)

resulting in lower emissions. Unfortunately, the modern devices are working in

higher frequencies (i.e. gigahertz range) where they become significant radiators of

electromagnetic (EM) energy. So the manufacturers of electronic devices must

control properly the RE of their products before they can sell them globally [3].

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Overview of EMC

According to the glossary of the International Electrotechnical Commission (IEC)

[4], EMC is defined as the ability of any electronic device or equipment to function

satisfactorily in its electromagnetic environment, and at the same time, not contribute

excessive electromagnetic disturbance to other devices/equipment/systems. The

electronic product may interfere with itself or with another product in that certain

environment as illustrated in Figure 1.1. In other words, a system or product is

classified as electromagnetically compatible if successfully fulfilled the following

requirements [5]-[7]:

i. it does not cause interference with other systems;

ii. it can tolerate the radiation or emission from other systems;

iii. it does not cause interference with itself.

Figure 1.1: EMC aspects and their interrelationship [6]

Electromagnetic Compatibility (EMC) (Ability to function satisfactorily)

Electromagnetic Susceptibility (EMS) (Suffering interference)

Electromagnetic Emission (EME) (Introducing disturbances)

Inside system

To other systems

By other systems

By system itself

Intersystem compatibility

Intrasystem compatibility

RE as the main scope of this research

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1.2.1 EMC standards

FCC [8] and CISPR [9] standards are the most widely adopted regulations for

commercial digital products around the world. FCC standard regulates the EM

emissions of digital devices marketed in US while the CISPR regulations focuses on

the EM emissions of digital devices sold in other countries of the world except U.S.

FCC standards classified the digital devices into two classes; Class A and

Class B [10]. The digital devices intended to be used in a business; industrial,

scientific and medical fields are classified to Class A, while the digital devices

applied in a residential environment belong to Class B.

Commonly, Class A limits are less strict than the Class B limits. This is due

to two reasons. The first reason is that the interference is more significant in the

residential environment since the electronic devices are working closer to each other,

so that it is not easy to minimize the interference effect. The second reason is to

consider the inability of the owners and users of Class B to protect their devices from

the electromagnetic interference. FCC standards specified the range for radiated

emissions from 30 MHz to 1 GHz and it can be extended further up to 40 GHz.

Figure 1.2 [6] shows the RE limits for Class A and Class B digital devices in the

FCC standards with 3 meters measurement distance.

In addition to FCC standards, CISPR is the European standard which had

been adopted in many countries around the world excepting U.S. One of most

important CISPR recommendations is CISPR 22 [9] which regulates the EM

emissions of information technology equipment (ITE). Similar to FCC, CISPR 22

also classified the digital devices to Class A and Class B. The RE is regulated in the

frequency range from 30 MHz to 1 GHz. Figure 1.3 [9] shows the radiated emission

limits for Class A and Class B ITE equipment in CISPR 22.

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Figure 1.2: RE limits for Class A and Class B digital device in the FCC standards [6]

In CISPR22, both Class A and Class B are always set at 10 meters distance

between the digital device under test and the receiving antenna. In contrast to that,

FCC regulations for Class A digital devices are set at 10 meters whereas Class B

regulations are set at 3 meters for the digital devices. Hence, the FCC regulations for

Class A for digital devices can be compared quite straight forward with CISPR 22

Class A whereas the CISPR 22 Class B for digital devices cannot be compared with

FCC Class B.

Figure 1.3: RE limits for Class A ITE equipment in CISPR 22 [9]

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In this case, FCC limits for Class B digital devices must be scaled at -10.45 dB to be

compared with the CISPR22 Class B limits. Therefore, the emissions at 3 meters are

assumed to be reduced by a factor of 3/10 if the measurement distance is moved to a

farther distance of 10 meters and vice versa. However, most countries adopt the

CISPR recommendations. This sets limits for the radiated and conducted emissions

of ITE equipment. Although there are a large number of EMC standards, the primary

one is the European norm EN 55022.

Characterization methods of radiated emissions

Generally, the radiated emissions of electronic devices can be quantified using

measurement or modelling methods. Open area test site (OATS) is one of the

measurement methods preferred by FCC. However, it requires making the test site

free of the unwanted noise signals which cannot be obtained easily considering the

widespread of the electronic devices everywhere. Alternatively, semi-anechoic

chamber (SAC) is employed as test site for measurement of RE. SAC provides an

all-weather measurement capability as well as security.

A SAC consists of a shielded room lined with radio-frequency absorber

material on the sides and at the top of the room to prevent reflections and simulate

free space. The floor of the room constitutes a ground plane without an absorber, and

this causes reflections that must be accounted for when performing simulations by

models as described in details in Chapter 2.

Although these measurement methods can evaluate the RE accurately, they

are not proper option to avoid the EMC issues earlier before fabrication of first

prototype. Alternatively, the RE can be estimated earlier at the design stage of the

product using one of the modelling methods [3], [6], [10] such as the numerical and

analytical methods.

Full-wave numerical solver is one of the modelling methods [11]–[15] used

to compute numerically the RE of the electronic devices. It employs one of the most

popular numerical techniques such as Finite Difference Time Domain method

(FDTD), Finite Element Method (FEM), Method of Moments (MoM) and the Partial

Element Equivalent Circuit (PEEC) method. These numerical techniques are widely

employed in a lot of the commercial software such as Ansys High Frequency

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Structural Simulator (HFSS) [16] which adopts the FEM method. Although those full

wave numerical solvers can compute the RE accurately, they require extensive

computational resources and time especially for today’s high speed devices.

Near Field (NF) to Far Field (FF) transformation is the second modelling

method for estimating the EMC-RE [17]–[23]. This method is relatively much more

efficient than the conventional full wave numerical methods. However, it is not easy

to obtain the equivalent model. In addition to that, it is difficult to adjust the

dipoles/current sources with the computation time. Furthermore, it still requires the

first prototype to be built for performing the NF scanning. Hence, it is not suitable

option for early estimation of RE from electronic devices.

In this research, novel analytical methods have been developed for RE

estimation. The analytical methods, however, require to identify the RE sources on

the electronic product. Two main sources of RE are identified in this study; namely

PCB-traces and the peripheral cables attached to PCB shown in Figure 1.4.

Figure 1.4: Radiated emissions of PCB attached with cables

Problem statement

The advanced technologies of integrated circuits as well as the ever increasing of

clock speed toward the Gigahertz range have enhanced the trends to smaller size

electronic devices. As a result, many electronic devices can work in less space which

increases the probability of EMI occurrence. So, the circuit designers must take into

the account the EMC-RE regulatory standards to avoid the excessive EMI above the

standard limit.

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Several solutions are available for compliance of the electronic products with

EMC-RE standards. However, solutions implemented at the end of the design cycle

resulted in product delays, as well as added cost [24]–[26]. Figure 1.5 shows that

implementations of EMC in the product phase are expensive and rather iterative

since it require fabricating the first prototype for measurement of RE. In contrast to

that, the implementation of EMC at the design stage is relatively cheaper than at the

production phase. In addition to that, many solutions and techniques are available at

the design stage of the electronic device [27]. Therefore, it is becoming critically

important to include EMC very early in the design phase of high speed systems.

Hence, properly designed PCB’s can offer a cost-effective approach for achieving

EMC compliance.

In the high frequencies, which may be existed due to the harmonics of the

digital signal outside the operating frequency, PCB has dimensions of the order of

several wavelengths and therefore it produces a significant amount of emissions.

Therefore, it is important to focus on the issue of EM emissions from PCBs to

provide a simple measure of the EMC based performance of PCB to the circuit

designers.

Figure 1.5: Cost of EMC implementation on the electronic product [27]

As described previously, the measurement of RE from PCB is not proper

option since it requires the first prototype to be built which may be involved in the

redesign process resulting in more delay in product marketing and increasing in the

unit cost. So, it cannot be employed for early prediction of RE from PCBs.

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Alternatively, 3D full wave numerical solvers can estimate the RE of PCB

accurately. Unfortunately, it still requires intensive computational time and powerful

resources considering the complexity of the today’s high speed PCBs. Therefore, it is

essential to develop new modelling approach which can estimate the RE from PCB

accurately and efficiently.

On PCB level, there are many sources of the unintentional RE located on

PCB such as the ICs, PCB-traces and PCB-attached cables. However, the two

predominant sources of RE on PCB are the PCB-traces and the PCB-attached cables

as shown in Figure 1.6. In high speed PCBs, the traces are electrically long and thus

become efficient radiators of EM energy. Therefore, it is important to model and

estimate the RE produced from these traces.

In addition to that, PCBs are mostly attached with cables to act as interfacing

and communication cables with other devices. These cables are well-known as a

significant source of the unintentional common-mode RE. Thus, it is important to

take into the account the emission produced from the cables attached to PCB to avoid

the product failure due to these cables.

Figure 1.6: Basic idea of the proposed models

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This research clearly address the issue of RE from PCB-traces and PCB

attached cables. First, a simple circuit consists of source trace and load is analysed.

Once the RE of this PCB-trace is estimated, the same idea can be employed to

compute the RE of PCB fully populated by traces. Finally, the compliance of EMC-

RE of entire product can be checked based on these proposed models as shown in

Figure 1.6.

Objectives of the research

In this research, novel models are proposed for estimating the RE from electrically

long PCB-traces as well as from cables attached to PCB. Specifically, the major

objectives of this study are:

i. To investigate and apply novel analytical methods for estimating the RE from

electrically long PCB-traces.

ii. To develop novel analytical methods for computing the RE from cables

attached to electrically long PCB-traces

iii. To build neural network model for estimating RE of PCB-traces for non-

Quasi-TEM current distribution.

iv. To verify and validate the proposed models using HFSS simulation and

measurements in SAC.

Scope of the research

The scope of this research involved analysis, modelling, analytical study, simulation

and measurement. In order to achieve the objectives, more constraints are applied to

limit the scope as follows:

i. The PCB traces and the attached cables are electrically long (trace length >>

wavelength).

ii. PCBs under test are designed and fabricated with two configurations (single

sided and double-sided PCB) according to the available resources and

facilities.

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iii. The cables attached to PCB-traces are connected at the source / load junctions

only according to the practical design of PCB-cables structures

iv. The RE of PCB-traces is studied analytically within the frequency range from

30 MHz to 3GHz to satisfy the condition of electrically long PCB-traces and

the condition of QTEM operation mode (cross-section of PCB-trace is short

compared to wavelength).

v. The effects of PCB coupling to grounding are omitted.

Significance of the research

Any electronic product needs to be electromagnetically compatible with EMC

regulations before it can be sold to the consumer. If the product failed to satisfy the

RE standards, it will require second cycle of redesign and validation. This process is

rather iterative and thus it increases the unit cost and minimizes the market sharing.

Therefore, the estimation of PCB RE would give more options in the design

modification as well as to avoid many of RE issues.

This research work focuses on developing new analytical modelling methods

for estimating the RE of electrically long PCB-traces based on transmission-line

theory, dipole antenna and travelling-wave antenna. Additionally, novel methods are

developed for estimating common-mode RE from cables attached to PCB using

imbalance difference model and asymmetrically dipole antenna. These proposed

methods can be employed to overcome many realistic problems of PCB-cables

structure at reasonable computational time with high level of accuracy comparing to

the experimental results.

Organisation of the thesis

This thesis consists of 7 Chapters. Each chapter is described briefly as follows:

Chapter 1 presents the background of the study, the problem statement, the objective

of the study, scope and limitations of the research work, and the significance of the

research.

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Chapter 2 describes critically the latest literature review of this research and specifies

the research scope. Therefore, this research work is comparable with any other new

research around the world.

The research methodology is given in Chapter 3 which details the approaches taken

to introduce novel techniques including analyses, modelling, simulation and

measurement in the semi-anechoic chamber.

Chapter 4 describes the analytical method to determine the RE from PCB traces. It

presents in details the analytical method for computing both DM and CM RE. This

chapter describes also the ANN model for estimating the total RE.

The methods for computing the RE of cables attached to PCB are given in Chapter 5.

A detailed analytical solution has been presented in this chapter for computing RE

from one or two cables attached to PCB.

Chapter 6 discusses the analysis of the research results based on the works

introduced in Chapter 4 and Chapter 5. Detailed analysis of all the proposed models

are presented and verified by comparing with both 3D full wave HFSS simulation

results and the results taken from measurement in SAC.

Chapter 7 provides a conclusion and details on future work, on how to improve this

research, and gives suggestions for forthcoming improvement to the research.

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2CHAPTER 2

LITERATURE REVIEW

This chapter describes in details the literature related to EM emissions of PCB. First,

a brief description of the basic radiators such as dipole antenna has been presented.

Secondly, the main sources of EM emissions on PCB are identified; namely PCB

traces and PCB-attached cables. Third, the previous studies for estimating the

emissions produced from the PCB-traces and PCB-attached cables are presented. At

the end of this chapter, a quick review of the imbalance difference model and the

artificial neural network are introduced to address the gap of this research work.

Fundamentals of the radiating systems

Basically, EM emission is generated due to the time-varying current flowing through

the electronic devices. However, these emissions can be produced from intentional

radiators such as dipole antenna or from unintentional radiators such as the PCBs that

are designed for specific application. Therefore, it is important to give a brief

description of the basic intentional radiators for better understanding of the EM

emissions of PCBs. The most important radiating antennas such as infinitesimal

dipole, short dipole and long dipole antennas are described in this section.

2.1.1 Infinitesimal dipole

For a z-directed dipole located with current I and length dl at the origin of the

coordinate system, as shown in Figure 2.1, the vector potential 퐴(푟) can be

expressed as [28]:

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퐴(푟) =휇4휋

퐽 푟 푒

푟 − 푟 푑푉 =

휇 퐼푑푙4휋

푒푟 푍

(2.1)

x

z

dlI

H

ErE

y

r

Figure 2.1: A z-directed electric point dipole [28]

where

k is the free space wave number (k=2π/λ)

λ is the wavelength,

푟, 푟 are the vector and scalar distance from the antenna respectively

휇 is the free space permeability

푍 is the characteristic impedance of the antenna

퐼 is the flowing current on the antenna

푑푙 is the antenna length

퐽 is the electric current density source.

The electric (퐸) and magnetic field (퐻) can be obtained based on the following

expressions [28]

퐻 =1휇 ∇ × 퐴

(2.2)

퐸 =1

푗휔휀 ∇ × 퐻 (2.3)

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D

Based on (2.1) and (2.2), the electric field components of the point dipole are written

as [28]

퐸 = −퐼푑푙푘 푒

4휋 푍 sin휃 1푗푘푟 +

1(푗푘푟) +

1(푗푘푟)

(2.4)

퐸 = −퐼푑푙푘 푒

4휋 푍 2 cos 휃 1

(푗푘푟) +1

(푗푘푟)

(2.5)

퐻 = −퐼푑푙푘 푒

4휋 sin휃 1푗푘푟 +

1(푗푘푟)

(2.6)

where the free space impedance 푍 = 휇휀 =377 Ohm. The other components

퐸 ,퐻 and 퐻 are zeros.

According to (2.4), (2.5) and (2.6), it is observed that the radiated field

depends on the distance from the source to the observation point. Considering

radiation source with the largest dimension D, the emissions can be divided into

three regions, namely the reactive near-field ( 362.0 Dr ), radiating near field

( 23 262.0 DrD ) and far field ( 22Dr ) starting from closest point to

source to the farthest point [28], as shown in Figure 2.2. In fact the boundaries

between the regions are only vaguely defined and changes between them are gradual.

Figure 2.2: Definition of near-field and far-field regions [28]

The wave impedance, which is the ratio of the electric and magnetic field,

also depends on the distance from the source. In the far field ( 3r ) , only the

r

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radiating term ( r1 term) of the field is significant so the electric and magnetic field

is related by the free space wave impedance as [28]

푍| ≫ =퐸퐻

≫= 푍

(2.7)

But in the near field, the wave impedance varies widely and depends on the

characteristics of the source. It can be expressed as [28]

푍 =퐸퐻 = 푍

1 − 푗 1(푘푟)

1 + 푗 1(푘푟)

(2.8)

Figure 2.3: Wave impedance of electric and magnetic sources [28]

It is well-known that the wave impedance near the electric source is very high

whereas the wave impedance near the magnetic source is very small as shown in

Figure 2.3. In the NF situation, the characteristics of the source are reflected in the

EM wave properties. But in the far-field there is nothing in the field properties to

identify the characteristics of its source. Thus, the electric/magnetic field components

in the far-field region ( 1kr ) can be simplified and approximated by [28]

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퐸 ≅ 푗휂푘퐼푑푙푒

4휋푟 sin휃

(2.9)

퐻 ≅ 푗푘퐼푑푙푒

4휋푟 푠푖푛 휃

(2.10)

where η is intrinsic impedance (377 ohm for free space). The other components

퐸 ≅ 퐸 = 퐻 = 퐻 are approximately zero.

2.1.2 Short dipoles

Commonly, the dipoles are classified based on the ratio between wavelength and the

physical length of the dipole. While the infinitesimal dipole has a length 50dl ,

the short dipoles are that dipoles with length 5010 dl .

x

z

y

r

2l

2l

R),,( rP

zd

Figure 2.4 : A z-directed short dipole [28]

Following the same procedure in Section 2.1.1, the far-field of both the

electric and magnetic fields can be written as [28]

퐸 ≅ 푗휂푘퐼푙푒

8휋푟 sin휃

(2.11)

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퐻 ≅ 푗푘퐼푑푙푒

4휋푟 sin휃

(2.12)

The other components 퐸 ≅ 퐸 = 퐻 = 퐻 are approximately zero.

2.1.3 Long dipoles

The long dipoles are those dipoles with length greater than one tenth wavelength and

no longer than one wavelength. Although the near field and far fields of the dipoles

can be expressed analytically, it is very difficult to be obtained for the near fields. In

addition to that, the most important in EMC is the maximum emission at 3 or 10 m

far from the source which is in the far-field region. Therefore, we focused on the

electric field in the far-field region. Following the same procedure in Section 2.1.1,

the electric field can be given as [28]

퐸 ≅ 푗휂퐼푒

2휋푟 cos 푘푙

2 cos휃 − cos 푘푙2

sin휃 (2.13)

The length is varied with maximum one wavelength. However, a special case

can be extracted by replacing the dipole length by half-wave length 2l . This

antenna is widely known as half-wavelength dipole antenna which its electric field in

the far-field region is expressed as [28]

퐸 ≅ 푗휂퐼푒

2휋푟 cos 휋

2 cos휃sin휃 (2.14)

Measurement and modelling of EM emissions of PCB

In Section 1.3, a brief introduction has been presented for the measurement and

modelling methods of RE. In this section, a detailed description of several

measurement and modelling methods are introduced. In fact, the modern PCBs

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compose of ICs, active and passive components, connecting traces, as well as

input/output ports and attached cables.

Although the PCB may function properly as it is designed to do, it may

produce unintentional emissions which may disturb the functional operation of the

other nearby devices [6]. So, the electronic systems must satisfy all the design

requirements not only the functional operation but also controlled emissions. Thus

emissions from electronic equipment have to be compliant with the limits defined in

national and international EMC standards [29]. Hence, those emissions can be

characterized using measurement or modelling methods [30] as illustrated in the next

sections.

2.2.1 Measurement of EM emissions of PCBs

Several measurement methods of RE from PCBs are available in the literature such

as TEM cell method [31], [32] and reverberation chamber method [33] and SAC

method. However, each method has its own measurement set-up as described in the

next sub-sections.

2.2.1.1 TEM cell method

The emissions of a device under test (DUT) can be measured inside TEM cell, as

shown in Figure 2.5 in the frequency range from 150 KHz to 1 GHz, and can be

extended beyond 1 GHz with gigahertz TEM (GTEM) cell [31], [32].

In this method, the DUT is positioned in the top or bottom of the TEM cell.

The RF voltage detected by TEM cell is then passed to the spectrum analyser via pre-

amplifier. The EM emissions of the DUT can be then computed based on the

detected RF voltage. However, this method requires performing the measurement

with at least two orientations of DUT to capture the total emission which is time

consuming process.

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Figure 2.5: Measurement set-up of EMC-RE using TEM cell [31]

2.2.1.2 Near-field scanning method

The other option for measurement of RE is to perform near-field scanning [34] of the

DUT at the distance less than one sixth of the wavelength. During the scanning

process, the magnetic probe is positioned closer to the DUT to detect the radiated

signals as illustrated in in Figure 2.6.

Figure 2.6: Measurement set-up of RE using NF scanning method [34]

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The outputs of the probes are then converted to a 2D field map showing the

field strength distribution. Using NF scanning results, the far-field electric field can

be computed using one of the transformation techniques [35]–[37].

2.2.1.3 Radiation pattern measurement method

This method is widely used for evaluating EMC-RE because the standard test EN-

55022 [9] adopted this method for measuring the RE of DUT. Based on FCC and

CISPR 22, the DUT is placed at 10m, or 3m in case of high ambient noise level far

away from the receiving antenna. A biconical antenna is used as a receiving antenna

for frequencies below 1 GHz. For frequencies above 1 GHz, a log-periodical antenna

or a horn antenna should be used. The DUT is mounted on a turntable that can rotate

through 360° to find the maximum emission. The receiving antenna is scanned in

height from 1m to 4m to find the maximum level of RE as shown in Figure 2.7. The

radiation pattern of DUT is then obtained by varying the antenna azimuth and

polarization through 360° during the measurement.

Figure 2.7: Measurement of RE using radiation pattern method

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2.2.1.4 Reverberation chamber method

In this method, the maximum REs are computed by measurement of the total

radiated power (TRP) of a DUT instead of direct measurement of the electric field.

For the purpose of measurement of TRP, an electrically large reverberation chamber

is used. This chamber is basically highly conductive overmoded enclosed cavity

equipped with one or more metallic tuners/stirrers [33] to achieve a statistically

uniform and isotropic electromagnetic environment as shown in Figure 2.8.

The stirrers are adjusted to rotate very slowly compared to the sweep time of

the EMI receiver to obtain sufficient number of samples. During a cycle period of the

stirrers, the maximum received power or averaged received power can be measured

and recorded. Those recorded signals are then converted to the TRP and the free

space field strength generated from the DUT.

Although all the measurement methods provide accurate results, the main

disadvantage is that they require fabricating the first prototype to be used as DUT.

Thus, they are not suitable option for early prediction of RE from PCBs. So, it is

crucially important to employ another modelling approach for estimating the

emissions of PCBs based on the PCB design specifications before the product is

fabricated to ensure time and cost savings.

Figure 2.8: Measurement of RE using reverberation chamber method [33]

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2.2.2 Modelling methods for EM emissions of PCB

Today, PCBs in high speed devices have a very complex design. Therefore, it is

difficult and expensive to modify the PCB-layout at the production phase of the

electronic devices. Thus, it is important to consider EMC-RE at the design process.

Modelling methods are preferable option since they can estimate the RE of the

electronic products earlier in the design phase. This section describes briefly several

modelling methods for computing the RE of PCB such as design-rule checking,

equivalent modelling and analytical modelling.

2.2.2.1 Full-wave numerical modelling

In this method, the entire structure of PCBs must be modelled by discretizing the

space in terms of grid or mesh and then solving numerically Maxwell equations in

free space, dielectrics, and conductors. Actually, the physical geometries of all the

elements (dielectrics, traces, excitations, loads, etc.) are modelled and specific

attention is given for the material properties and frequency.

The models are then solved numerically using one of the computational

electromagnetics (CEM) methods [38]. Many numerical methods have been

developed over the past decades for solving the Maxwell equations either in

differential forms such FDTD [14], FEM [39], and transmission line matrix (TLM)

[40] or integral forms such as MoM [41] and PEEC [42]. Many commercial tools

have adopted these numerical techniques such as FEKO which adopted MoM [43],

FEM based HFSS [16] and CST which employed hybrid methods [44].

For better illustration of these numerical techniques, detailed descriptions are

introduced for each technique as follows:

i. FDTD technique

In this technique, the structure is chunked into small cells each of which do not

exceed one tenth of the wavelength of the maximum operating frequency. Figure 2.9

shows the discretized cell, which is defined as Yee cell [45]. The EM field

components ZYXZYX E,E,E,H,H,H are given as below.

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⎝

⎜⎜⎜⎛휕퐻휕푦 −

휕퐻휕푧

휕퐻휕푧 − 휕퐻

휕푥휕퐻휕푥 − 휕퐻

휕푦 ⎠

⎟⎟⎟⎞

=

⎝

⎜⎜⎛휀

휕퐸휕푡

휀휕퐸휕푡

휀 휕퐸휕푡 ⎠

⎟⎟⎞

+퐽퐽퐽

(2.15)

⎝

⎜⎜⎜⎜⎛휕퐸푌휕푧 − 휕퐸푍

휕푦휕퐸푍휕푥 − 휕퐸푋

휕푧휕퐸푋휕푦 − 휕퐸푌

휕푥 ⎠

⎟⎟⎟⎟⎞

=

⎝

⎜⎜⎛휇

휕퐻휕푡

휇 휕퐻휕푡휇 휕퐻휕푡 ⎠

⎟⎟⎞

(2.16)

where J is the electric current density, is the material permittivity and is the

material permeability.

Figure 2.9: Electromagnetic field components of one cell using FDTD [26]

It is necessary to compute the electric and magnetic field components of the

entire structure for solving the equations (2.15) and (2.16). FDTD is a good option

for computing the maximum RE of PCBs. However, it is not proper option since it

needs to be involved in post processing to obtain the far-field electric field.

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i. FEM technique

FEM is the second technique for finding the maximum RE of PCBs. The entire

structure is modelled by large number of finite elements each of which has its

modelling equation. Generally, FEM method can solve more complex geometries

and more complex loading conditions comparing to the other methods such as

FDTD. But, it is time-consuming process especially for complex structures of PCBs.

ii. MoM technique

In contrast to the FDTD and FEM, this technique does not desire more computational

process of the structure. Instead, it desires to calculate the boundary values only. As a

result, it is efficient for solving problems with small surfaces. However, it is also not

preferred for estimating the RE from PCBs since it still requires intensive

computational time especially for large structures.

Although the full wave numerical solvers can provide accurate estimation of

EM emissions, we can conclude that these numerical solvers are not well-suited

option due to the huge requirement for computer speed and memory to estimate the

RE of PCBs.

2.2.2.2 Design rule checking

Design rule checker is software that can check whether the rules of design are

violated or not in the same manner as professional EMC expert does. It reads the

board layout information from automated board layout tools (such as Allegro, Protel,

Board Station, etc.) and checks if certain EMC design guidelines have been

breached. This method does not provide quantitative estimation of EM emissions

from the PCB. However, it gives indication of the overall correctness of the design.

DRC can also help to identify and locate the potential source of emissions on the

designs which is very difficult to point it out. In the recent days, several DRC

softwares are available such as EMI Stream [46], EMSAT [47], and Zuken CR-5000

Lightning EMC [48].

Although these DRC softwares are easy to use even for non-expert peoples,

they can do not provide any quantitative estimation of RE from the device. In

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