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Sensors and Actuators A 165 (2011) 200–206

Contents lists available at ScienceDirect

Sensors and Actuators A: Physical

journa l homepage: www.e lsev ier .com/ locate /sna

assive single chip wireless microwave pressure sensor

mr Ibrahim, D.R.S. Cumming ∗

epartment of Electronics and Electrical Engineering, University of Glasgow, Rankine Building, Oakfield Avenue, Glasgow G12 8LT, UK

r t i c l e i n f o

rticle history:eceived 12 May 2010eceived in revised form 7 October 2010ccepted 31 October 2010vailable online 9 November 2010

a b s t r a c t

A novel micromachined passive wireless pressure sensor is presented. The device consists of a tunedcircuit operating at 10 GHz fabricated on to a SiO2 membrane, supported on a silicon wafer. A pressuredifference across the membrane causes it to deflect so that an antenna circuit detunes. The circuit isremotely interrogated to read off the sensor data wirelessly. The chip area is 5 mm × 4 mm and the mem-brane area is 2 mm2 with a thickness of 4 �m. Two on-chip passive resonant circuits were investigated:

eywords:ireless pressure sensor

esonance frequencyembraneicromachining

a meandered dipole and a zigzag antenna. Both have a physical length of 4.25 mm. The sensors show ashift in their resonant frequency in response to changing pressure of 10.28–10.27 GHz for the meandereddipole, and 9.61–9.58 GHz for the zigzag antenna. The sensitivities of the meandered dipole and zigzagsensors are 12.5 kHz/mbar and 16 kHz/mbar respectively.

© 2010 Published by Elsevier B.V.

igzag antennaeander dipole antenna

. Introduction

Wireless pressure sensors are increasingly being used inany practical applications such as automotive and biomedical

pplications [1,2]. Reliable, easy to fabricate, and low cost micro-lectromechanical systems (MEMS) are required as the marketemand increases. Most work on pressure sensors concentratesn two transduction techniques based on the use of a deformableembrane: piezoresistive and capacitive sensors. The capacitive

ressure sensor, when compared to the piezoresistive sensor hasigher sensitivity to changes in pressure, lower thermal drift, and

ower noise. Both sensors require read out circuits in close prox-mity to the sensing element [3], or on a separate chip [4]. Manypplications require that sensors should require little or no poweronsumption over extended in-service periods [5]. This is particu-arly challenging for wireless devices.

Wireless systems can be made to operate either actively or pas-ively. In an active system, the sensor will be connected to anlectronic circuit that requires power to operate its componentarts. In a passive system, the sensor will be limited to simple com-onents such as inductors and capacitors, and will need no power.ctive systems are desirable where communication is needed over

elatively long distances, or sophisticated wireless technology isequired.

Such devices must be either battery operated, or be capable ofower scavenging from a reader device over very short working dis-

∗ Corresponding author. Tel.: +44 0141 5233.E-mail address: d.cumming@elec.gla.ac.uk (D.R.S. Cumming).

924-4247/$ – see front matter © 2010 Published by Elsevier B.V.oi:10.1016/j.sna.2010.10.020

tances (typically a few centimeters). Passive systems also requireenergy from a reader device in order to be read, but the powerlevels can be very small. Passive low frequency devices have beendemonstrated based on the use of lumped-element components(inductors and capacitors) to achieve inductively coupled sensorsfor biomedical applications in [6,7], and miniaturization on to singleintegrated circuit devices operating at 60–75 MHz, 750–950 MHz,and 11–13 MHz has also been reported in [8–10].

In this paper we demonstrate an alternative method to using lowfrequency inductively coupled devices that is to use of microwavesignals to read the data from a remote sensor that has been specifi-cally designed to operate at the required frequency. Fig. 1(a) showsthe basic principle of operation in schematic form. The sensor iscomposed of a deformable membrane, as is usual, that has a pla-nar metal antenna fabricated on to its surface extending to thesupporting silicon that surrounds the membrane.

The antenna is designed to have a resonant frequency in theband of interest. As the membrane is deformed, the antenna expe-riences a slight change in its physical characteristics, causing theantenna resonance to shift. The sensor is read by directing a beamof microwave radiation from, for example, a horn antenna towardsthe sensor and recording the reflected spectrum. By this means, theresonant frequency of the antenna is measured. We have investi-gated in detail the methods by which such a passive wireless sensormay be implemented. The design frequency was chosen to be in

the microwave X-band (8–12 GHz) since it is widely used for arange of microwave technologies including RADAR and telecom-munications [11]. Furthermore, the wavelength is relatively small(�0 = 30 mm in free space) so that there is the possibility of mak-ing relatively small single chip devices with appropriate antenna

A. Ibrahim, D.R.S. Cumming / Sensors and Actuators A 165 (2011) 200–206 201

or and (b) current distribution of a dipole antenna.

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antennas on silicon are very useful for high frequency commu-nications. The antenna that was studied for this work consists of300 �m long sections (ls) laid out as shown in Fig. 3. Additionalparameters were the section gaps WZ = 460 �m, and the distance

Fig. 1. (a) Working principle of proposed sens

esign. As a further consideration, the substrate was chosen to beilicon since it is cheap, widely available, and there are many easyo use fabrication processes available.

One of the most common types of antenna is the half wavelengthipole antenna. The amplitude of the current in a dipole antennaas the profile of a half-cycle of a sine wave, reaching a peak athe centre, as illustrated in Fig. 1(b), and the length of a simpleipole from end-to-end is L = �0/2 where �0 is the wavelength athe resonant frequency.

A simple half-wave dipole offers a balanced feed but its lengthor 10 GHz operation in free space (approximately 15 mm) makes itnsuitable for integration as a small on-chip sensors. The meanderipole and zigzag antennas are excellent candidates for investiga-ion since they allow for miniaturization, low fabrication cost, andan be excited by single or differential feeds. Integrated log-periodicigzag antennas and meander antennas have been demonstratedn low resistivity silicon substrates and 130 nm CMOS processesn [12,13], and meander antennas have been shown to work on

suspended, thin, low loss, benzocyclobutene (BCB) membrane14]. We show that these antenna concepts are suitable for makingn integrated microwave readable pressure sensor as a conse-uence of their inherent properties and operation. We furtherresent detailed design and fabrication methods, and data fromeveral micro-machined antennas is analyzed to evaluate sensitiv-ty to pressure, return loss, impedance bandwidth, and resistivity.he final devices are compact and can be readily fabricated on amm × 4 mm chip for operation at 10 GHz.

. Antennae design

The elementary requirements for the dimensions of a simpleipole antenna may be easily calculated. For a printed dipole onhe surface of a solid substrate the resonant frequency is given by:

c = c

2L√

εeff

(1)

εeff is the effective dielectric constant of the media surroundinghe dipole, fc is the resonant frequency and c is the speed of lightn vacuo. It is therefore possible to make a dipole antenna with< �0/2 by using εeff > 1. However, another technique for reducinghe length is to meander the branches of the dipole. By introducinghe meander, the inductance of each arm of the dipole antenna isncreased, thus adding to the effective electrical length, L, of the

tructure. The total inductance is made up of the sum of the induc-ances for each meander, as shown in Fig. 2. The input impedance,0, individual meander inductance, and overall inductance can alle controlled in the design process by selecting values for thearameters WS (meander separation), WC (conductor width) and

Fig. 2. A meander dipole antenna.

a (meander pitch). The sensor devices are intended for wirelessoperation and interrogation by free-space radiation. However, aspart of our design and test procedure we desire to be able to makeprober measurements to a calibrated 50 � vector network analyzer.Therefore, as an additional design constraint we require Z0 of theantenna to be 50 �. Z0 is given by [15]:

Z0 = ZC

�log

2a

WA(2)

The printed antenna had a length of 4.25 mm (<�0/2),WS = 200 �m, WA = 650 �m, distance between arms DM = 250 �mnarrowing to 100 �m between the feed lines, and metal (alu-minium) thickness of 1 �m. ZC refers to intrinsic impedance of theantenna.

Similarly, another method for designing a short-compact dipoleantenna is to bend the dipole into a zigzag shape. Such zigzag

Fig. 3. A zigzag antenna.

2 rs and Actuators A 165 (2011) 200–206

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etween arms DZ = 470 �m, narrowing to 50 �m between the feedines. Additional parameters shown in Fig. 3 are: WB = 100 �m and

L = 150 �m. Based on previous results obtained for similar anten-ae on a CMOS chips, the angle between sections was chosen toe 30◦ [12]. The separation between the feed lines was 50 �m. Thentenna width, WA, was 610 �m and the total length was 4.25 mm,s for the meander antenna.

For both the meander and zigzag antennae a membrane wastched into the silicon substrate from behind. In each case thentenna is part on and part off the membrane as illustrated by theoxes shown in Figs. 2 and 3. The membrane serves two purposes.t is necessary for the antenna to be able to flex in response to aressure gradient, and by having part of the sensor on the mem-rane, the losses generated in the silicon substrate are reduced. Bynly putting part of the antenna on the membrane the overall dipoleength was reduced. High Frequency Surface Simulator HFSSTM wassed for designing both antennae.

. Antennae fabrication

Fig. 4 shows the fabrication steps. A 2 �m thick SiO2 layeras deposited by plasma enhanced chemical vapour deposition

n to a 3 k�-cm, 300 �m thick, 4′′ silicon wafer. A double-layerf polymethylmethacrylate (PMMA) electron-beam resist, of lowolecular weight (1.2 �m) followed by high molecular weight

0.9 �m), was spin-cast on to the chip, and pre-baked at 180 ◦C for20 min after each layer. Direct-write electron-beam lithographysing a Vistec VB6 was used to delineate the antenna patterns fol-

owed by development in 1:1 MIBK:IPA for 60 s at 23 ◦C. 1 �m ofluminium was deposited using a sputter coating followed by lift-ff in acetone by soaking for 3 h. Aluminium was chosen as thentenna material because of its ease of deposition, good conductiv-ty, and low temperature processing.

After patterning the antennae, the chips were prepared for etch-ng to form the membrane. The etched area was located behindhe feed points of the antenna in order to minimise the dielectricosses in this critical section [16]. A thick layer of photoresist wassed to protect the un-etched areas. AZ4562 was spin-cast andre-baked at 90 ◦C and the mask was exposed for 30 s in a Sussicrotec mask aligner. The resist was developed for 4 min in AZ

00 K. The membrane was then etched using an inductive-coupledlasma in a STS 100 with C4F8/SF6/O2 gases for 68 min at an etchate of 4.38 �m/min. The etch depth was 298 �m, thus leaving aesidual support layer under the membrane of 2 �m of silicon torovide greater strength (devices without this layer broke easily).

ig. 5. Wafer level fabrication of meander dipole (right) and zigzag antenna (bottom-leftograph of etched area of chips. (For interpretation of the references to colour in this figu

Fig. 4. Fabrication process of devices. (a) SiO2 deposition. (b) Double layers of PMMAcoating. (c) Development. (d) Al metallization. (e) Lift-off. (f) AZ4562 coating. (g)Development. (h) ICP-etching.

The results from etch processing are shown in Fig. 5. The devicewafers are sawn to make 5 mm × 4 mm sensor chips from a finalwafer.

4. Simulation and measurement results

4.1. Probed antennas characteristics and discussions

Several micromachined devices were tested and the char-acteristic impedance of the antennae at their feed-points wascharacterized by making on-wafer measurements to the litho-graphically defined devices using an Agilent E8362B network

analyzer. The chips were mounted on an absorbing chuck to preventreflections from surrounding metal parts. The measurement resultswere compared directly with the design simulations as shown inFigs. 6 and 7 for both the meander and zigzag antenna-based sen-sors. For comparison we present antenna characteristic for before

). Sensing area is defined by red lines for clarity, and bottom-right SEM micropho-re legend, the reader is referred to the web version of this article.)

A. Ibrahim, D.R.S. Cumming / Sensors and Actuators A 165 (2011) 200–206 203

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1.0–1.4 bar. Using this simple method, changes in the resonant fre-

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ig. 6. Simulated and measurement results for meander dipole antenna. S11 (top)nd antenna response normalized to 50 � input impedance.

nd after the etching of the membrane. Table 1 summarises theesults at the resonant frequency for the meander dipole and zigzagntennas.

As was to be expected the results show very strongly the effectf the presence of the silicon substrate beneath the antenna. Datarom either antenna design prior to etching shows that the reso-ant frequency is shifted to a lower frequency, consistent with theresence of a material with a dielectric constant >1 (Eq. (1)). Afterhe formation of the membrane below the central feed-point of thentennae the resonant frequency increases considerably by 0.7 GHzor the meander antenna and 0.4 GHz for the zigzag antenna.

The losses of the antennae are reduced after the etch. This isgain consistent with the removal of a portion of lossy siliconubstrate. The improvement in the loss is 14 dB for the mean-er antenna and 5 dB for the zigzag device. A change of the orderf 10% in Re[Z0] is seen between before and after etching. How-ver, as expected, the change in Im[Z0] is very large, consistentith the change in the return loss. Final, the band-width of the

ntenna, as measured in the frequency response of the character-stic impedance also improves dramatically after the membraneas been etched. There is a slight discrepancy between simulatednd measured results that were obtained using a dielectric con-

able 1imulated and Measured results for micromachined antennas.

Parameters

Return loss (dB) fc (GH

Meander antennaSimulated on membrane −40.45 10.22Measured prior etching −26.80 9.90Measured on membrane −36.80 10.60Zigzag antennaSimulated on membrane −37.40 9.40Measured prior etching −22.50 9.60Measured on membrane −27.65 10.00

Fig. 7. Simulated and measurement results for zigzag antenna. S11 (top) and antennaresponse normalized to 50 � input impedance.

stant for the composite SiO2/Si membrane of 4.1. The measuredresonant frequencies of the devices differ by 3.5% and 6% for mean-der and zigzag devices respectively from the simulated results onmembrane. We attribute this discrepancy to a potential variationbetween the properties of the aluminium that was sputtered, thebilayer that forms the membrane and the inconsistency of siliconwafer resistivity, which varies between 3 and 5 k�, and finally dis-crepancy associated with probes and the surroundings metal parts.

4.2. Probed antenna-pressure measurements

To study the effects of membrane deflection on antennas, a sim-ple un-calibrated experiment was carried out as shown in Fig. 8(a).A jet of nitrogen is blown onto the surface of the membrane usinga plastic pipette positioned approximately 1 mm from the surface.The gas flow-rate is controlled using a regulator connected to a gascylinder. The pressure at the output of the regulator is in the range

quency of 10.6–10.62 GHz for the meander and 10–10.325 for thezigzag antennas was achieved. This is due to mechanical changesin physical dimensions of antennas (WS, WZ) and gap between feedlines. As the data in Fig. 8(b and c) shows the meander antenna

z) Impedance (�) −10 dB bandwidth (%)

50.85 + j 0.435 32.0055.95 − j 0.475 43.2051.2 + j 0.050 44.50

51.4 + j 0.10 31.0057.35 − j 3.30 48.0056.15 + j 0.09 52.00

204 A. Ibrahim, D.R.S. Cumming / Sensors and Actuators A 165 (2011) 200–206

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ig. 8. (a) Simple experiment setup. (b) and (c) Resonant frequency shift for the mesonant frequency increases with gas pressure, as indicated by the solid arrows.

hows a continuous shift in the resonant frequency throughouthe experimental pressure range. By contrast, the zigzag antennahows a more pronounced shift in the frequency for pressure as gasow increases. At higher gas flow there is little further shift in the

requency (Fig. 8(d and e)).

.3. Wireless pressure measurements

Calibrated wireless telemetry performance of the sensor’sehaviour was obtained by monitoring the shift in resonant fre-uency for both devices in response to an applied pressure usingmicrowave reflection experiment. A simple base-station readeras set up for use in the X-band using a horn antenna as shown

Fig. 9. (a) The membrane has higher pressure behind the m

r dipole antenna. (d) and (e) Resonant frequency shift for the zigzag antenna. The

in Fig. 9. The horn was directly connected to the Agilent E8362BPNA, and the pressure senor to horn antenna distance was 10 cm.The differential pressure between atmospheric room pressure(1000 mbar) and the back-side of the membrane was accuratelycontrolled by mounting the sensor over an aperture on a sealed cop-per adaptor. The sensor was surrounded by microwave absorbingfoam. The pressure was controlled using a regulator connected toa nitrogen gas cylinder. The pressure in the adaptor was measured

using a barometric sensor (Campbell Scientific CS100). As can beseen from the sketch of the experimental configuration, the sensormembrane is deflected outwards from the chip in this set-up, but itis expected that similar results would be obtained if a vacuum wasapplied behind the membrane and it was deflected inwards.

embrane and (b) a sketch of the experimental set up.

A. Ibrahim, D.R.S. Cumming / Sensors and

Fig. 10. Resonant frequency shift detected at horn for meander antenna (top) anddevice sensitivity (bottom).

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ig. 11. Resonant frequency shift detected at horn for zigzag antenna (top) andevice sensitivity (bottom).

The sensor’s calibrated wireless telemetry’s repeatability wasonfirmed by performing three independent measurements ofhree devices of each design. For each experiment, the interro-ation power radiated from the horn was −5 dBm at 23 ◦C. Theesults were measured for 120 s per 50 mbar during which timeo drift was observed. Figs. 10 and 11 show devices sensitivi-

ies to pressure reflections from the horn antenna. The error barshown in Figs. 10 and 11 indicate device-to-device variation thate attribute to tolerances in the photolithography and membrane

tch. The measured shift in resonant frequency of the meander

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Actuators A 165 (2011) 200–206 205

antenna is between 10.28 GHz and 10.27 GHz in response to a pres-sure range between atmospheric pressure and 1.8 bar. This givesa device sensitivity of 12.5 kHz/mbar for the device containing themeander antenna. Data acquired from zigzag antenna device showsa resonance shift from 9.6078 to 9.5809 GHz according to appliedpressure, yielding a device sensitivity of 16 kHz/mbar.

5. Conclusion

A passive wireless microwave readable pressure sensor has beendeveloped on a single chip. The devices were implemented by fabri-cating folded antenna structures on to a silicon dioxide membrane.Designs based on meander dipole and zigzag antennas were sim-ulated prior to fabrication. As expected, the antenna design wassensitive to the presence of the silicon substrate before the mem-brane was etched, and good agreement between simulation andexperiment was achieved for the completed devices. The applica-tion of a differential pressure to deflect the membrane gave riseto a shift in the resonant frequency of 10 MHz over 800 mbar forthe meander antenna. The equivalent frequency shift for the zigzagantenna was 27.8 MHz. These data translate into pressure sensi-tivities of 12.5 kHz/mbar and 16 kHz/mbar respectively. A workingwireless operating distance of 10 cm was demonstrated using a verysimple laboratory set up. The new device used electromagneticallysafe readout, and does not need a line of sight assembly and a point-ing system (as opposed to using a laser) at a widely used frequencyband. In future simple readers and smaller processing units couldbe built for use with devices in the V-band using cheap consumerelectronic parts [17].

Acknowledgment

The authors would like to thank the UK Engineering and PhysicalScience Research Council (EPSRC) for their funding and full supportfor this project.

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[4] J. Ramos, A. Luque, J.M. Quero, Integration of a MEMS capacitive sensor on awireless transmitter, in: IEEE Proc. Spanish Conference on Electron Devices,2009, pp. 330–333.

[5] Q. Huang, M. Oberle, A 0.5-mW passive telemetry IC for biomedical applica-tions, IEEE Journal of Solid-State Circuits 33 (7) (1998).

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Actuators A 165 (2011) 200–206

Biographies

Amr Ibrahim received his BSc in Physics from University of Khartoum, Sudan in 2002and his MSc from Heriot-Watt University-Edinburgh in Microsystems Technology in2006. He is currently working towards his PhD at Glasgow University-Scotland. Hisresearch interests includes on chip antennas and passive components for wirelesssensing applications.

D.R.S. Cumming received the BEng Degree from the University of Glasgow, U.K.

in 1989 and the PhD degree from the University of Cambridge, U.K., in 1993. Hehas worked variously on mesoscopic device physics, RF characterization of noveldevices, fabrication of diffractive optics for optical and submillimeter wave appli-cations, diagnostic systems, and microelectronic design. He served as a EPSRCAdvanced Research Fellow, and he is currently a Professor and Chair of ElectronicSystems at Glasgow University.