NE602 App Note

Philips Semiconductors

AN1993High sensitivity applications of low-powerRF/IF integrated circuits

1997 Aug 20

RF COMMUNICATIONS PRODUCTS

Philips Semiconductors Application note

AN1993High sensitivity applications of low-power RF/IFintegrated circuits

6–21997 Aug 20

ABSTRACTThis paper discusses four high sensitivity receivers and IF(Intermediate Frequency) strips which utilize intermediatefrequencies of 10.7MHz or greater. Each circuit utilizes a low-powerVHF mixer and high-performance low-power IF strip. The circuitconfigurations are

1. 45 or 49MHz to 10.7MHz narrowband,2. 90MHz to 21.4MHz narrowband,3. 100MHz to 10.7MHz wideband, and4. 152.2MHz to 10.7MHz narrowband.

Each circuit is presented with an explanation of component selectioncriteria, (to permit adaptation to other frequencies and bandwidths).Optional configurations for local oscillators and data demodulatorsare summarized.

INTRODUCTIONTraditionally, the use of 10.7MHz as an intermediate frequency hasbeen an attractive means to accomplish reasonable image rejectionin VHF/UHF receivers. However, applying significant gain at a highIF has required extensive gain stage isolation to avoid instability andvery high current consumption to get adequate amplifier gainbandwidth. By enlightened application of two relatively new lowpower ICs, Philips Semiconductors SA602 and SA604A, it ispossible to build highly producible IF strips and receivers with inputfrequencies to several hundred megahertz, IF frequencies of 10.7 or21.4MHz, and sensitivity less than 2µV (in many cases less than1µV). The Philips Semiconductors new SA605 combines thefunction of the SA602 and the SA604A. All of the circuits describedin this paper can also be implemented with the SA605. The SA602andSA604A were utilized for this paper to permit optimum gainstage isolation and filter location.

THE BASICSFirst let’s look at why it is relevant to use a 10.7 or 21.4MHzintermediate frequency. 455kHz ceramic filters offer good selectivityand small size at a low price. Why use a higher IF? The fundamentalpremise for the answer to this question is that the receiverarchitecture is a hetrodyne type as shown in Figure 1.

A pre-selector (bandpass in this case) precedes a mixer and localoscillator. An IF filter follows the mixer. The IF filter is only supposedto pass the difference (or sum) of the local oscillator (LO) frequencyand the preselector frequency.

The reality is that there are always two frequencies which cancombine with the LO: The pre-selector frequency and the “image”frequency. Figure 2 shows two hypothetical pre-selection curves.Both have 3dB bandwidths of 2MHz. This type of pre-selection istypical of consumer products such as cordless telephone and FMradio. Figure 2A shows the attenuation of a low side image with10.7MHz. Figure 2B shows the very limited attenuation of the lowside 455kHz image.

PRE SELECTFILTER

MXR

LO

IF FILT IF AMP DEMOD

AUDIOAND/ORDATA

SR00820

Figure 1. Basic Hetrodyne Receiver

PR

ES

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CT

OR

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TE

NU

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ION

(dB

)

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)

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–20

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–20 –10 0 +10 +20

PRESELECTORFILTER

10.7MHz IF

DESIREDFREQUENCY

LO

IMAGE

∆ PRESELECTOR FREQ (MHz)

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–30

–40

–20 –10 0 +10 +20

PRESELECTORFILTER

455kHz IF

DESIREDFREQUENCY

LO

IMAGE

∆ PRESELECTOR FREQ (MHz) SR00821

Figure 2. Effects of Preselection on Images

PRE SELECTFILTER

1ST MIXER

1ST LO

1ST IF FILTER IF AMP DEMOD

AUDIOAND/ORDATA

2ND MIXER

2ND IF FILTER

2ND LO

SR00822

Figure 3. Dual Conversion



1997 Aug 20 6–3

ZI

ZIZI

BPF

BPF

ZIZI

SR00823

Figure 4. Feedback Paths

6

7 4

8

5

2

3

1

BIAS BIAS

BIAS

BUFFER1.5k 1.5k

VCC

80k

GND

10k

20k

1.5k

SR00824

Figure 5. SA602 Equivalent Circuit

If the single conversion architecture of Figure 1 were implementedwith a 455kHz IF, any interfering image would be received almost aswell as the desired frequency. For this reason, dual conversion, asshown in Figure 3, has been popular.

In the application of Figure 3, the first IF must be high enough topermit the pre-selector to reject the images of the first mixer andmust have a narrow enough bandwidth that the second mixerimages and the intermod products due to the first mixer can beattenuated. There’s more to it than that, but those are the basics.The multiple conversion hetrodyne works well, but, as Figure 3suggests, compared to Figure 2 it is more complicated. Why, then,don’t we use the approach of Figure 2?

THE PROBLEMHistorically there has been a problem: Stability! Commerciallyavailable integrated IF amplifiers have been limited to about 60dB of

gain. Higher discrete gain was possible if each stage was carefullyshielded and bypassed, but this can become a nightmare on aproduction line. With so little IF gain available, in order to receivesignals of less than 10µV it was necessary to add RF gain and this,in turn, meant that the mixer must have good large signal handlingcapability. The RF gain added expense, the high level mixer addedexpense, both added to the potential for instabilities, so the multipleconversion started looking good again.

But why is instability such a problem in a high gain high IF strip?There are three basic mechanisms. First, ground and the supply lineare potentially feedback mechanisms from stage-to-stage in anyamplifier. Second, output pins and external components create fieldswhich radiate back to inputs. Third, layout capacitances becomefeedback mechanisms. Figure 4 shows the fields and capacitancessymbolically.



1997 Aug 20 6–4

16 15 14 13 11 10 9

87654321

12

42k

GND

42k

700

7k

1.6k40k

70035k

1.6k40k

2k4.5k

2k 8kFULLWAVERECT.

VOLTAGE/CURRENT

CONVERTER

VOLTREG

VOLTREG

MUTE QUAD

DET

BANDGAPVOLT

GND

80k 55k 55k

40k40k

80k 80k

FULLWAVERECT.

VEE

VCC

VCC

VCC

SR00825

Figure 6. SA604A Equivalent Circuit

RF

MIXER

IFT

S1

LO

IF AMP

QUADDETECTOR

AUDIOOUT

IFT

IF AMP

SR00826

Figure 7. Symbolic Circuit

If ZF represents the impedance associated with the circuit feedbackmechanisms (stray capacitances, inductances and radiated fields),and ZIN is the equivalent input impedance, a divider is created. Thisdivider must have an attenuation factor greater than the gain of theamplifier if the amplifier is to remain stable.

• If gain is increased, the input-to-output isolation factor must beincreased.

• As the frequency of the signal or amplifier bandwidth increases,the impedance of the layout capacitance decreases therebyreducing the attenuation factor.

The layout capacitance is only part of the issue. In order fortraditional 10.7MHz IF amplifiers to operate with reasonable gainbandwidth, the amount of current in the amplifiers needed to bequite high. The CA3089 operates with 25mA of typical quiescentcurrent. Any currents which are not perfectly differential must becarefully bypassed to ground. The higher the current, the moredifficult the challenge. And limiter outputs and quadraturecomponents make excellent field generators which add to thefeedback scenario. The higher the current, the larger the field.



1997 Aug 20 6–5

LO INPUT

GND

RSSI AUDIO DATA

GND

GND

OFF

ON VCC

PHILIPSNE602/604 DEMO BOARD

MUTE

RF INPUT

SR00827

SA602/604

Figure 8. Circuit Board Layout

THE SOLUTIONThe SA602 is a double balanced mixer suitable for input frequenciesin excess of 500MHz. It draws 2.5mA of current. The SA604A is anIF strip with over 100dB of gain and a 25MHz small signalbandwidth. It draws 3.5mA of current. The circuits in this paper willdemonstrate ways to take advantage of this low current and 75dB ormore of the SA604A gain in receivers and IF strips that would not bepossible with traditional integrated circuits. No special tricks areused, only good layout, impedance planning and gain distribution.

THE MIXERThe SA602 is a low power VHF mixer with built-in oscillator. Theequivalent circuit is shown in Figure 5. The basic attributes of thismixer include conversion gain to frequencies greater than 500MHz,a noise figure of 4.6dB @ 45MHz, and a built-in oscillator which canbe used up to 200MHz. LO can be injected.

For best performance with any mixer, the interface must be correct.The input impedance of the SA602 is high, typically 3kΩ in parallel

with 3pF. This is not an easy match from 50Ω. In each of theexamples which follow, an equivalent 50:1.5k match

was used. This compromise of noise, loss, and match yielded goodresults. It can be improved upon. Match to crystal filters will requirespecial attention, but will not be given focus in this paper.

This oscillator is a single transistor with an internal emitter followerdriving the mixer. For best mixer performance, the LO level needs tobe approximately 220mVRMS at the base of the oscillator transistor(Pin 6). A number of oscillator configurations are presented at theend of this paper. In each of the prototypes for this paper, the LOsource was a signal generator. Thus, a 51Ω resistor was used toterminate the signal generator. The LO is then coupled to the mixerthrough a DC blocking capacitor. The signal generator is set for0dBm. The impedance at the LO input (Pin 6) is approximately20kΩ. Thus, required power is very low, but 0dBm across 51Ω doesprovide the necessary 220mVRMS.

The outputs of the SA602 are loaded with 1.5kΩ internal resistors.This makes interface to 455kHz ceramic filters very easy. Other filtertypes will be addressed in the examples.

THE IF STRIPThe basic functions of the SA604A are ordinary at first glance:Limiting IF, quadrature detector, signal strength meter, and muteswitch. However, the performance of each of these blocksis superb. The IF has 100dB of gain and 25MHz bandwidth.This feature will be exploited in the examples. The signalstrength indicator has a 90dB log output characteristic withvery good linearity. There are two audio outputs with greaterthan 300kHz bandwidth (one can be muted greater than70dB). The total supply current is typically 3.5mA. This is theother factor which permits high gain and high IF.

Figure 6 shows an equivalent circuit of the SA604A. Each of the IFamplifiers has a 1.6kΩ input impedance. The input impedance isachieved by splitting a DC feedback bias resistor. The inputimpedance will be manipulated in each of the examples to aidstability.

BASIC CONSIDERATIONSIn each of the circuits presented, a common layout and systemmethodology is used. The basic circuit is shown symbolically inFigure 7.

At the input, a frequency selective transformation from 50Ω to 1.5kΩpermits analysis of the circuit with an RF signal generator. A secondgenerator provides LO. This generator second generator providesLO. This generator is terminated with a 51Ω resistor. The output ofthe mixer and the input of the first limiter are both high impedance(1.5Ω nominal). As indicated previously, the input impedance of thelimiter must be low enough to attenuate feedback signals. So, theinput impedance of the first limiter is modified with an externalresistor. In most of the examples, a 430Ω external resistor was usedto create a 330Ω input impedance (430//1.5kΩ). The first IF filter isthus designed to present 1.5kΩ to the mixer and 330Ω to the firstlimiter.

The same basic treatment was used between the first and secondlimiters. However, in each of the 10.7MHz examples, this interstagefilter is not an L/C tank; it is a ceramic filter. This will be explained inthe first example.



1997 Aug 20 6–6

After the second limiter, a conventional quadrature detectordemodulates the FM or FSK information from the carrier and a

simple low pass filter completes the demodulation process at theaudio outputs.

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9101112131416 15

1 2 3 4

568 7

0.1µF

0.1µF

0.1µF

0.1µF

0.1µF

0.1µF

0.1µF

0.1µF

0.1µF

51Ω

10µF

0.28µH

12µH

1.5k

+6V

LO INPUT10.7MHz

CERAMICFILTER

10.7MHzCRYSTAL

FILTER10M15A

SA602

RF INPUT

220pF

47pF

3.9pF

18pF

56pF

430

SA604A

MUTE +6V

RSSIAUDIO DATA

1nF 91k

15nF

150pF

1pF 180pF1.1µH

430

SR00828

Figure 9. SA602/604A Demonstration Circuit with RF Input of 45MHz and IF of 10.7MHz ±7.5kHz

0

10

20

3–

40

10.5 10.7 10.9

RE

LAT

IVE

FIL

TE

R A

TT

EN

UA

TIO

N (

dB)

FREQ|UENCY (MHz)

CRYSTALFILTER

CERAMICFILTER

SR00829

Figure 10. Passband Relationship



1997 Aug 20 6–7

• 45MHz RF input • 10.7MHz IF• 55.7MHz LO • 15kHz IF BW (odBm) • 7kHz deviation

5.0

4.5

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(VO

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AUDIO

RSSI

THD+NOISE

NOISE

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DIO

RF INPUT (dBm) (50Ω)

0.7µV 7µV 70µV 0.7mV 7mV 70mV

SR00830

Figure 11. VHF or UHF 2nd Conversion (Narrow Band)

As mentioned, a single layout was used for each of the examples.The board artwork is shown in Figure 8. Special attention was givento: (1) Creating a maximum amount of ground plane with connectionof the component side and solder side ground at locations all overthe board; (2) careful attention was given to keeping a ground ringaround each of the gain stages. The objective was to provide a

shunt path to ground for any stray signal which might feed back toan input; (3) leads were kept short and relatively wide to minimizethe potential for them to radiate or pick up stray signals; finally (andvery important), (4) RF bypass was done as close as possible tosupply pins and inputs, with a good (10µF) tantalum capacitorcompleting the system bypass.

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9101112131416 15

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568 7

0.1µF

0.1µF

0.1µF

0.1µF

0.1µF

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0.1µF

0.1µF51Ω

10µF

0.06µH

+6V

LO INPUT

21.4MHzCRYSTAL

FILTER

SA602

RF INPUT

330pF

56pF

SA604A

MUTE +6V

RSSIAUDIO DATA

1nF 91k

15nF

150pF

0.82pF

180pF0.3µH

430

TANT .33µH

390pF

270pF

0.1µF

820

C34

220pF

15pF

47pF

.33µH

0.1µF

SR00831

Figure 12. SA602/604A Demonstration Circuit with RF Input of 90MHz and IF of 21.4MHz ±7.5kHz



1997 Aug 20 6–8

5.0

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AUDIO

RSSI

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NOISE

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• 90MHz RF input • 21.4MHz IF• 68.6MHz LO • 15kHz IF BW (odBm) • 7kHz deviation SR00832

Figure 13. UHF Second Conversion (Narrow Band) or VHF Single Conversion (Narrow Band)

EXAMPLE: 45MHZ TO 10.7MHZ NARROWBANDAs a first example, consider conversion from 45MHz to 10.7MHz.There are commercially available filters for both frequencies so thisis a realistic combination for a second IF in a UHF receiver. Thiscircuit can also be applied to cordless telephone or short rangecommunications at 46 or 49MHz. The circuit is shown in Figure 9.

The 10.7MHz filter chosen is a type commonly available for 25kHzchannel spacing. It has a 3dB bandwidth of 15kHz and a terminationrequirement of 3kΩ/2pF. To present 3kΩ to the input side of the filter,a 1.5kΩ resistor was used between the SA602 output (which has a1.5kΩ impedance) and the filter. Layout capacitance was closeenough to 2pF that no adjustment was necessary. Thisseries-resistance approach introduces an insertion loss whichdegrades the sensitivity, but it has the benefit of simplicity.

The secondary side of the crystal filter is terminated with a 10.7MHztuned tank. The capacitor of the tank is tapped to create atransformer with the ratio for 3k:330. With the addition of the 430Ωresistor in parallel with the SA604A 1.6kΩ internal input resistor, thecorrect component of resistive termination is presented to the crystalfilter. The inductor of the tuned load is adjusted off resonanceenough to provide the 2pF capacitance needed. (Actual means ofadjustment was for best audio during alignment).

If appropriate or necessary for sensitivity, the same type of tunedtermination used for the secondary side of the crystal filter can alsobe used between the SA602 and the filter. If this is desired, thecapacitors should be ratioed for 1.5k:3k. Alignment is more complexwith tuned termination on both sides of the filter. This approach isdemonstrated in the fourth example.

A ceramic filter is used between the first and second limiters. It isdirectly connected between the output of the first limiter and the

input of the second limiter. Ceramic filters act much like ceramiccapacitors, so direct connection between two circuit nodes withdifferent DC levels is acceptable. At the input to the second limiter,the impedance is again reduced by the addition of a 430Ω externalresistor in parallel with the internal 1.6kΩ input load resistor. Thispresents the 330Ω termination to the ceramic filter which themanufacturers recommend.

On the input side of the ceramic filter, no attempt was made tocreate a match. The output impedance of the first limiter is nominally1kΩ. Crystal filters are tremendously sensitive to correct match.Ceramic filters are relatively forgiving. A review of themanufacturers’ data shows that the attenuation factor in thepassband is affected with improper match, but the degree of changeis small and the passband stays centered. Since the principalselectivity for this application is from the crystal filter at the input ofthe first limiter, the interstage ceramic filter only has to suppresswideband noise. The first filter’s passband is right in the center ofthe ceramic filter passband. (The crystal filter passband is less than10% of the ceramic filter passband). This passband relationship isillustrated in Figure 10.

After the second limiter, demodulation is accomplished in thequadrature detector. Quadrature criteria is not the topic of this paper,but it is noteworthy that the choice of loaded Q will affectperformance. The SA604A is specified at 455kHz using aquadrature capacitor of 10pF and a tuning capacitor of 180pF.(180pF gives a loaded Q of 20 at 455kHz). A careful look at thequadrature equations (Ref 3.) suggests that at 10.7MHz a value ofabout 1pF should be substituted for the 10pF at 455kHz.

The performance of this circuit is presented in Figure 11. The –12dBSINAD (ratio of Signal to Noise And Distortion) was achieved with a0.6µV input.



1997 Aug 20 6–9

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9101112131416 15

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568 7

0.1µF

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0.1µF

0.1µF

0.1µF

51Ω

10µF

0.28µH

+6V

LO INPUT10.7MHz

CERAMICFILTER

10.7MHzCERAMIC

FILTER

SA602

RF INPUT

220pF

47pF

430

SA604A

MUTE +6V RSSI AUDIO DATA

1nF 91k

15nF

150pF

1pF 180pF1.1µH

430

SR00833

Figure 14. SA602/604A Demonstration Circuit with RF Input of ~100MHz and IF of 10.7MHz ±140kHz

EXAMPLE: 90MHZ TO 21.4MHZ NARROWBANDThis second example, like the first, used two frequencies whichcould represent the intermediate frequencies of a UHF receiver. Thiscircuit can also be applied to VHF single conversion receivers if thesensitivity is appropriate. The circuit is shown in Figure 12.

Most of the fundamentals are the same as explained in the firstexample. The 21.4MHz crystal filter has a 1.5kΩ/2pF terminationrequirement so direct connection to the output of the SA602 ispossible. With strays there is probably more than 2pF in this circuit,

but the performance is good nonetheless. The output of the crystalfilter is terminated with a tuned impedance-step-down transformeras in the previous example. Interstage filtering is accomplished witha 1kΩ:330 step-down ratio. (Remember, the output of the first limiteris 1kΩ and a 430Ω resistor has been added to make the secondlimiter input 330Ω). A DC blocking capacitor is needed from theoutput of the first limiter. The board was not laid out for an interstagetransformer, so an “XACTO” knife was used to make some minormods. Figure 13 shows the performance. The +12dB SINAD waswith 1.6µV input.

• 94.7MHz RF input • 10.7MHz IF• 84MHz LO • 280kHz IF BW (odBm) • 75kHz deviation

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AUDIO

RSSI

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NOISE

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SR00834

Figure 15. FM Broadcast Receiver (Wide Band)



1997 Aug 20 6–10

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568 7

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0.1µF

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0.1µF

51Ω

10µF

0.28µH

+6V

LO INPUT

10.7MHzCERAMIC

FILTER

10.7MHzCERAMIC

FILTER

SA602

RF INPUT

220pF

47pF

430

SA604A

MUTE +6V

RSSIAUDIO DATA

1nF 91k

15nF

150pF

1pF 180pF1.1µH

430

10M15A

SR00835

Figure 16. SA602/604A Demonstration Circuit with RF Input of 152.2MHz and IF of 10.7MHz ±7.5kHz

• 152.2MHz RF input • 10.7MHz IF• 141.5MHz LO • 15kHz IF BW (odBm) • 7kHz deviation

3.50

3.46

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3.34

3.30-10 0 10 20 30 40 50 60 70 80

OU

TP

UT

BIA

S V

OLT

AG

E (

V)

AMBIENT TEMPERATURE (°C)

VCC = 5.0V

PIN 14

PIN 12

SR00836

Figure 17. VHF Single Conversion (Narrow Band)

EXAMPLE: 100MHZ TO 10.7MHZ WIDEBANDThis example represents three possible applications: (1) low cost,sensitive FM broadcast receivers, (2) SCA (SubsidiaryCommunications Authorization) receivers and (3) data receivers.The circuit schematic is shown in Figure 14. While this example hasthe greatest diversity of application, it is also the simplest. Two10.7MHz ceramic filters were used. The first was directly connectedto the output of the SA602. The second was directly connected to

the output of the first IF limiter. The secondary sides of both filterswere terminated with 330Ω as in the two previous examples. Whilethe filter bandpass skew of this simple single conversion receivermight not be tolerable in some applications, to a first order theresults are excellent. (Please note that sensitivity is measured at+20dB in this wideband example.) Performance is illustrated inFigure 15. +20dB SINAD was measured with 1.8µV input.

EXAMPLE: 152.2MHZ TO 10.7MHZ NARROWBANDIn this example (see Figure 16) a simple, effective, and relativelysensitive single conversion VHF receiver has been implemented. Allof the circuit philosophy has been described in previous examples.In this circuit, tuned-transformed termination was used on the inputand output sides of the crystal filter. Performance is shown in Figure17. The +12dB SINAD sensitivity was 0.9µV.

OSCILLATORSThe SA602 contains an oscillator transistor which can be used tofrequencies greater than 200MHz. Some of the possibleconfigurations are shown in Figures 18 and 19.

L/CWhen using a synthesizer, the LO must be externally buffered.Perhaps the simplest approach is an emitter follower with the baseconnected to Pin 7 of the SA602. The use of a dual-gate MOSFETwill improve performance because it presents a fairly constantcapacitance at its gate and because it has very high reverseisolation.



1997 Aug 20 6–11

3

4

3

4

a. FundamentalColpitts Crystal

b. OvertoneColpitts Crystal

c. OvertoneButler Crystal

d. HartleyL/C Tank

e. ColpittsL/C Tank

63

4

6

7

63

4

4

3

SR00837

Figure 18. Oscillator Configurations

CRYSTALWith both of the Colpitts crystal configurations, the load capacitancemust be specified. In the overtone mode, this can become asensitive issue since the capacitance from the emitter to ground isactually the equivalent capacitive reactance of the harmonicselection network. The Butler oscillator uses an overtone crystalspecified for series mode operation (no parallel capacitance). It mayrequire an extra inductor (Lo) to null out Co of the crystal, butotherwise is fairly easy to implement (see references).

The oscillator transistor is biased with only 220µA. In order to assureoscillation in some configurations, it may be necessary to increasetransconductance with an external resistor from the emitter toground. 10kΩ to 20kΩ are acceptable values. Too small a resistancecan upset DC bias (see references).

DATA DEMODULATIONIt is possible to change any of the examples from an audio receiverto an amplitude shift keyed (ASK) or frequency shift keyed (FSK)receiver or both with the addition of an external op amp(s) orcomparator(s). A simple example is shown in Figure 20. ASK

decoding is accomplished by applying a comparator across thereceived signal strength indicator (RSSI). The RSSI will track IFlevel down to below the limits of the demodulator (–120dBm RFinput in most of the examples). When an in-band signal is above thecomparator threshold, the output logic level will change.

FSK demodulation takes advantage of the two audio outputs of theSA604A. Each is a PNP current source type output with 180° phaserelationship. With no signal present, the quad tank tuned for thecenter of the IF passband, and both outputs loaded with the samevalue of capacitance, if a signal is received which is frequencyshifted from the

IF passband, and both outputs loaded with the same value ofcapacitance, if a signal is received which is frequency shifted fromthe IF center, one output voltage will increase and the other willdecrease by a corresponding absolute value. Thus, if a comparatoris differentially connected across the two outputs, a frequency shiftin one direction will drive the comparator output to one supply rail,and a frequency shift in the opposite direction will cause thecomparator output to swing to the opposite rail. Using this technique,and L/C filtering for a wide IF bandwidth, NRZ data at rates greaterthan 4Mb have been processed with the new SA605.



1997 Aug 20 6–12

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568 76.8µF

0.09µH

100nF

SA602

10nF

0.01

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0.001

MV2105OR EQUIV

0.01

FROMSYNTH LOOP

FILTER

18k

9pF

2pF

11pF 0.001

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18k

MV2105OR EQUIV

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0.01

0.001

0.8

2k

100k

100k

47k 0.010.01

330Ω

0.08µH

**12pF*10.7MHz

IFK&L 38780OR EQUIV

11µH 11µH

3SK126 OR EQUIVALENT

TOPRESCALER

VCC

VCC

*PERMITS IMPEDANCE MATCH OF NE602 OUTPUT OF 1.8k/8pF TO 3.0k FILTER IMPEDANCE**CHOOSE FOR IMPEDANCE MATCH TO SR00838

Figure 19. Typical Varactor Tuned Application

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0.1µF

0.1µF

0.1µF

0.1µF

51Ω

10µF+6V

LO INPUT10.7MHz

CERAMICFILTER

10.7MHzCERAMIC

FILTER

SA602

RF INPUT

430

SA604A

MUTE

FSK OUTPUT

91k

1pF 68pF3.3µH

430

+6V ASK OUTPUT

10

100nF

TANT

SR00839

Figure 20. Basic SA602/604A Data Receiver



1997 Aug 20 6–13

SUMMARYThe SA602, SA604A and SA605 provide the RF system designerwith the opportunity for excellent receiver or IF system sensitivitywith very simple circuitry. IFs at 455kHz, 10.7MHz and 21.4MHzwith 75 to 90dB gain are possible without special shielding. Theflexible configuration of the built-in oscillator of the SA602/605 addto ease of implementation. Either data or audio can be recoveredfrom the SA604A/605 outputs.

REFERENCES1) Anderson, D.: “Low Power ICs for RF Data Communications”,Machine Design , pp 126-128, July 23, 1987.

2) Krauss, Raab, Bastian: Solid State Radio Engineering , p. 311,Wiley, 1980.

3) Matthys, R.: “Survey of VHF Crystal Oscillator Circuits,” RFTechnology Expo Proceedings, pp 371-382, February, 1987.

4) Philips Semiconductors: “SA604A High Performance Low PowerFM IF System”, Linear Data and Applications Manual, PhilipsSemiconductors, 1987.

5) Philips Semiconductors; “SA602 Double Balanced Mixer andOscillator”, Linear Data and Applications Manual, PhilipsSemiconductors, 1985.

6) Philips Semiconductors: “AN1982–Applying the Oscillator of theSA602 in Low Power Mixer Applications”, Linear Data andApplications Manual, Philips Semiconductors, 1985.

NE602 App Note

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