My Thesis

Misr University for Science & Technology Faculty of Engineering

Department of Electronics and Communication Engineering

Simulation and Implementation for Several PAPR Reduction Techniques in OFDM using USRP with

LABVIEW

Authors: 1. Mohamed Rabie Ragab 40115 2. Ahmed Fathi Gouda 40023 3. Amal Hamed Abdelmonem 43034 4. Yasmin Hamdy Baioumy 40424 5. Ahmed Mahmoud Anwar 40121

Associated Professor:

Dr. MAMDOUH GOUDA

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Acknowledgements

First of all I would like to thank my supervisor Dr.Mamdouh Gouda who believed in me and was continuously supporting me in every problem that occurred during the writing of this report. I especially thank him for his encouragement and his accurate comments which were of critical importance, during this work. Our cooperation was truly an inspiring experience, I feel grateful. Very special thanks to Eng.Mohamed Hussien, doctoral student of communication engineering, for his eagerness and great assistance in order to accomplish the writing of this report. I would also want to thank the members of the examining committee, assistant professors as well as the audience during this report’s presentation. Special thanks goes to all my friends for their advice, their continuous support and their tolerance during the writing of this report, without your moral support and love I could not have made it until the end. I would like to dedicate this work to my beloved parents and families who always believed in us, always were there and will always be whenever we need them. we owe you everything.

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List of Figure Fig. 2.1 Discrete Fourier Transform ............................................................................................................ 10 Fig. 2.2 ISI Due to channel filtering .............................................................................................................. 12 Fig. 2.3 Overlap of neighboring subcarriers, Inter Carrier Interference (ICI) ............................. 13 Fig. 2.4 Multipath reception leads to location and frequency-selective fading ........................... 13 Fig. 2.5 Echo Delay .............................................................................................................................................. 14 Fig. 2.6 Inter Symbol Interference (ISI) ...................................................................................................... 14 Fig. 2.7 Multi Carrier .......................................................................................................................................... 15 Fig. 2.8 Channel Impulse Response .............................................................................................................. 16 Fig. 2.9 Flat Channel ........................................................................................................................................... 16 Fig. 2.10 Orthogonal subcarriers .................................................................................................................. 17 Fig. 2.11 Block diagram of Orthogonal subcarriers ............................................................................... 18 Fig 2.12 Subcarrier with Guard Band .......................................................................................................... 19 Fig. 2.13 Difference between spectra of FDM and OFDM ..................................................................... 19 Fig. 2.14 OFDM Implementation.................................................................................................................... 20 Fig. 2.15 Discrete Fourier Transform (DFT) ............................................................................................. 21 Fig. 2.16 OFDM transmitter ............................................................................................................................. 22 Fig. 2.17 OFDM receiver ................................................................................................................................... 22 Fig. 2.18 Guard Period ...................................................................................................................................... 23 Fig. 3.1 Power samples of one symbol OFDM signal .............................................................................. 28 Fig. 3.2 Block diagram of OFDM system...................................................................................................... 29 Fig. 3.3 Histogram of Real part of OFDM signal amplitude ................................................................. 31 Fig. 3.4 Histogram of Imaginary part of OFDM signal amplitude ...................................................... 31 Fig. 3.5 OFDM signal magnitude .................................................................................................................... 32

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Fig. 3.6 OFDM transmitter basic scheme ..................................................................................................... 32 Fig. 3.7 Linear and Non-Linear Power Amplifier .................................................................................... 33 Fig. 3.8 256-QAM constellations: (a) regular and (b) modified mapping to reduce ...................... 35 Fig. 3.9 Types of PAPR Reduction Techniques ......................................................................................... 38 Fig. 4.1 SDR Sender and Receiver module diagram ............................................................................... 41 Fig. 4.2 USRP Connection Diagram ............................................................................................................... 42 Fig. 4.3 Detailed view of Front Panel of NI USRP-2920 ......................................................................... 46 Fig. 4.4 NI USRP-2920 Front Panel ............................................................................................................... 46 Fig. 4.5 MIMO Expansion cable ...................................................................................................................... 47 Fig. 4.6 USRP Antenna ....................................................................................................................................... 47 Fig. 4.7 Gigabit Ethernet Cable ....................................................................................................................... 48 Fig. 4.8 Power Supply ........................................................................................................................................ 48 Fig. 4.9 The USRP Configuration .................................................................................................................... 49 Fig. 4.10 Detailed block diagram of NI-USRP 2920 ................................................................................. 50 Fig. 4.11 Power connections to the USRP kit ............................................................................................ 51 Fig. 4.12 Ethernet cable and antenna connections to the USRP ......................................................... 52 Fig. 4.13 Connections of the NI-USRP kit to the host PC ........................................................................ 52 Fig. 4.14 Verifying the device in NI-USRP-Configuration Utility ........................................................ 53 Fig. 4.15 Changing the device IP of the device .......................................................................................... 54 Fig. 5.1 The block diagram of the Communication System .................................................................. 58 Fig. 5.2 The block diagram of OFDM Transceiver ................................................................................... 58 Fig. 5.3 The block diagram of OFDM transmitter using Labview ...................................................... 59 Fig. 5.4 The block diagram of OFDM Receiver using Labview ............................................................ 59 Fig. 5.5 SLM General Block Diagram ............................................................................................................ 60 Fig. 5.6 SLM Block Diagram using Labview ............................................................................................... 61

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Fig. 5.7 Received 4-QAM Constellation for SLM ....................................................................................... 62 Fig. 5.8 Received 16-QAM Constellation for SLM ..................................................................................... 62 Fig. 5.9 Received 64-QAM Constellation for SLM ..................................................................................... 63 Fig. 5.10 Received 256-QAM Constellation for SLM ............................................................................... 63 Fig. 5.11 CCDF for SLM with different types of Modulation ................................................................. 64 Fig. 5.12 PTS General Block Diagram .......................................................................................................... 65 Fig. 5.13 PTS Block Diagram using Labview ............................................................................................. 66 Fig. 5.14 Received 4-QAM Constellation for PTS (V=4 & V=16) respectively ................................ 67 Fig. 5.15 Received 4-QAM Constellation for PTS (V=32 & V=64) respectively ............................. 67 Fig. 5.16 CCDF for PTS 4-QAM with different number of Sub-blocks (V) ........................................ 68 Fig. 5.17 Received 16-QAM Constellation for PTS (V=4 & V=16) respectively ............................. 69 Fig. 5.18 Received 16-QAM Constellation for PTS (V=32 & V=64) respectively .......................... 69 Fig. 5.19 CCDF for PTS 16-QAM with different Sub-blocks (V) ........................................................... 70 Fig. 5.20 Received 64-QAM Constellation for PTS (V=4 & V=16) respectively ............................. 71 Fig. 5.21 Received 64-QAM Constellation for PTS (V=32 & V=64) respectively .......................... 71 Fig. 5.22 CCDF for PTS 64-QAM with different Sub-blocks (V) ........................................................... 72 Fig. 5.23 Active Constellation Extension (a) for QPSK (b) for 16 QAM ............................................ 73 Fig. 5.24 ACE Block Diagram using Labview ............................................................................................. 73 Fig. 5.25 Received 4-QAM Constellation for ACE ..................................................................................... 74 Fig. 5.26 Received 16-QAM Constellation for ACE................................................................................... 74 Fig. 5.27 Received 64-QAM Constellation for ACE................................................................................... 75 Fig. 5.28 Received 256-QAM Constellation for ACE ................................................................................ 75 Fig. 5.29 CCDF for ACE with different types of Modulation.................................................................. 76 Fig. 5.30 TI General Block Diagram .............................................................................................................. 77 Fig. 5.31 TI Block Diagram using Labview ................................................................................................. 77 Fig. 5.32 Received 4-QAM Constellation for TI ......................................................................................... 78 Fig. 5.33 Received 16-QAM Constellation for TI ...................................................................................... 78

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Fig. 5.34 Received 64-QAM Constellation for TI ...................................................................................... 79 Fig. 5.35 Received 256-QAM Constellation for TI ................................................................................... 79 Fig. 5.36 CCDF for TI with different types of Modulation ..................................................................... 80 Fig. 5.37 Interleaving General Block Diagram ......................................................................................... 81 Fig. 5.38 Interleaving Block Diagram using Labview ............................................................................ 81 Fig. 5.39 Received 4-QAM Constellation for Interleaving .................................................................... 82 Fig. 5.40 Received 16-QAM Constellation for Interleaving .................................................................. 82 Fig. 5.41 Received 64-QAM Constellation for Interleaving .................................................................. 83 Fig. 5.42 CCDF for Interleaving with different types of Modulation ................................................. 83 Fig. 5.43 ACF General Block Diagram .......................................................................................................... 85 Fig. 5.44 ACF Block Diagram using Labview ............................................................................................. 85 Fig. 5.45 Received 4-QAM Constellation for ACF ..................................................................................... 86 Fig. 5.46 Received 16-QAM Constellation for ACF ................................................................................... 86 Fig. 5.47 CCDF for ACF with different types of Modulation .................................................................. 87 Fig. 5.48 Hardware Component ..................................................................................................................... 88 Fig. 5.49 Received signal for various V=4,16,32,64 using 4-QAM ...................................................... 90 Fig. 5.50 Received signal in time domain for various V = 4,16,32,64 using 4-QAM ................... 91 Fig. 5.51 CCDF for PTS 4-QAM with different Sub-blocks (V) .............................................................. 92 Fig. 5.52 Received signal for various V=4,16,32,64 using 16-QAM ................................................... 93 Fig. 5.53 Received signal in time domain for various V=4,16,32,64 using 16-QAM ................... 94 Fig. 5.54 CCDF for PTS 16-QAM with different Sub-blocks (V) ........................................................... 95 Fig. 5.55 Received signal for various V=4,16,32,64 using 64-QAM ...................................................... 96 Fig. 5.56 Received signal in time domain for various V=4,16,32,64 using 64-QAM .................. 97 Fig. 5.57 CCDF for PTS 64-QAM with different Sub-blocks (V) ........................................................... 98

VII

List of Table Table 1 Summary of all Generation ................................................................................................................................ 7 Table 2 Comparison between FDM and OFDM ....................................................................................................... 20 Table 3 PAPR for picked modulation formats ........................................................................................................ 35 Table 4 PAPR Reduction Techniques .......................................................................................................................... 37 Table 5 The Features of NI-USRP-2920 TX & RX ................................................................................................. 43 Table 6 Connectors of NI USRP-2920 .......................................................................................................................... 44 Table 7 NI USRP-2920 Module LEDs .......................................................................................................................... 45 Table 8 The System Parameters used for Simulation ............................................................................................ 57 Table 9 CCDF values for SLM with different types of Modulation ................................................................ 64 Table 10 CCDF values for PTS 4-QAM Modulation with different number of sub-blocks(V)........... 68 Table 11 CCDF values for PTS 16-QAM Modulation with different number of sub-blocks(V) ........ 70 Table 12 CCDF values for PTS 16-QAM Modulation with different number of sub-blocks(V) ........ 72 Table 13 CCDF values for ACE with different types of Modulation .............................................................. 76 Table 14 CCDF values for TI with different types of Modulation ................................................................... 80 Table 15 CCDF values for Interleaving with different types of Modulation ............................................... 84 Table 16 CCDF values for ACF with different types of Modulation .............................................................. 87 Table 17 The System Parameters used for Implementation ............................................................................... 89 Table 18 CCDF values for PTS 4-QAM Modulation with different number of sub-blocks(V)........... 92 Table 19 CCDF values for PTS 16-QAM Modulation with different number of sub-blocks(V) ........ 95 Table 20 CCDF values for PTS 64-QAM Modulation with different number of sub-blocks(V) ........ 98

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Abbreviations OFDM : Orthogonal Frequency-Division Multiplexing OFDMA : Orthogonal Frequency-Division Multiple Access FDM : Frequency-Division Multiplexing FDMA : Frequency-Division Multiple Access TDM : Time-Division Multiplexing TDMA : Time-Division Multiple Access CDMA : Code-Division Multiple Access COFDM : Coded Orthogonal Frequency-Division Multiplexing PAPR : Peak to Average Power Ratio ISI : Inter Symbol Interference ICI : Inter Carrier Interference 1G : First Generation 2G : Second Generation 3G : Third Generation 4G : Fourth Generation 5G : Fifth Generation DUC : Digital up Converter HDSL : High-bit-rate Digital Subscriber Line ADSL : Asymmetric Digital Subscriber Line VDSL : Very-high-speed Digital Subscriber Line DAB : Digital Audio Broadcasting DVB : Digital Video Broadcasting HDTV : High-Definition Television

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IEEE : Institute of Electrical and Electronics Engineers UMTS : Universal Mobile Telecommunication System PDS : Pacific Digital Cellular D-AMPS : Digital-Advanced Mobile Phone Services NMT : Nordic Mobile Telephony TACS : Total Access Communication System AMPS : Advanced Mobile Phone Service AWGN : Additive White Gaussian Noise CCDF : Complementary Cumulative Distribution Function PDF : Probability Density Function FFT : Fast Fourier Transform IFFT : Inverse Fast Fourier Transform S/P : Serial to Parallel P/S : Parallel to Serial DFT : Discrete Fourier Transform DAC : Digital to Analog Converter ADC : Analog to Digital Converter Tx : Transmitter Rx : Receiver RF : Radio Frequency HPA : High Power Amplifier BER : Bit Error Rate FM : Frequency Modulation CP : Cyclic Prefix QAM : Quadrature Amplitude Modulation QPSK : Quadrature Phase-Shift Keying

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M-PSK : Multiple Phase-Shift Keying SDR : Software Defined Radio MIMO : Multi Input Multi Output NI : National Instruments USRP : Universal Software Radio Peripheral SLM : Selected Level Mapping PTS : Partial Transmit Sequences TR : Tone Reservation TI : Tone Injection ACE : Active Constellation Extension AI : Adaptive Interleaving ACF : Amplitude Clipping and Filtering

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Contents Chapter One: Introduction to wireless Communication 1.1 Aim of this Thesis .................................................................................................................................................... 2 1.2 Evolution of wireless communications ............................................................................................................ 3

1.2.1 First generation (1G) cellular systems ............................................................................................. 3 1.2.2 Second generation (2G) cellular systems ......................................................................................... 3 1.2.3 Third generation (3G) cellular systems ............................................................................................ 4 1.2.4 Fourth generation (4G) wireless broadband systems ....................................................................... 5 1.2.5 Next generation technology ............................................................................................................. 6

1.3 Summary of all generations ................................................................................................................................. 7 Chapter Two: Orthogonal Frequency Division Multiplexing 2.1 OFDM Historical Overview ............................................................................................................................. 11 2.2 Important Definitions.......................................................................................................................................... 11

2.2.1 Inter-Symbol Interference (ISI) .................................................................................................... 12 2.2.2 Inter-Carrier Interference (ICI) ..................................................................................................... 12

2.3 OFDM as a Multicarrier Transmission Technique ................................................................................. 13 2.4 OFDM Concept ..................................................................................................................................................... 15

2.4.1 Analysis.......................................................................................................................................... 16 2.4.1.1 Time Domain Analysis ........................................................................................................... 16 2.4.1.2 Frequency Domain Analysis ................................................................................................... 16 2.4.1.3 Conclusion from both Domains Analysis ............................................................................... 17

2.5 Orthogonality of OFDM .................................................................................................................................... 17 2.6 Comparing FDM to OFDM ............................................................................................................................. 19

2.6.1Frequency Division Multiplexing (FDM) ....................................................................................... 19 2.6.2 Orthogonal Frequency Division Multiplexing (OFDM) ................................................................ 19

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2.7 OFDM Implementation ...................................................................................................................................... 20 2.7.1 Implementation using “FFT/IFFT” ................................................................................................ 21

2.8 OFDM Transmitter .............................................................................................................................................. 21 2.9 OFDM Receiver .................................................................................................................................................... 22 2.10 Cyclic Prefix ........................................................................................................................................................ 23 2.11 Strengths and drawbacks of OFDM ........................................................................................................... 24

2.11.1 OFDM advantages ....................................................................................................................... 24 2.11.2 OFDM disadvantages ................................................................................................................... 24

Chapter Three: The Peak-to-Average power Ratio 3.1 Introduction to Peak to Average Power Ratio (PAPR) ......................................................................... 27 3.2 Definitions of PAPR ............................................................................................................................................ 27

3.2.1 (PAPR) of a continuous time signal ............................................................................................... 28 3.2.2 (PAPR) of a discrete time signal .................................................................................................... 29

3.3 PAPR of OFDM signal ...................................................................................................................................... 29 3.3.1 Calculation of PAPR ........................................................................................................................ 30 3.3.2 Complementary Cumulative Distribution Function ....................................................................... 30

3.4 Identification of the Problem ........................................................................................................................... 32 3.5 PAPR Effect ........................................................................................................................................................... 34 3.6 Factors influencing the PAPR ........................................................................................................................ 34

3.6.1 The number of sub carriers ............................................................................................................ 34 3.6.2 The order of Modulation ................................................................................................................ 34 3.6.3 Constellation shape ........................................................................................................................ 35 3.6.4 Pulse Shaping ................................................................................................................................. 35

3.7 Existing Approaches for PAPR Reduction ................................................................................................. 36 3.8 PAPR Reduction Techniques ........................................................................................................................... 37

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Chapter Four: Getting Start with Software Defined Radio 4.1 Software Defined Radio ..................................................................................................................................... 41 4.2 The Hardware of USRP and Software of Labview .................................................................................. 42

4.2.1 Hardware Specifications ................................................................................................................ 43 4.2.2 Connectors of NI USRP-2920........................................................................................................ 44 4.2.3 NI USRP-2920 Module LED ......................................................................................................... 45 4.2.4 NI USRP-2920 Front Panel ........................................................................................................... 46

4.3 USRP Parameters configuration .................................................................................................................... 49 4.4 System Implementation....................................................................................................................................... 50

4.4.1 NI-USRP 2920 Block diagram ...................................................................................................... 50 4.4.2 Interfacing of the Host PC with the Kit ......................................................................................... 51

Chapter Five: Simulation and Implementation Results 5.1 The overall system design specifications .................................................................................................... 57 5.2 OFDM Physical Layer ....................................................................................................................................... 57 5.3 The Simulation Results ....................................................................................................................................... 60

5.3.1 Selected Level Mapping (SLM) ..................................................................................................... 60 5.3.1.1 Block Diagram ........................................................................................................................ 60 5.3.1.2 SLM Equations ....................................................................................................................... 61 5.3.1.3 SLM Results ............................................................................................................................ 62

5.3.2 Partial Transmit Sequences (PTS) ................................................................................................. 65 5.3.2.1 Block Diagram ........................................................................................................................ 65 5.3.2.2 PTS Equations ......................................................................................................................... 66 5.3.2.3 PTS Results ............................................................................................................................. 67

5.3.3 Active Constellation Extension( ACE ) ......................................................................................... 73 5.3.3.1 Block Diagram ........................................................................................................................ 73

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5.3.3.2 ACE Results ............................................................................................................................ 74 5.3.4 Tone Injection (TI) ......................................................................................................................... 77

5.3.4.1 Block Diagram ........................................................................................................................ 77 5.3.4.2 TI Results ................................................................................................................................. 78

5.3.5 Interleaving .................................................................................................................................... 81 5.3.5.1 Block Diagram ........................................................................................................................ 81 5.3.5.2 Interleaving Results................................................................................................................. 82

5.3.6 Amplitude Clipping and Filtering (ACF) ....................................................................................... 84 5.3.6.1 Block Diagram ........................................................................................................................ 85 5.3.6.2 ACF Equations ........................................................................................................................ 85 5.3.6.3 ACF Results ............................................................................................................................ 86

5.4 The Implementation Results for PTS technique using USRP .............................................................. 88 5.4.1 The System Parameters Used for Implementation ........................................................................ 89 5.4.2 Implementation Results of PTS using USRP ................................................................................. 90

Conclusion ............................................................................................................................................100 Future Work .........................................................................................................................................101 DISSEMINATION ..................................................................................................................................102 Reference ..............................................................................................................................................104

XV

PREFACE Orthogonal frequency-division multiplexing (OFDM) is an orthogonal waveform technique for encoding digital data on multiple carrier frequencies. OFDM has developed into a popular scheme for wide band digital communication, used in applications such as wireless networks and 4G mobile communications such as LTE, WIMAX and Wi-Fi. It still exists with some of the drawbacks, out of which, high peak to average power ratio (PAPR) gives rise to non-linear distortion, inter-symbol interference and out-of-band radiation. There has been various ways developed and implemented to reduce PAPR . we will describe all of these techniques to reduce PAPR such as : Partial Transmit Sequence (PTS) , Selective Level Mapping ( SLM ) , Interleaving , Tone Injection ( TI ) and Active Constellation Extension( ACE ) which comes under Signal scrambling techniques . And Amplitude Clipping and Filter (ACF) which comes under signal distortion technique. In this thesis, we simulate all these techniques using LABVIEW software then implement Partial Transmit Sequence (PTS) technique on Universal Software Radio Peripheral (USRP). Organization of this thesis

Chapter One: Overview of wireless communication. Chapter Two: Fundamentals of OFDM Systems. Chapter Three: Peak to Average Power Ratio (PAPR). Chapter Four: OFDM System Implementation Using Software Defined Radio. Chapter Five: Simulation and Implementation Results.

Chapter One

Introduction to wireless

Communication

Chapter One Introduction of Wireless Communication

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Chapter 1 Introduction of wireless communication

1.1 Aim of this Thesis The pervasive use of wireless communications is more and more conditioning lifestyle and working habits in many developed countries. Examples of this trend are the ever increasing number of users that demand Internet connection when they are traveling, the use of cellular phones to check bank accounts and make remote payments, or the possibility of sharing moments in our lives with distant friends by sending them images and video clips. In the last few years, the proliferation of laptop computers has led to the development of wireless local area networks (WLANs), which are rapidly supplanting wired systems in many residential homes and business offices. More recently, wireless metropolitan area networks (WMANs) have been standardized to provide rural locations with broadband Internet access without the costly infrastructure required for deploying cables. A new generation of wireless systems wherein multimedia services like speech, audio, video and data will converge into a common and integrated platform is currently under study and is expected to become a reality in the near future.[1] The promise of portability is clearly one of the main advantages of the wireless technology over cabled networks. Nevertheless, the design of a wireless communication system that may reliably support emerging multimedia applications must deal with several technological challenges that have motivated an intense research in the field. One of this challenge is the harsh nature of the communication channel. In wireless applications, the radiated electromagnetic wave arrives at the receiving antenna after being scattered, reflected and diffracted by surrounding objects. As a result, the receiver observes the superposition of several differently attenuated and delayed copies of the transmitted signal. The constructive or destructive combination of these copies induces large fluctuations in the received signal strength with a corresponding degradation of the link quality. In addition, the characteristics of the channel may randomly change in time due to unpredictable variations of the propagation environment or as a consequence of the relative motion between the transmitter and receiver. A second challenge is represented by the limited amount of available radio spectrum, which is a very scarce and expensive resource. It suffices to recall that European telecommunication companies spent over 100 billion dollars to get licenses for third-generation cellular services. To obtain a reasonable return from this investment, the purchased spectrum must be used as efficiently as possible. A further impairment of wireless transmissions is the relatively high level of interference arising from channel reuse. Although advanced signal processing techniques based on multiuser detection have recently been devised for interference mitigation, it is a fact that mobile wireless communications will never be able to approach the high degree of stability, security and reliability afforded by cabled systems. Nevertheless, it seems that customers are ready to pay the price of a lower data throughput and worse link quality in order to get rid of wires.[1]


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1.2 Evolution of wireless communications Before proceeding to a systematic study of OFDM and OFDMA, we think it useful to review some basic applications of such schemes and highlight the historical reasons that led to their development. The current section is devoted to this purpose, and illustrates the evolution of wireless communication systems starting from the theoretical works of Maxwell in the nineteenth century till the most recent studies on broadband wireless networks. Some historical notes on multicarrier transmissions are next provided in the last section of this introductory chapter. 1.2.1 First generation (1G) cellular systems Despite its theoretical relevance, the cellular concept was not widely adopted during the 1960s and 1970s. To make an example, in 1976 the Bell Mobile Phone had only 543 paying customers in the New York City area, and mobile communications were mainly supported by heavy terminals mounted on cars. Although the first patent describing a portable mobile telephone was granted to Motorola in 1975 mobile cellular systems were not introduced for commercial use until the early 1980s, when the so-called first generation (1G) of cellular networks were deployed in most developed countries. The common feature of 1G systems was the adoption of an analog transmission technology. Frequency modulation (FM) was used for speech transmission over the 800-900 MHz band and frequency division multiple access (FDMA) was adopted to separate users' signals in the frequency domain. In practice, a fraction of the available spectrum (sub-channel) was exclusively allocated to a given user during the call set-up and retained for the entire call.[1] In the early 1980s, 1G cellular networks experienced a rapid growth in Europe, particularly in Scandinavia where the Nordic Mobile Telephony(NMT) appeared in 1981, and in United Kingdom where the Total Access Communication System (TACS) started service in 1985. The Advanced Mobile Phone Service (AMPS) was deployed in Japan in 1979, while in the United States it appeared later in 1983. These analog systems created a critical mass of customers. Their main limitations were the large dimensions of cell phones and the reduced traffic capacity due to a highly inefficient use of the radio spectrum. At the end of the 1980s, progress in semiconductor technology and device miniaturization allowed the production of small and light-weight hand held phones with good speech quality and acceptable battery lifetime. This marked the beginning of the wireless cellular revolution that took almost everyone by surprise since in the meantime many important companies had stopped business activities in cellular communications, convinced that mobile telephony would have been limited to rich people and would have never attracted a significant number of subscribers. 1.2.2 Second generation (2G) cellular systems The limitations of analog radio technology in terms of traffic capacity became evident in the late 1980s, when 1G systems saturated in many big cities due to the rapid growth of the cellular market. Network operators realized that time was ripe for a second generation (2G) of cellular systems that would have marked the transition from analog to digital radio technology.


4

This transition was not only motivated by the need for higher network capacity, but also by the lower cost and improved performance of digital hardware as compared to analog circuitry. Driven by the success of NMT, in 1982 the Conference of European Posts and Telecommunications (CEPT) formed the Group Special Mobile (GSM) in order to develop a pan-European standard for mobile cellular radio services with good speech quality, high spectral efficiency and the ability for secure communications. The specifications of the new standard were approved in 1989 while its commercial use began in 1993. Unlike 1G systems, the GSM was developed as a digital standard where users' analog signals are converted into sequences of bits and transmitted on a frame-by-frame basis. Within each frame, users transmit their bits only during specified time intervals (slots) that are exclusively assigned at the call setup according to a time-division multiple-access (TDMA) approach.[1] Actually, the GSM is based on a hybrid combination of FDMA and TDMA, where FDMA is employed to divide the available spectrum into 200 kHz wide sub-channels while TDMA is used to separate up to a maximum of eight users allocated over the same sub-channel. In Europe the operating frequency band is 900 MHz, even though in many big cities the 1800 MHz band is also being adopted to accommodate a larger number of users. Many modern European GSM phones operate in a "dual-band" mode by selecting either of the two recommended frequencies. In the United States, the 1900 MHz frequency band is reserved to the GSM service. In addition to circuit-switched applications like voice, the adoption of a digital technology enabled 2G cellular systems to offer low-rate data services including email and short messaging up to 14.4 kbps. The success of GSM was such that by June 2001 there were more than 500 million GSM subscribers all over the world while in 2004 the market penetration exceeded 80% in Western Europe. The reasons for this success can be found in the larger capacity and many more services that the new digital standard offered as compared to previous. 1G analog systems. Unfortunately, the explosive market of digital cell phones led to a proliferation of incompatible 2G standards that sometimes prevent the possibility of roaming among different countries. Examples of this proliferation are the Digital Advanced Mobile Phone Services (D-AMPS) which was introduced in the United States in 1991 and the Japanese Pacific Digital Cellular (PDS). The Interim Standard 95 (IS-95) became operative in the United States starting from 1995 and was the first commercial system to employ the code-division multiple-access (CDMA) technology as an air interface. 1.2.3 Third generation (3G) cellular systems At the end of the 1990s it became clear that GSM was not sufficient to indefinitely support the explosive number of users and the ever-increasing data rates requested by emerging multimedia services. There was the need for a new generation of cellular systems capable of supporting higher transmission rates with improved quality of service as compared to GSM. After long deliberations, two prominent standards emerged: the Japanese-European Universal Mobile Telecommunication System (UMTS) and the American CDMA-2000 [161]. Both systems operate around the 2 GHz frequency band and adopt a hybrid FDMA/CDMA approach.


5

In practice, groups of users are allocated over disjoint frequency sub-bands, with users sharing a common sub-band being distinguished by quasi-orthogonal spreading codes. The CDMA technology has several advantages over TDMA and FDMA, including higher spectral efficiency and increased flexibility in radio resource management. In practical applications, however, channel distortions may destroy orthogonality among users' codes, thereby resulting in multiple access interference (MAI). In the early 1990s, problems related to MAI mitigation spurred an intense research activity on CDMA and other spread-spectrum techniques. This led to the development of a large number of multiuser detection (MUD) techniques, where the inherent structure of interfering signals is exploited to assist the data detection process. The introduction of 3G systems offered a wide range of new multimedia applications with the possibility of speech, audio, images and video transmissions at data rates of 144-384 kbps for fast moving users up to 2 Mbps for stationary or slowly moving terminals. In addition to the increased data rate, other advantages over 2G systems are the improved spectral efficiency, the ability to multiplex several applications with different quality of service requirements, the use of variable bit rates to offer bandwidth on demand and the possibility of supporting asymmetric services in the uplink and downlink directions, which is particularly useful for web browsing and high-speed downloading operations. Unfortunately, the impressive costs paid by telecom providers to get 3G cellular licenses slackened the deployment of the 3G infrastructure all over the world and led to a spectacular crash of the telecom stock market during the years 2000/2001. As a result, many startup companies went bankrupt while others decreased or stopped at all their Investments in the wireless communication area. This also produced a significant reduction of public funding for academic research.[1] 1.2.4 Fourth generation (4G) wireless broadband systems 4G refers to the fourth generation of cellular wireless standards. It is a successor to 3G and 2Gfamilies of standards. The nomenclature of the generations generally refers to a change in the fundamental nature of the service, non-backwards compatible transmission technology and new frequency bands. The first was the move from 1981 analogue (1G) to digital (2G) transmission in 1992.This was followed, in 2002, by 3G multi-media support, spread spectrum transmission and at least200 Kbit/s, soon expected to be followed by 4G,which refers to all-IP packet-switched networks, mobile ultra-broadband (gigabit speed) access and multi-carrier transmission. Pre-4G technologies such as mobile WiMAX and first-release 3G Long Term Evolution (LTE) have been available on the market since 2006and 2009 respectively. It is basically the extension in the 3G technology with more bandwidth and services offers in the 3G.The expectation for the 4G technology is basically the high quality audio/video streaming over end to end Internet Protocol. If the Internet Protocol (IP)multimedia sub-system movement achieves what it going to do, nothing of this possibly will matter. WiMAX or mobile structural design will become progressively more translucent, and therefore the acceptance of several architectures by a particular network operator ever more common.[1]


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1.2.5 Next generation technology 5G (5th generation mobile networks or 5th generation wireless systems) is a name used in some research papers and projects to denote the next major phase of mobile telecommunications standards beyond the upcoming 4G standards, which are expected to be finalized between approximately 2011 and 2016. Currently 5G is not a term officially used for any particular specification or in any official document yet made public by telecommunication companies or standardization bodies such as 3GPP, WiMAX Forum or ITU-R. New 3GPP standard releases beyond 4G and LTE Advanced are in progress, but not considered as new mobile generations.[2] 5G Technology stands for 5th Generation Mobile technology. 5G technology has changed the means to use cell phones within very high bandwidth. User never experienced ever before such a high value technology. Nowadays mobile users have much awareness of the cell phone (mobile) technology. The 5G technologies include all type of advanced features which makes 5G technology most powerful and in huge demand in near future.[2] The gigantic array of innovative technology being built into new cell phones is stunning. 5G technology which is on hand held phone offering more power and features than at least 1000 lunar modules. A user can also hook their 5G technology cell phone with their Laptop to get broadband internet access. 5G technology including camera, MP3 recording, video player, large phone memory, dialing speed, audio player and much more you never imagine. For children rocking fun Bluetooth technology and Pico nets has become in market.[2] 5G technology going to be a new mobile revolution in mobile market. Through 5G technology now you can use worldwide cellular phones and this technology also strike the china mobile market and a user being proficient to get access to Germany phone as a local phone. With the coming out of cell phone alike to PDA now your whole office in your finger tips or in your phone. 5G technology has extraordinary data capabilities and has ability to tie together unrestricted call volumes and infinite data broadcast within latest mobile operating system. 5G technology has a bright future because it can handle best technologies and offer priceless handset to their customers. May be in coming days 5G technology takes over the world market. 5G Technologies have an extraordinary capability to support Software and Consultancy. The Router and switch technology used in 5G network providing high connectivity. The 5G technology distributes internet access to nodes within the building and can be deployed with union of wired or wireless network connections. The current trend of 5G technology has a glowing future.[2] A new revolution of 5G technology is about to begin because 5G technology going to give tough completion to normal computer and laptops whose marketplace value will be effected. There are lots of improvements from 1G, 2G, 3G, and 4G to 5G in the world of telecommunications. The new coming 5G technology is available in the market in affordable rates, high peak future and much reliability than its preceding technologies.[2]


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1.3 Summary of all generations

Table 1 Summary of all Generation Generation Feature

1G 2G 3G 4G 5G

Deployment 1970-1980 1990-2001 2001-2010 2011 2015-20 onwards

Data Rates 2kbps 14.4-64 kbps 2Mbps 200Mbps to 1Gbps 1Gbps and higher

Technology

Analog Cellular Technology

Digital Cellular Technology: Digital narrow band circuit

data, Packet data

Digital Broadband Packet data: CDMA 2000

EVDO, UMTS, EDGE.

Digital Broadband Packet data: WiMax LTE

Wi-Fi

wwww Unified IP seamless

combination of broadband LAN,

PAN, MAN, WLAN

Service Analog voice

service No data service

Digital voice with higher clarity

SMS,MMS Higher capacity packetized

data

Enhanced audio video streaming

video conferencing support Web browsing at

higher speeds IPTV support

Enhanced audio-video streaming IP telephony HD

mobile TV

Dynamic Information access Wearable

devices AI Capabilities

Multiplexing Switching FDMA TDMA,CDMA CDMA CDMA CDMA

Standards MTS, AMTS, IMTS

2G:GSM 2.5:GPRS 2.75:EDGE

IMT-2000 3.5G HSDPA 3.75G

HSUPA Single unified standard LTE,

WiMaX Single unified

standard

WEB Standard www www (IPv4) www (IPv4) wwww (IPv6)

Chapter Two

Orthogonal Frequency Division Multiplexing

Chapter Two T Orthogonal Frequency Division Multiplexing

10

Chapter 2 Orthogonal Frequency Division Multiplexing

Due to the high data rate transmission and the ability to against frequency selective

fading, orthogonal frequency division multiplexing (OFDM) is a promising technique in the current broadband wireless communication system.

Orthogonal frequency division multiplexing (OFDM) technology is to split a high-rate data stream into a number of lower rate streams that are transmitted simultaneously over a number of subcarrier. Because the symbol duration increases for the lower rate parallel subcarrier, the relative amount of dispersion in time causes by multipath delay spread is decreased.

In a classical parallel data system, the total signal frequency band is divided into “N” non-overlapping frequency sub-channels, each sub-channel is modulated with a separate symbol and then the “N” sub-channels are frequency-multiplexed. It seems good to avoid spectral overlap of channels to eliminate inter-channel interference. However, this leads to inefficient use of the available spectrum. To cope with the inefficiency, the ideas proposed from the mid- 1960s were to use parallel data and FDM with overlapping sub-channels, in which, each carrying a signaling rate b is spaced apart in frequency to avoid the use of high-speed equalization and to combat impulsive noise and multipath distortion, as well as to fully use the available bandwidth.

The employment of discrete Fourier transform to replace the banks of sinusoidal generator as shown in “Figure (2.1)”and the demodulation significantly reduces the implementation complexity of OFDM modems.

Fig. 2.1 Discrete Fourier Transform


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Inter-symbol interference is eliminated almost completely by introducing a guard interval with zero padding in every OFDM symbol. In the guard time, the OFDM symbol is cyclically extended to avoid inter-carrier interference (cyclic prefix).

2.1 OFDM Historical Overview In 1971, Weinstein and Ebert applied the Discrete Fourier Transform (DFT) to parallel

data transmission systems as part of modulation and demodulation process. In the 1980s, OFDM was studied for high-speed modems digital mobile communication

and high-density recording and pilot tone is used to stabilize carrier and frequency control in addition to Trellis code is implemented.

In 1980, Hirosaki suggested an equalization algorithm in order to suppress both inter-symbol and inter-carrier interference caused by the channel impulse response or timing and frequency errors.

In the 1990s, OFDM was exploited for wideband data communications Mobile radio FM channels:

Fix-wire network High-bit-rate digital subscriber line (HDSL) Asymmetric digital subscriber line (ADSL) Very-high-speed digital subscriber line (VDSL) Digital audio broadcasting (DAB) Digital video broadcasting (DVB) High-definition television (HDTV) terrestrial broadcasting There exist three mechanisms about the digital terrestrial television broadcasting system

European (COFDM ) Wireless LAN HIPERLAN2 (European)

IEEE 802.11a (U.S.A) IEEE 802.11g (U.S.A)

Now, OFDM technique has been adopted as the new European DAB standard, DVB standard, widely used in all WiMAX implementations, a candidate of 4G mobile communication, in IEEE 802.16 broadband wireless access system standards, and IEEE 802.20 mobile broadband wireless access and more other advanced communications systems.[16 , 17]

2.2 Important Definitions Inter-symbol interference (ISI) and inter-carrier interference (ICI) are very important

phenomena that a signal can face so we will focus here on their detailed meaning.


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2.2.1 Inter-Symbol Interference (ISI) In telecommunication, inter-symbol interference (ISI) is a form of distortion of a signal

in which one symbol interferes with subsequent symbols as shown below in “Figure (2.2)”. This is an unwanted phenomenon as the previous symbols have similar effect as noise, thus making the communication less reliable. ISI is usually caused by multipath propagation or the inherent non-linear frequency response of a channel causing successive symbols to "blur" together. The presence of ISI in the system introduces errors in the decision device at the receiver output. Therefore, in the design of the transmitting and receiving filters, the objective is to minimize the effects of ISI, and thereby deliver the digital data to its destination with the smallest error rate possible. Ways to fight inter-symbol interference include adaptive equalization and error correcting codes.

From the receiver point of view, the channel introduces time dispersion in which the duration of the received symbol is stretched; extending the symbol duration causes the current received symbol to overlap previous received symbols and results in inter-symbol interference (ISI).

Fig. 2.2 ISI Due to channel filtering

2.2.2 Inter-Carrier Interference (ICI) Interference caused by data symbols on adjacent subcarriers, ICI occurs when the multipath channel varies over one OFDM symbol time as shown below in “Figure (2.3)”, when this happens, the Doppler shifts on each multipath component cause a frequency offset on the subcarriers, resulting in the loss of orthogonality among them. This situation can be viewed from the time domain perspective, in which the integer number of cycles for each subcarrier within the FFT interval of the current symbol is no longer maintained due to the phase transition introduced by the previous symbol, Finally, any offset between the subcarrier frequencies of the transmitter and receiver also introduces ICI to an OFDM symbol.


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Fig. 2.3 Overlap of neighboring subcarriers, Inter Carrier Interference (ICI)

2.3 OFDM as a Multicarrier Transmission Technique OFDM is a special case of multicarrier transmission, where a single data stream is

transmitted over a number of lower rate subcarrier. Multi-carrier methods belong to the most complicated transmission methods of all and are in no way inferior to the code division multiple access (CDMA) methods, But why this complexity? The reason is simple: the transmission medium is an extremely difficult medium to deal with.

The terrestrial transmission medium involves Terrestrial transmission paths. Difficult line-associated transmission conditions.

The terrestrial transmission paths, in particular, exhibit the following characteristic features: Multipath reception via various echo paths caused by reflections from

buildings, mountains, trees, vehicles. Additive white Gaussian noise (AWGN). Narrow-band or wide-band interference sources caused by internal combustion

engines, streetcars or other radio sources. Doppler Effect i.e. frequency shift in mobile reception.

Fig. 2.4 Multipath reception leads to location and frequency-selective fading

An effect known as “red-light effect” in car radios, the car stops at a red stop light and radio reception ceases. If one were to select another station or move the car slightly forward, reception would be restored. If information is transmitted by only one discrete carrier precisely at


14

one particular frequency, echoes will cause cancellations of the received signal at particular locations at exactly this frequency. This effect is a function of the frequency as shown in “Figure (2.5)”, the intensity of the echo and the echo delay.

Fig. 2.5 Echo Delay

If high data rates of digital signals are transmitted by vector modulated (I/Q modulated) carriers, they will exhibit a bandwidth which corresponds to the symbol rate.

The available bandwidth is usually specified. The symbol rate is obtained from the type of modulation and the data rate. However, single carrier methods have a relatively high symbol rate, often within a range of more than 1 MS/s up to 30 MS/s. This leads to very short symbol periods of 1 μs and shorter (inverse of the symbol rate). However, echo delays can easily be within a range of up to 50 μs or more in terrestrial transmission channels. Such echoes would lead to inter-symbol interference between adjacent symbols or even far distant symbols and render transmission more or less impossible. An obvious trick would now be to make the symbol period as long as possible in order to minimize inter-symbol interference as shown in “Figure (2.6)”and, in addition, pauses could be inserted between the symbols, so-called guard intervals.

Fig. 2.6 Inter Symbol Interference (ISI)

However, there is still the problem of the location- and frequency selective fading

phenomena. If then the information is not transmitted via a single carrier but is distributed over many, up to thousands of subcarriers and a corresponding overall error protection is built in, the


15

available channel bandwidth remaining constant, individual carriers or carrier bands will be affected by the fading, but not all of them. Many thousands of subcarriers are used instead of one carrier, the symbol rate is reduced by the factor of the number of subcarriers and the symbols are correspondingly lengthened several thousand times up to a millisecond. The fading problem is solved and, at the same time, the problem of inter-symbol interference is also solved due to the longer symbols and the appropriate pauses between them. A multi-carrier method is born and is called Coded Orthogonal Frequency Division Multiplex (COFDM). It is now only necessary to see that the many adjacent carriers do not interfere with one another, i.e. are orthogonal to one another.[18]

2.4 OFDM Concept An OFDM signal consists of a number of closely spaced modulated carriers. When

modulation of any form - voice, data, etc. is applied to a carrier, then sidebands spread out either side. It is necessary for a receiver to be able to receive the whole signal to be able to successfully demodulate the data. As a result when signals are transmitted close to one another they must be spaced so that the receiver can separate them using a filter and there must be a guard band between them. This is not the case with OFDM. Although the sidebands from each carrier overlap, they can still be received without the interference that might be expected because they are orthogonal to each another. This is achieved by having the carrier spacing equal to the reciprocal of the symbol period.

Fig. 2.7 Multi Carrier

To see how OFDM works, it is necessary to look at the receiver. This acts as a bank of

demodulators, translating each carrier down to DC. The resulting signal is integrated over the symbol period to regenerate the data from that carrier. The same demodulator also demodulates the other carriers. As the carrier spacing equal to the reciprocal of the symbol period means that they will have a whole number of cycles in the symbol period and their contribution will sum to zero - in other words there is no interference contribution.


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2.4.1 Analysis To completely understanding the concept of OFDM we make analysis in time and

frequency domains. 2.4.1.1 Time Domain Analysis

Assume a channel that has the following impulse response “h (t)”, if we send a pulse “S(t)” over this channel, the pulse shape would be convolved with the channel impulse response as shown in “Figure (2.8)”.

Fig. 2.8 Channel Impulse Response

Note that the pulse becomes dispersed or extended in time interfering with surrounding pulses and causing inter-symbol interference (ISI) and Now compare the two cases of T >> τ and T < τ:

T < τ: Pulse completely distorted. ISI is significant in this case. T > τ: Pulse extended but the extension is much smaller than “T” the output behaves like

the transmitted rectangular pulse. 2.4.1.2 Frequency Domain Analysis

A wideband signal is completely distorted while a narrow band signal is essentially seeing a flat channel as shown in “Figure (2.9)”.

Fig. 2.9 Flat Channel


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A high data rate transmitted signal has a consequent large bandwidth, this means that it subjected to frequency selective or frequency dependent fading, which can distort the signal significantly, one solution is to divide the bandwidth available for transmission into many narrowband sub channels.

Low data rate sub channel‘s frequency component encounter an almost flat channel. (The band over which the channel is almost constant is called the coherence bandwidth of the channel) Relative to the narrow sub channel, the channel is basically a frequency-independent complex number, i.e.; amplitude and a phase shift. 2.4.1.3 Conclusion from both Domains Analysis

Short pulses suffer severely from channel, But we need these short pulses in order to send a greater number of them at given time period, i.e., to increase data or bit transmission rate, one solution is to use short pulses (high data rate) and an equalizer at receiver, the equalizer is a filter that compensates for the distortion induced by channel characteristics.

For very high data rates a sophisticated equalizer is needed and may not be ever feasible, is there another solution? Yes, stick to long pulses. But how can we then increase the data rate? We can use many frequency channels (called sub channels), and hence the name FDM (Frequency Division Multiplexing), over each of these sub channels the data rate is low, but taken together and since they operate in parallel, a very high data rate can be achieved while circumventing the dispersive influence of the channel.

2.5 Orthogonality of OFDM In OFDM, the spectra of subcarriers overlap but remain orthogonal to each other “Figure

(2.10)”, this means that at the maximum of each subcarrier spectrum, all the spectra of other subcarriers are zero.[19]

Fig. 2.10 Orthogonal subcarriers


18

The receiver samples data symbols on individual subcarriers at the maximum points and demodulates them free from any interference from the other subcarriers and hence no ICI, the orthogonality of subcarriers can be viewed in either the time domain or in frequency domain:

From the time domain perspective, each subcarrier is a sinusoid with an integer number of cycles within one FFT interval.

From the frequency domain perspective, this corresponds to each subcarrier having the maximum value at its own center frequency and zero at the center frequency of each of the other subcarriers. Two functions, , are orthogonal over an interval.

[a, b] mathematically this can be written as shown in “Equation(2.1)”:

The output of the integrator can be expressed as follows in “Equation (2.2)”:

This integral satisfies the orthogonality definition, to implement this integral sine wave

carriers are used, Consequently, the harmonic exponential functions (sine wave carriers) with frequency separation “Δf= 1/TU” are orthogonal as shown below in “Figure (2.11)”.

Fig. 2.11 Block diagram of Orthogonal subcarriers


19

2.6 Comparing FDM to OFDM 2.6.1Frequency Division Multiplexing (FDM)

In Frequency Division Multiplexing system, signals from multiple transmitters are transmitted simultaneously (at the same time slot) over multiple frequencies. Each frequency range (sub-carrier) is modulated separately by different data stream and a spacing (guard band) is placed between sub-carriers to avoid signal overlap as shown “Figure (2.12)”.

Fig 2.12 Subcarrier with Guard Band

2.6.2 Orthogonal Frequency Division Multiplexing (OFDM)

OFDM is a multiplexing technique that divides the bandwidth into multiple frequency sub carriers. OFDM also uses multiple sub-carriers but the sub-carriers are closely spaced to each other without causing interference, removing guard bands between adjacent sub-carriers. Here all the sub carriers are orthogonal to each other. Two periodic signals are orthogonal when the integral of their product, over one period, is equal to zero. The use of OFDM results in bandwidth saving as seen in the “Figure (2.13)”. [20]

Fig. 2.13 Difference between spectra of FDM and OFDM


20

Table 2 Comparison between FDM and OFDM

2.7 OFDM Implementation Keeps a time slot of length TS fixed, considers modulation in frequency direction for each time slot, starts with a base transmit pulse “g(t)”.then obtains frequency-shifted replicas of this pulse, that is, if “g(t) = g0(t)” is located at the frequency “f = 0”, then “gk(t)” is located at “f =f*k”, in contrast to the first scheme, for each time instant l, the set of K (or K + 1) modulation symbols is transmitted by using different pulse shapes “gk(t)”, the parallel data stream excites a filter bank of K (or K + 1) different band pass filters. The filter outputs are then summed up before transmission. This setup is depicted in “Figure (2.14)”.

Fig. 2.14 OFDM Implementation

The transmitted signal in the complex baseband can be represented as shown below in “Equation (2.3)”.

It is obvious that we come back to the first setup if we replace the modulation symbols “Skl” by “Skl exp (−j2πfklTs)” in Equation, such a time-frequency-dependent phase rotation does not change the performance, so both methods can be regarded as equivalent.

Point of Comparison

Orthogonal Frequency Division Multiplexing ( OFDM)

Frequency Division Multiplexing (FDM)

Bandwidth All sub-channels are dedicated to a single data source Bandwidth dedicated to several sources

Carrier Sum of a number of orthogonal carriers No relationship between the carriers Guard band No guard band between carriers There is a guard band between carriers Efficiency Better spectral efficiency Low spectral efficiency

ISI Overcomes ISI and delay spread More subject to ISI and external interference from other RF sources

(


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However, the second – the filter bank – point of view is closer to implementation, especially for the case of OFDM, where the filter bank is just an FFT, as it will be shown later. 2.7.1 Implementation using “FFT/IFFT” In order to overcome the daunting requirement for L RF radios in both the transmitter and the receiver, OFDM uses an efficient computational technique, discrete Fourier transform (DFT) as shown “Figure (2.15)”.

Fig. 2.15 Discrete Fourier Transform (DFT)

Discrete Fourier transform (DFT) leads itself to a highly efficient implementation commonly known as the Fast Fourier Transforms (FFT), this FFT and its inverse operation which is called (IFFT) can create a multitude of orthogonal subcarriers using a single radio.

2.8 OFDM Transmitter The OFDM transmitter shown below in “Figure (2.16)” will be briefly discussed in the next few words. Initially, the bit sequence is first subjected to channel encoding to reduce the probability of error at the receiver due to the channel effects, these bits are mapped to symbols of QPSK, QAM, or any other modulation schematic, the symbol sequence is converted to parallel format, IFFT (OFDM modulation) is applied to convert the block of frequency data to a block of time data that modulates the carrier, the sequence is once again converted to the serial format.

Now OFDM symbol already generated, after then, guard time is provided between the OFDM symbols and the guard time is filled with the cyclic extension of the OFDM symbol, and windowing is applied to the OFDM symbols to make the fall-off rate of the spectrum steeper, the resulting sequence is converted to an analog signal using a DAC and passed on to the RF modulation stage, finally, The resulting RF modulated signal is, then, transmitted to the receiver using the transmit antennas.


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Fig. 2.16 OFDM transmitter

2.9 OFDM Receiver The OFDM receiver shown below in “Figure (2.17)” will be briefly discussed in the next few words. Initially, At the receiver, first RF demodulation is performed, Then the signal is digitized using an ADC, Timing and frequency synchronization are performed, then the guard time is removed from each OFDM symbol, then The sequence is converted to parallel format, After then FFT (OFDM demodulation) is applied to get back to the frequency domain, The output is then serialized. Symbol de-mapping is done to get back the coded bit sequence, and Channel decoding is, then, done to get the user bit sequence.

Fig. 2.17 OFDM receiver


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2.10 Cyclic Prefix The key to making OFDM realizable in practice is the use of the FFT algorithm, due to its low complexity. In telecommunications, the term cyclic prefix refers to the prefixing of a symbol with a repetition of the end. Although the receiver is typically configured to discard the cyclic prefix samples, the cyclic prefix serves two purposes. As a guard interval, it eliminates the inter-symbol interference from the previous symbol, while as a repetition of the end of the symbol; it allows the linear convolution of a frequency-selective multipath channel to be modeled as circular convolution, which in turn may be transformed to the frequency domain using a discrete Fourier transform. In order for the cyclic prefix to be effective (i.e. to serve its aforementioned objectives), the length of the cyclic prefix must be at least equal to the length of the multipath channel. Although the concept of cyclic prefix has been traditionally associated with OFDM systems, the cyclic prefix is now also used in single carrier systems to improve the robustness to multipath. In order for the IFFT/FFT to create an ISI-free channel, the channel must appear to provide a circular convolution, adding cyclic prefix to the transmitted signal, as the following copy “n” values from the end of the symbols sequence and use them in the guard region as shown “Figure (2.18)”.

Fig. 2.18 Guard Period

In addition to preventing interference between original signals, signal after adding cyclic prefix the cyclic prefix above realizes the purpose of the OFDM by making each sub channel see a flat channel response. One must take into consideration that the cyclic prefix must be larger than the delay spread to overcome the inter-symbol interference.[21]


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2.11 Strengths and drawbacks of OFDM 2.11.1 OFDM advantages

Immunity to selective fading: One of the main advantages of OFDM is that is more resistant to frequency selective fading than single carrier systems because it divides the overall channel into multiple narrowband signals that are affected individually as flat fading sub-channels.

Resilience to interference: Interference appearing on a channel may be bandwidth limited and in this way will not affect all the sub channels. This means that not all the data is lost.

Spectrum efficiency: Using close-spaced overlapping sub-carriers, a significant OFDM advantage is that it makes efficient use of the available spectrum.

Resilient to ISI: Another advantage of OFDM is that it is very resilient to inter-symbol and inter-frame interference. This results from the low data rate on each of the sub-channels.

Simpler channel equalization: One of the issues with CDMA systems was the complexity of the channel equalization which had to be applied across the whole channel. An advantage of OFDM is that using multiple sub-channels, the channel equalization becomes much simpler.

2.11.2 OFDM disadvantages Sensitive to carrier offset and drift: It is very sensitive to phase noise and frequency

synchronization errors, which translates into more stringent specifications for local oscillators.

High peak to average power ratio: An OFDM signal has a noise like amplitude variation and has a relatively large dynamic range, or peak to average power ratio. This impacts the RF amplifier efficiency as the amplifiers need to be linear and accommodate the large amplitude variations and these factors mean the amplifier cannot operate with a high efficiency level.

Chapter Three

The Peak-to-Average

power Ratio

Chapter Three The Peak to Average Power Ratio

27

Chapter 3 The Peak-to-Average power Ratio

3.1 Introduction to Peak to Average Power Ratio (PAPR) In the orthogonal frequency division multiplexing (OFDM) the peak power might be much larger than the average power, due to adding up subcarriers coherently which resulting in large peak-to-average power ratio (PAPR). PAPR is a very important situation in the communication system because it has big effects on the transmitted signal. Low PAPR makes the transmit power amplifier works efficiently, on the other hand, the high PAPR makes the signal peaks move into the non-linear region of the RF power amplifier which reduces the efficiency of the RF power amplifier. In addition, high PAPR requires a high-resolution digital- to- analog converter (DAC) at the transmitter, high-resolution analog -to -digital converter (ADC) at the receiver and a linear signal. Any non-linearity in the signal will cause distortion such as inter-carrier-interference (ICI) and inter symbol interference (ISI). [3] There are a number of techniques to deal with the problem of PAPR. Some of them are "amplitude clipping", "clipping and filtering", "coding", "partial transmit sequence (PTS)", "selected mapping (SLM) ‟ and "interleaving‟. These techniques achieve PAPR reduction at the expense of transmit signal power increase, bit error rate (BER) increase, data rate loss, computational complexity increase, and so on.

3.2 Definitions of PAPR Presence of large number of independently modulated sub-carriers in an OFDM system the peak value of the system can be very high as compared to the average of the whole system. This ratio of the peak to average power value is termed as Peak-to-Average Power Ratio

Coherent addition of N signals of same phase produces a peak which is N times the average signal. The PAPR effect is shown in "Figure. 3.1". And it can be seen that the peak power is about 17 times the average power.


28

Fig. 3.1 Power samples of one symbol OFDM signal

This figure shows that one of the samples have high peaks, and in order to transmit a signal with these high peaks, a high signal span is required from the power amplifier. But such types of power amplifiers are costly. [3] 3.2.1 (PAPR) of a continuous time signal For a continuous time baseband OFDM signal, the PAPR of any signal is defined as the proportion of the maximum instantaneous power of the signal and its average power. If x (t) is a transmitted baseband OFDM signal, then PAPR is defined as:

Where, is the original signal. Τ is the interval time. is the peak signal power. is the average signal power.


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3.2.2 (PAPR) of a discrete time signal For a discrete OFDM signal, the PAPR is computed from 'L' time oversampled OFDM signal as:

Where, L is the discrete signal. N is the total number of subcarriers. is the expectation.

3.3 PAPR of OFDM signal "Figure 3.2" shows the end-to-end block diagram of an OFDM system in which the discrete-time signal after IFFT at the transmitter can be expressed as:

(3.4)

Fig. 3.2 Block diagram of OFDM system.

Let's have a look on the mathematical explanation of the PAPR. The input bit stream given to OFDM system is first mapped on a selected modulation technique Ex: QAM, QPSK, BPSK etc. From this, a complex OFDM vector of symbols is obtained. Let’s this complex valued


30

vector is a data block 'X' consisting of complex data , , , .the complex base band representation of this multicarrier signal can be determined by the above "Equation 3.4". In the above equation ,'N' tells the number of subcarriers in the OFDM system. is the modulated data carried by the sub-carrier. This equation is actually performing the Inverse Fast Fourier Transform (IFFT) through which, a time domain symbol is obtained .it can be represented as: x=IFFT(X). Therefore, 'x' represents a vector of N elements (o to N-1) of time domain signals and x (n) is element. Whereas 'X' is a set of modulated data carried by 'N' different subcarriers. 3.3.1 Calculation of PAPR Now the peak to Average power ratio of this transmitted signal can be calculated in dB from equation (3.5) given below

(3.5)

In order to observe the PAPR performance of a signal, the Cumulative Distributed Function (CDF) or the Complementary Cumulative Distribution Function (CCDF) is used. Through the CCDF, it can be observed that the PAPR of the signal exceeded a certain value. 3.3.2 Complementary Cumulative Distribution Function In the modern communication word, CCDF measurements prove as one of the precious tool. The CCDF plots offer a comprehensive analysis of signal power peaks. It is a statistical technique that provides the amount of time, a signal spends any given power level. However by using these plots, a probability can be seen that a signal data block exceeds a given threshold. These CCDF plots can be used to analyze the PAPR performance of the signal. Mathematical Explanation: in order to calculate the CCDF of a given data, the following steps should be followed:

Cal. The probability density function (PDF) of the data Take the integral of the PDF to get the CDF (Cumulative Distributed Function) Subtract the CDF from '1' to get the CCDF (as: CCDF=1-CCDF) Or equivalently, it can summarize as: CDF= & CCDF=1-CCDF (3.6) Mathematically, it can be explained as follows:


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P (PAPR>z) =1-P (PAPR z) =1- =1- (3.7) Where, F (z) represent the CDF and z is the given reference level. The histogram plots for the real part, imaginary part and the absolute value of a time domain OFDM signal are shown in Figure 3.3, 3.4 and 3.5 respectively. The plots shown in Figures 3.3(a) and (b) are obtained after performing the computer simulations of an OFDM system having N=256 QPSK modulated subcarriers as shown in "Figure 3.2". The signal obtained from IFFT block of “Figure 3.2"is complex OFDM signal. After that real, imaginary and absolute values of OFDM signal (x[n]) are calculated and their histograms are plotted. [5] The power of OFDM signal has chi-square distribution. The distribution of PAPR is often expressed on the one hand Complementary Cumulative Distribution Function (CCDF). In probability theory and statistics, the CCDF describes the probability that a real-valued random variable X with a given probability distribution will be found at a value greater than or equal to x. [5 and 6]

Fig. 3.3 Histogram of Real part of OFDM signal amplitude

Fig. 3.4 Histogram of Imaginary part of OFDM signal amplitude


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Fig. 3.5 OFDM signal magnitude

3.4 Identification of the Problem Multi-carrier phenomena is considered to be one of the major development in wireless communication and among them OFDM is becoming the important standard. However, high PAPR is the major drawback of OFDM, which results in lower power efficiency hence impedes in implementing OFDM. To overcome the low power efficiency requires not only large back off and large dynamic range DAC but also highly efficient HPA and linear converters. These demands result in costly hardware and complex systems. Therefore to lessen the difficulty of complex hardware design it has become imperative to employ efficient PAPR reduction techniques. This high RAPR can make a cause of certain drawbacks for Example: Increased complexity in the analog to digital and digital to analog converter.

Fig. 3.6 OFDM transmitter basic scheme In some applications, the drawbacks of high PAPR overcome the benefits of OFDM To understand how and where the PAPR of the signal becomes a problem, it is necessary to take a look at the OFDM transmitter," Figure 3.6" shows a basic scheme. The information data is generated and shaped in the Data block, this takes place in frequency domain, then comes the IFFT block and the signal is afterwards in digital time domain. To send the signal through the


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antenna, it must be first converted to analog time domain by means of an D/A converter and then amplified with an RF power amplifier. After the RF amplifier comes the antenna which sends the signal to the OFDM receiver over the wireless channel.

Reduction is efficiency of RF amplifiers

Before transmitting the signal it must path through present high power amplifier (HPA). We already know that we have two main types of power amplifiers which are:

Linear Power Amplifiers (Ideal): They are perfect amplifiers from linearity as shown below in “Figure (3.7)” while its destructive disadvantage operating with very low efficiency of maximum 25% so if we used it the battery of the mobile phone will rapidly exhausted.

Fig. 3.7 Linear and Non-Linear Power Amplifier

Non Linear Power Amplifiers(Actual): They are perfect amplifiers from the

efficiency point of view as they can operate with very high efficiency hence battery can survive for long time, and on the other side this type main disadvantage is operating in a nonlinear region of operation as shown above in “Figure (3.7)” which leads to loss of orthogonality & hence leads to intermarries interference.

When transmitted through a nonlinear device, a high peak signal generates out-of-band energy (spectral re-growth) and in-band distortion (constellation tilting and scattering). These degradations may affect the system performance severely, so, we must operate our system in the linear region of the nonlinear power amplifier. [4 and 9].


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3.5 PAPR Effect

The power amplifiers at the transmitter need to have a large linear range of operation.

Non-linear distortions and peak amplitude limiting introduced by the High Power amplifier (HPA) will produce inter-modulation between the different carriers and introduce additional interference into the system.

Additional interference leads to an increase in the Bit Error Rate (BER) of the system.

The Analog to Digital converters and Digital to Analog converters need to have a wide dynamic range and this increases complexity.

A way to avoid non-linear distortion is by forcing the amplifier to work in its linear region, but unfortunately such solution is not power efficient and thus not suitable for wireless communication. And at manually clipping the output signal in-band distortion (additional noise) and out-of-band radiation (ACI) took place.

3.6 Factors influencing the PAPR 3.6.1 The number of sub carriers In Multi-Carrier Systems the complex base band signal for one symbol in an OFDM system can be expressed as follows: t (3.8) Where N is the modulating symbol and is the number of sub carriers. For moderately large numbers of m-PSK (multiple phase-shift keying) sub carriers the quadrature components of x (t) each tends towards a Gaussian distribution (giving the sum of their power amplitude a Rayleigh distribution). Consequently, whilst the peak value possible is N times the individual sub carrier peak, the probability of any value close to that peak occurring is very low. For example, with only 24 sub carriers, the probability of the PAPR exceeding 4dB is and of exceeding 8dB is only .[5] 3.6.2 The order of Modulation High data bandwidth efficiency (in terms of b/s/Hz) this can be achieved by utilizing higher order modulations based, for instance, on QAM. When using a higher-order modulation such as QAM type, the PAPR of the summed OFDM signal is increased by the PAPR of the QAM constellation utilized. Nevertheless, the probability of these higher peaks happening is accordingly less. Furthermore, since among benefits of OFDM is one that sub carriers could


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have their modulation independently varied to adapt to channel conditions, the joined PAPR in any system utilizing this technique might are hard to predict and control. PAPR for an unfiltered base band signal is listed in the following Table 3 . [8]

Table 3 PAPR for picked modulation formats Modulation PAPR[dB]

M-PSK (reference) 0 16-QAM 2.55 64-QAM 3.68 256-QAM 4.23 256-QAM (modified) 2.85 3.6.3 Constellation shape The last entry in Table 3 is for a constellation obtained by modifying 256- QAM to reduce PAPR. This modified constellation shape is shown in figure 3.8. However, there is an additional processor load associated with encoding and decoding this constellation.

Fig. 3.8 256-QAM constellations: (a) regular and (b) modified mapping to reduce

PAPR 3.6.4 Pulse Shaping In terrestrial communications, it is popular to use pulse shaping to the base band signal, to decrease the bandwidth of the transmitted spectrum, but this causes overshoot and can increase the PAPR of the modulating signal by 4-5 dB [8].


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3.7 Existing Approaches for PAPR Reduction As in everyday life, we must pay some costs for PAPR reduction. There are many factors that should be considered before a specific PAPR reduction technique is chosen. These factors include PAPR reduction capability, power increase in transmit signal, BER increase at the receiver, loss in data rate, computational complexity increase, and so on. Next we briefly discuss each of them and corresponding known PAPR reduction methods. You can find more details about these techniques in "Chapter 5".

PAPR Reduction Capability Careful attention must be paid to the fact that some techniques result in other harmful effects. For example, the Amplitude Clipping technique in Page 80, clearly removes the time domain signal peaks, but results in in-band distortion and out-of-band radiation.

Power Increase in Transmit Signal Some techniques require a power increase in the transmit signal after using PAPR reduction techniques. For example, Tone Reservation (TR) , requires more signal power because some of its power must be used for the peak reduction carriers. Tone Injection (TI) in page 73, uses a set of equivalent constellation points for an original constellation point to reduce PAPR. Since all the equivalent constellation points require more power than the original constellation point, the transmit signal will have more power after applying TI.

BER Increase at the Receiver other techniques may have an increase in BER at the receiver if the transmit signal power is fixed or equivalently may require larger transmit signal power to maintain the BER after applying the PAPR reduction technique. For example, the BER after applying Active Constellation Extension in Page 69, will be degraded if the transmit signal power is fixed. In some techniques such as SLM in Page 56, the entire data block may be lost if the side information is received in error. This may also increase the BER at the receiver. Loss in Data Rate Some techniques require the data rate to be reduced, the block coding technique needs a portion of information symbols to be dedicated to control the PAPR. In SLM and PTS, the data rate is reduced due to the side information used to inform the receiver of what has been done in the transmitter. In these techniques the side information may be received in error unless some form of protection such as channel coding is employed. When channel coding is used, the loss in data rate due to side information is increased further.

Computational Complexity Techniques such as PTS in chapter 5 found a solution for the PAPR reduced signal by using many iterations and performing exhaustive search, so it costs a lot to implement it in hardware. This report will propose a low complexity version of PTS which use more intellectual method to find optimum phase for PAPR minimization. [10, 11, 12]


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3.8 PAPR Reduction Techniques Several PAPR reduction techniques have been proposed in the literature .These techniques are divided into two groups - signal scrambling techniques and signal distortion techniques which are given below:

a) Signal Scrambling Techniques: Block Coding Techniques Block Coding Scheme with Error Correction Selected Mapping (SLM) Partial Transmit Sequence (PTS) Interleaving Technique Tone Reservation (TR) Tone Injection (TI) Signal scrambling techniques work with side information which minimized the effective throughput since they commence redundancy. Signal distortion techniques introduce band interference and system complexity also. Signal distortion techniques minimize high peak dramatically by distorting signal before amplification.

b) Signal Distortion Techniques: Peak Windowing Envelope Scaling Peak Reduction Carrier Clipping and Filtering

Table 4 PAPR Reduction Techniques


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Fig. 3.9 Types of PAPR Reduction Techniques

Chapter Four

Getting Start with Software

Defined Radio

Chapter Four Getting Start with Software Defined Radio

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Chapter 4 Getting Start with Software Defined Radio

4.1 Software Defined Radio Definition: SDR is an idea which has been in focus since the early nineties. The main purpose of Software Defined Radio was to create a device which should be capable of working with many radios operating at different parameters. Moreover, it can adjust to any range and any modulation scheme by using a powerful software along with programmable hardware. An alternative definition for SDR is to merge hardware and software technologies to make the system flexible for wireless communication [7] [8]. Software Defined Radio provide an effective and inexpensive solution for building multi-mode, multi-range and multifunctional wireless devices that can be improved by means of software advancements. By using SDR enabled devices, the same piece of “hardware" can be modified to perform different functions at different times. SDR performs noteworthy amount of signal processing in a common PC. In SDR, signal handled in digital domain instead in analog domain as in the conventional radio. The analog signal can be converted to the digital domain with the help of Analog to Digital Converter. This figure shows that the ADC process is taking place after the Front End (FE) circuit. The A/D converter will convert the signal to digital form and pass it to the baseband processor for more processes; detection, channel coding, source coding and etc.

Fig. 4.1 SDR Sender and Receiver module diagram


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4.2 The Hardware of USRP and Software of Labview Communication between the USRPs and the computers is handled by LabVIEW drivers. Initial user input includes the TX/RX frequency, IQ sample rate, IP address of the USRP, and the TX/RX gain. Thereafter LabVIEW can continuously fetch or transmit samples and then close the connection. The USRPs also have the ability to share information and communicate with each other through a MIMO (multiple input multiple output) cable. This was used solely for debugging parameters like internal vs actual frequency offset. "Figure 4.2" shows the high level flow of data, from LabVIEW on a computer to a USRP, into the air, received on another USRP, and recovered in LabVIEW on another computer. All arrows are bidirectional, because the hardware supports duplex communications.

Fig. 4.2 USRP Connection Diagram


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4.2.1 Hardware Specifications The features of NI-USRP-2920 are discussed below:

Table 5 The Features of NI-USRP-2920 TX & RX

Parameter Value in Transmitter Kit Value in Receiver Kit Frequency Range 50 MHz to 2.2 GHz 50 MHz to 2.2 GHz Frequency Step < 1 kHz < 1 kHz

Max. Output Power (Pout) 50 MHz to 1.2 GHz 50 mW to 100 mW (17 dBm to 20 dBm)

1.2 GHz to 2.2 GHz 30 mW to 70 mW (15 dBm to 18 dBm)

----------------------

Max. Input Power (Pin) ---------------------- 0 dBm Gain Range 10 dB to 31 dB 0 dB to 31.5 dB Gain Step 1.0 dB 0.5 dB Frequency Accuracy 2.5 ppm 2.5 ppm Max. Real-Time instantaneous-bandwidth

16-bit sample-width 20 MHz 8-bit sample-width 40 MHz

16-bit sample-width 20 MHz 8-bit sample width 40 MHz

Max. I/Q Sampling Rate 16-bit sample-width 25 MS/s 8-bit sample-width 50 MS/s

16-bit sample-width 25 MS/s 8-bit sample-width 50 MS/s

DAC 2 Channels 16-bit, 400 MS/s 14 bit, 100 MS/s


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4.2.2 Connectors of NI USRP-2920 The front end connectors of the USRP-2920 kit are discussed in the following table.

Table 6 Connectors of NI USRP-2920

Connector Use RX1 TX1

RF signal input and output terminal. RX1/TX1 is an SMA (f) connector with an impedance of 50 Ω, and is a single-ended input or output channel.

RX2

RF signal input and output terminal. RX2 is an SMA (f) connector with an impedance of 50 Ω, and is a single-ended input or output channel.

REF IN Input terminal for an external reference signal for the local oscillator (LO) on the NI USRP-2921. REF IN is an SMA (f) connector with an impedance of 50 Ω, and is a single-ended reference input. REF IN accepts a 10 MHz signal with a range of 0 dBm to 15 dBm.

PPS IN Pulse per second timing reference input terminal. PPS IN is an SMA (f) connector with an impedance of 50 Ω, and is a single-ended input. PPS IN accepts 0 V to 3.3 V TTL and 0 V to 5 V TTL signals.

MIMO EXPANSION

The multiple-input, multiple-output (MIMO) EXPANSION interface port connects two USRP devices using a compatible MIMO cable.

GB ETHERNET The gigabit Ethernet port accepts an RJ-45 connector and gigabit Ethernet compatible cable (Category 5, Category 5e, or Category 6).

POWER The power input accepts a 6 V, 3 A external DC power connector.


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4.2.3 NI USRP-2920 Module LED The module LEDs are listed in the following table:

Table 7 NI USRP-2920 Module LEDs

LED Indication A

Indicates the transmit status of the NI USRP-2921 module: OFF—The module is not transmitting data. GREEN—The module is transmitting data.

B Indicates the status of the physical MIMO cable link: OFF—The devices are not connected using the MIMO cable. GREEN—The devices are connected using the MIMO cable.

C Indicates the receive status of the NI USRP-2921 module: OFF—The device is not receiving data. GREEN—The module is receiving data.

D Indicates the firmware status of the NI USRP-2921 module: OFF—The firmware is not loaded. GREEN—The firmware is loaded.

E Indicates the reference lock status of the LO on the NI USRP-2921 module: OFF—There is no reference signal or the LO is not locked to a reference signal. BLINKING—The LO is not locked to a reference signal. GREEN—The LO is locked to a reference signal.

F Indicates the power status of the NI USRP-2920 module: OFF—The device is powered off. GREEN—The device is powered on.


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4.2.4 NI USRP-2920 Front Panel The front panel of the USRP kit for the demonstration is shown in the following figure. The use of all the ports available is also shown in the diagram itself

Fig. 4.3 Detailed view of Front Panel of NI USRP-2920

Fig. 4.4 NI USRP-2920 Front Panel


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1) MIMO Expansion: The MIMO expansion cable is used to link a pair of USRP systems together. Length is 0.5 meters. Only one cable needed for every linked pair of USRP systems.

Fig. 4.5 MIMO Expansion cable

2) Antenna :

This version works with a central frequency in the range from 50 MHz up to 2.2 GHz, covering the FM radio, GPS, GSM, OFDM and radar bands.

Fig. 4.6 USRP Antenna


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3) Gigabit Ethernet: The Gigabit Ethernet is used to send the signal data from PC to USRP and from USPR to PC with a speed of up to 25 Msample/sec.

Fig. 4.7 Gigabit Ethernet Cable

4) Power Supply:

The power supply for USRP devices with input at 100 - 240 V, 50-60 Hz, 0.6A and output at 6V DC 3A.

Fig. 4.8 Power Supply


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4.3 USRP Parameters configuration

Fig. 4.9 The USRP Configuration

1) Device Name: IP address of one or multiple USRP.

2) IQ Rate: Quadrature sampling rate Equivalent to Bandwidth.

3) Carrier Frequency: Frequency of interest.

4) Antenna: Select which antenna port to receive from.

5) Gain: Amplification of signal before the digitizing the signal.

6) Fetch Size: how many samples to acquire each fetch.


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4.4 System Implementation For the OFDM system implementation with the help of LabVIEW and NI-USRP 2920, there were again two options available. Either we could use the different communication or signal processing blocks that are available to us. We could also design these blocks as per our requirement. OFDM system implementation was experimented in LabVIEW by using these blocks. 4.4.1 NI-USRP 2920 Block diagram The block diagram of NI-USRP 2920 kit is shown in the figure 4.2. The NI USRP kit connects to the PC to serve as a software-defined radio. For the purpose of transmission, baseband I/Q signal samples are generated by the computer and given to the USRP-2920 kit at the rate of 20 MS/s over Gigabit Ethernet. These are represented with 32-bits (16-bits for the in-phase and 16 for the quadrature phase components).

Fig. 4.10 Detailed block diagram of NI-USRP 2920

The USRP mixes the incoming signal with 400 MS/s with the help of a digital up-convertor (DUC) and then transforms the signal from digital form to analog with a dual-channel, 16- bit DAC. The resulting signal in analog form is then mixed up with some specified carrier frequency.


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4.4.2 Interfacing of the Host PC with the Kit Before installing the software for the functioning of USRP, we need to install the latest version of LabVIEW available in the market is LabVIEW 2015. After installing the LabVIEW software, follow the following steps:

NI USRP Software Configuration Utility should be inserted into the PC and installed. There are some optional products which if required may be installed such as: LabVIEW Modulation Toolkit, LabVIEW Digital Filter Design Toolkit and LabVIEW Math script RT module. Keep the host PC powered on. The power cable should be connected to the USRP kit as shown in the following figure. Now, attach the cable or antenna to the terminals of the NI USRP-2920 front panel according to the requirement.

Connect the device directly to your computer with the included Ethernet cable as shown in the following figure.

Insert the Ethernet cable in the available slot on the host PC. The whole procedure must be followed as shown in the following figure.

Fig. 4.11 Power connections to the USRP kit


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Fig. 4.12 Ethernet cable and antenna connections to the USRP

Fig. 4.13 Connections of the NI-USRP kit to the host PC

After ensuring that all the connections, are correct, setting up the network takes some

time to start the communication with the USRP device.


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The IP addresses-for the computer-and each connected USRP-device must be unique.

To confirm the network connection, we have to open the NI-USRP-Configuration Utility. The following window will be displayed.

Fig. 4.14 Verifying the device in NI-USRP-Configuration Utility

On this window, go to the Change IP Address tab of the utility. Your device should

appear in the selected IP address available on the left side of the tab.

If the device name is not present in the list, check all the connections and power supply again, then click the “find devices” button to scan for USRP devices.

We can also change the IP address of the device by selecting the device from the list. The IP address of the device which we select is displayed in the Selected IP Address textbox.

We can enter the new IP address which for the device in the New IP Address textbox shown in the following figure:


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Fig. 4.15 Changing the device IP of the device

After this step, we were ready to use the software so as to design and implement the System model.

Chapter Five

Simulation and

Implementation Results

Chapter Five Simulation and Implementation Results

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Chapter 5 Simulation and Implementation Results

In this chapter the techniques, simulation and implementation results are shown and discussed in the following sections, first we will present the explanation of all six techniques from point of view : Block Diagram, Overview and Equations with the simulation results and then we will present the implementation results of Partial Transmit Sequence (PTS) technique.

5.1 The overall system design specifications The overall system design specifications are given in the following table :

Table 8 The System Parameters used for Simulation Parameters Value used

Symbol rates supported 5 Msps Channel coding repetition (3,1) code Modulation schemes supported 4-QAM, 16-QAM, 64-QAM and

256-QAM Pulse Shaping Raised cosine pulse shape with

roll-off 0.5 Passband Bandwidth 5 MHz Number of subcarriers N = (FFT Size)

64 Length of Cyclic Prefix Lc 8 PAPR Oversample Factor 4 Number of generated OFDM signal 10000 Symbol Timing Extraction Max Energy, Early-Late Gate

Method Channel Estimation & Equalization IEEE 802.11a training sequence

5.2 OFDM Physical Layer Any Communication System should be contain some of these block : Source, Channel Encoder, Modulation, Add Control, filter ... etc in transmitter and contain : Matched Filter, Strip Control, Demodulation, Channel Decoder, ... etc in receiver. The block diagram of basic OFDM System consists a several blocks such as : Mapper, S/P, P/S, Insert Null Tones, IFFT, Cyclic Prefix , ...etc in transmitter and FFT, remove CP , remove Null Tones, De-mapper, ... etc in receiver.


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The following figures show the physical layer of USRP communication system that used in our project , the Transceiver of OFDM System and how we built it using software Labview.

Fig. 5.1 The block diagram of the Communication System

Fig. 5.2 The block diagram of OFDM Transceiver


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Fig. 5.3 The block diagram of OFDM transmitter using Labview

Fig. 5.4 The block diagram of OFDM Receiver using Labview


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5.3 The Simulation Results Now, we have to explain all techniques and show the simulation results of each one. These techniques are divided into two groups: signal scrambling techniques and signal distortion techniques. 5.3.1 Selected Level Mapping (SLM) This method is used for minimization of peak to average transmit power of multi-carrier transmission system with selected mapping. A complete set of candidate signal is generated signifying the same information in selected mapping, and then concerning the most favorable signal is selected as consider to PAPR and transmitted. In the SLM, the input data structure is multiplied by random series and resultant series with the lowest PAPR is chosen for transmission. To allow the receiver to recover the original data to the multiplying sequence can be sent as ‘side information’. One of the preliminary probabilistic methods is SLM method for reducing the PAPR problem. The good side of selected mapping method is that it doesn’t eliminate the peaks, and can handle any number of subcarriers. The drawback of this method is the overhead of side information that requires to be transmitted to the receiver of the system in order to recover information. 5.3.1.1 Block Diagram

Fig. 5.5 SLM General Block Diagram


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Fig. 5.6 SLM Block Diagram using Labview

5.3.1.2 SLM Equations The algorithm is as follows: Step 1: Sequences of data bits are mapped to constellation points M-QAM or BPSK to produce sequence symbols X0, X1, X2. . . Step 2: These symbol sequences are divided into blocks of length N. where N is the number of subcarriers. Step 3: Each block X = [X0, X1, X1 ,. . . XN-1] is multiplied (point wise multiplication) by U different phase sequence vectors B(u) = [B0(u) , B1(u) ,...., BN-1(u)] where each row of the normalized Riemann matrix B is taken as B(u), u=[1,2, . . ., U]. Step 4: A set of U different OFDM data blocks X(u) = [X0(u), X1(u),... , XN-1(u)] are formed, where Xn(u)= Xn. B(u) , n = 0, 1, . . ., N-1 , u = 1, 2, ..., U. Step 5: Transform X(u) into time domain to get x(u) = IDFTX(u). Step 6: Select the one from x(u) u = 1, 2, . . .,U, which has the minimum PAPR and transmit it.


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5.3.1.3 SLM Results 1) Received Constellation

Fig. 5.7, 5.8, 5.9 and 5.10 shows the shape of Received Constellation (Scattering plots) for 4, 16, 64 and 256 -QAM, and how the SLM technique is effect on them.

4-QAM Modulation

Fig. 5.7 Received 4-QAM Constellation for SLM

16-QAM Modulation



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64-QAM Modulation


256-QAM Modulation



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2) CCDF for SLM Fig. 5.11 shows the CCDF of PAPR for a 4,16,64 and 256-QAM/OFDM system without SLM and with SLM technique.

Fig. 5.11 CCDF for SLM with different types of Modulation

Table 9 CCDF values for SLM with different types of Modulation

Modulation Types PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No SLM ) 10 4-QAM 6 16-QAM 6 64-QAM 6.5 256-QAM 7

As shown in figure 5.11 : It is seen that the PAPR performance improves as the number of Modulation type decrease with M = 4, 16, 64 and 256. The Basic OFDM curve has PAPR equals to 10 dB. After applying SLM technique, the value was significantly reduced to 6 dB for 4-QAM and 16-QAM , 6.5 dB using 64-QAM and 7 dB using 256-QAM.


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5.3.2 Partial Transmit Sequences (PTS) In PTS, the original data block is divided into multiple non-overlapping sub-blocks. Then these sub-blocks are rotated with rotation factors which are statistically independent. After that, the signal with the lowest PAPR is chosen for transmission.[15] There are several ways for the partition of the data sequence into multiple sub-blocks, including adjacent partition, interleaved partition and pseudorandom partition . Among them, pseudo-random partitioning has been found to be the best choice.[15] The major drawback of PTS is also the computational complexity (complexity for optimal phase factor, and more than one IFFT blocks) and low data rate (required side information). Several techniques have been proposed in the literature to reduce the search complexity and overhead (by reducing/avoiding the usage of side information). The complexity of PTS is less than SLM.[15] In PTS method, the original frequency-domain data sequence is divided into multiple disjoint sub-blocks, which are then weighted by a set of phase sequences to create a set of candidates Finally, the candidate with the lowest PAPR is chosen for transmission.[15] 5.3.2.1 Block Diagram

Fig. 5.12 PTS General Block Diagram


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Fig. 5.13 PTS Block Diagram using Labview

5.3.2.2 PTS Equations In the PTS technique input data block X is partitioned into V or M disjoint sub blocks: The sub blocks are combined to minimize the PAPR in the time domain. The set of phase factors is denoted as a vector: The time domain signal after combining is given by: The objective is to find the set of phase factors that minimizes the PAPR. Minimization of PAPR is related to the minimization of max [x’ (b)].


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5.3.2.3 PTS Results

4-QAM Modulation Fig. 5.14 and 5.15 shows the shape of Received Constellation (Scattering plots) for 4-QAM, and how the PTS technique is effect on it when we changing the number of subblocks (V = 4,16,32 and 64) .

Fig. 5.14 Received 4-QAM Constellation for PTS (V=4 & V=16) respectively



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Fig. 5.16 : shows the CCDF of PAPR for a 4-QAM/OFDM system without PTS and with PTS technique as the number of subblocks varies.

Fig. 5.16 CCDF for PTS 4-QAM with different number of Sub-blocks (V)

Table 10 CCDF values for PTS 4-QAM Modulation with different number of sub-blocks(V)

Sub-Blocks (V) PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No PTS ) 10 Proposed V = 4 8 Proposed V = 16 6 Proposed V = 32 3.5 Proposed V = 64 2.5


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16-QAM Modulation Fig. 5.17 and 5.18 shows the shape of Received Constellation (Scattering plots) for 16-QAM, and how the PTS technique is effect on it when we changing the number of subblocks (V = 4,16,32 and 64).




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Fig. 5.19 shows the CCDF of PAPR for a 16-QAM/OFDM system without PTS and with PTS technique as the number of subblock varies.

Fig. 5.19 CCDF for PTS 16-QAM with different Sub-blocks (V)


Sub-Blocks (V) PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No PTS ) 10 Proposed V = 4 6 Proposed V = 16 5 Proposed V = 32 3.5 Proposed V = 64 3.5


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64-QAM Modulation

Fig. 5.20 and 5.21 shows the shape of Received Constellation (Scattering plots) for 64-QAM, and how the PTS technique is effect on it when we changing the number of subblocks (V = 4,16,32 and 64).




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Sub-Blocks (V) PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No PTS) 10 Proposed V = 4 7 Proposed V = 16 4.5 Proposed V = 32 3.5 Proposed V = 64 3.5 As shown in figure 5.16, 5.19, 5.22 : It is seen that the PAPR performance improves as the number of sub blocks increases with V = 4, 16, 32, and 64. The Basic OFDM curve has PAPR equals to 10 dB. After applying PTS technique , the value was significantly reduced to 2.5 dB using V=64 for 4-QAM , 3 dB using V=64 for 16 QAM and 3.5 dB using V=64 for 64-QAM. This proves that PTS gives better results which is superior performance in PAPR reduction. It is seen that the PAPR performance improves as the number of sub blocks increases with V = 4, 16, 32, and 64.


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5.3.3 Active Constellation Extension( ACE ) Active Constellation Extension (ACE) is another technique which is used for PAPR reduction. The dynamic extension of data block's outer signal constellations to periphery of original constellation in such a way that PAPR is decreased, this is the basic theme of this technique. This idea is schematically represented in figure 5.23 and can be elucidated for multicarrier transmission utilizing QPSK modulation for all sub-carriers.

Fig. 5.23 Active Constellation Extension (a) for QPSK (b) for 16 QAM This technique can also be used with MPSK and QAM. The main advantages of technique include significant reduction of PAPR without compromising data rate and no need for side information. There is additional slight decrease in BER also. The drawback is that the technique is useful for larger constellation size modulations only. 5.3.3.1 Block Diagram

Fig. 5.24 ACE Block Diagram using Labview


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5.3.3.2 ACE Results 1) Received Constellation

Fig. 5.25, 5.26, 5.27 and 5.28 shows the shape of Received Constellation (Scattering plots) for 4, 16, 64 and 256 -QAM, and how the ACE technique is effect on them.

4-QAM Modulation

Fig. 5.25 Received 4-QAM Constellation for ACE

16-QAM Modulation



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64-QAM Modulation


256-QAM Modulation



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2) CCDF for ACE Fig.5. 29 shows the CCDF of PAPR for a 4,16,64 and 256-QAM/OFDM system without ACE and with ACE technique.

Fig. 5.29 CCDF for ACE with different types of Modulation

Table 13 CCDF values for ACE with different types of Modulation

Modulation Types PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No ACE ) 10 4-QAM 5.5 16-QAM 6 64-QAM 7.5 256-QAM 7 As shown in figure 5.29 : It is seen that the PAPR performance improves as the number of Modulation type decrease (usually) with M = 4, 16, 64 and 256. The Basic OFDM curve has PAPR equals to 10 dB. After applying ACE technique , the value was significantly reduced to 5.5 dB for 4-QAM ,6 dB for 16-QAM , 7.5 dB using 64-QAM and 7 dB using 256-QAM.


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5.3.4 Tone Injection (TI) The idea of tone injection (TI) is to increase the constellation size so that each point in the original constellation can be mapped to several equivalent points in the expanded constellation. Since each symbol in a data block is with more mapping choices, it is likely to achieve better PAPR performance. The method is called tone injection because substituting a point in the original constellation for a new point in the expanded constellation is equivalent to add a tone carrying the information of the vector between the two points. The tone injection method may introduce a power increase in the transmit signal due to the injected signal. 5.3.4.1 Block Diagram

Fig. 5.30 TI General Block Diagram

Fig. 5.31 TI Block Diagram using Labview


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5.3.4.2 TI Results 1) Received Constellation

Fig. 5.32, 5.33, 5.34 and 5.35 shows the shape of Received Constellation (Scattering plots) for 4, 16, 64 and 256 -QAM, and how the TI technique is effect on them.

4-QAM Modulation

Fig. 5.32 Received 4-QAM Constellation for TI

16-QAM Modulation



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64-QAM Modulation


256-QAM Modulation



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2) CCDF for TI

Fig. 5.36 shows the CCDF of PAPR for a 4,16,64 and 256-QAM/OFDM system without TI and with TI technique.

Fig. 5.36 CCDF for TI with different types of Modulation

Table 14 CCDF values for TI with different types of Modulation

Modulation Types PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No TI ) 10 4-QAM 8 16-QAM 8.5 64-QAM 8.5 256-QAM 9 As shown in figure 5.36 : It is seen that the PAPR performance improves as the number of Modulation type decrease with M = 4, 16, 64 and 256. The Basic OFDM curve has PAPR equals to 10 dB. After applying TI technique , the value was significantly reduced to 8 dB for 4-QAM and 8.5 for 16-QAM and 64-QAM and 9 dB using 256-QAM.


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5.3.5 Interleaving The basic idea in adaptive interleaving is to set up an initial terminating threshold. PAPR value goes below the threshold rather than seeking each interleaved sequences. The minimal threshold will compel the adaptive interleaving (AI) to look for all the interleaved sequences. The main important of the scheme is that it is less complex than the PTS technique but obtains comparable result. This method does not give the assurance result for PAPR reduction. In this circumstance, higher order error correction method could be used in addition to this method. 5.3.5.1 Block Diagram

Fig. 5.37 Interleaving General Block Diagram

Fig. 5.38 Interleaving Block Diagram using Labview


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5.3.5.2 Interleaving Results 1) Received Constellation

Fig. 5.39, 5.40 and 5.41 shows the shape of Received Constellation (Scattering plots) for 4, 16 and 64-QAM, and how the Interleaving technique is effect on them.

4-QAM Modulation

Fig. 5.39 Received 4-QAM Constellation for Interleaving

16-QAM Modulation



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64-QAM Modulation


2) CCDF for Interleaving

Fig. 5.42 shows the CCDF of PAPR for a 4,16 and 64-QAM/OFDM system without Interleaving and with Interleaving technique.

Fig. 5.42 CCDF for Interleaving with different types of Modulation


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Table 15 CCDF values for Interleaving with different types of Modulation

Modulation Types PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No Interleaving ) 10 4-QAM 6 16-QAM 6.5 64-QAM 6.5 As shown in figure 5.42 : It is seen that the PAPR performance improves as the number of Modulation type decrease with M = 4, 16, 64 and 256. The Basic OFDM curve has PAPR equals to 10 dB. After applying Interleaving technique , the value was significantly reduced to 6 dB for 4-QAM , 6.5 dB for 16-QAM and 64-QAM. 5.3.6 Amplitude Clipping and Filtering (ACF) One of the simple and effective PAPR reduction techniques is clipping, which cancels the signal components that exceed some unchanging amplitude called clip level. However, clipping yields distortion power, which called clipping noise, and expands the transmitted signal spectrum, which causes interfering. Clipping is nonlinear process and causes in-band noise distortion, which causes degradation in the performance of bit error rate (BER) and out-of-band noise, which decreases the spectral efficiency. Clipping and filtering technique is effective in removing components of the expanded spectrum. Although filtering can decrease the spectrum growth, filtering after clipping can reduce the out-of-band radiation, but may also cause some peak re-growth, which the peak signal exceeds in the clip level. The technique of iterative clipping and filtering reduces the PAPR without spectrum expansion. However, the iterative signal takes long time and it will increase the computational complexity of an OFDM transmitter. The amplitude clipper limits the peak of the envelope of the input OFDM signal to a predetermined threshold value.


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5.3.6.1 Block Diagram

Fig. 5.43 ACF General Block Diagram

Fig. 5.44 ACF Block Diagram using Labview

5.3.6.2 ACF Equations

for n = 0 , ...., NL - 1 - 3pt

where ; that is the phase of c[n] is out of phase with by 180o , and A is the clipping level.

where is the threshold.

where is the clipped signal can be obtained by adding a time-shifted and scaled signal to the original signal .


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5.3.6.3 ACF Results 1) Received Constellation

Fig. 5.45 and 5.46 shows the shape of Received Constellation (Scattering plots) for 4 and 16-QAM, and how the ACF technique is effect on them (Distortion).

4-QAM Modulation

Fig. 5.45 Received 4-QAM Constellation for ACF

16-QAM Modulation

Fig. 5.46 Received 16-QAM Constellation for ACF


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2) CCDF for ACF Fig. 5.47 shows the CCDF of PAPR for a 4 and 16-QAM/OFDM system without ACF and with ACF technique.

Fig. 5.47 CCDF for ACF with different types of Modulation

Table 16 CCDF values for ACF with different types of Modulation

Modulation Types PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No ACF ) 10 4-QAM 4.5 16-QAM 5 As shown in figure 5.47 : It is seen that the PAPR performance improves as the number Modulation type decrease with M = 4 and 16. The Basic OFDM curve has PAPR equals to 10 dB. After applying ACF technique , the value was significantly reduced to 4.5 dB for 4-QAM and 5 dB for 16-QAM.


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5.4 The Implementation Results for PTS technique using USRP SDR refers to the technology wherein software modules running on a generic hardware platform are used to implement radio functions. Combine the NI USRP hardware with LabVIEW software for the flexibility and functionality to deliver a platform for rapid prototyping involving physical layer design, wireless signal record & playback, signal intelligence, algorithm validation, Implementation was performed to test simulations result through a real wireless communication link using two Universal Software Radio Peripheral (USRP 29-20) one as transmitter and the other as a receiver. [13-14]

Fig. 5.48 Hardware Component


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5.4.1 The System Parameters Used for Implementation The parameter used for calculation of PAPR are illustrated in following Table:

Table 17 The System Parameters used for Implementation

Parameters Value used Data Random bits Symbol rate 5 Msps Modulation schemes 4,16,64 QAM Number of subcarrier 64 Oversampling factor 4 Length of cyclic prefix 8 bit Number of sub-blocks used in PTS methods (V)

4,16,32,64

Packet header/tail (8bit) IEEE 802.11a Short Training Channel encoding (3,1) repetition code Pulse shaping filter Raised cosine, α=0.5 TX average power level 0 dBm

RX reference level -20 dBm


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5.4.2 Implementation Results of PTS using USRP

Implementation Results Using 4-QAM Fig. 5.49 shows The PAPR characteristics of the PTS-OFDM signals which includes the distributions of the imaginary and real parts using various V for N=64 and 4-QAM.

Fig. 5.49 Received signal for various V=4,16,32,64 using 4-QAM


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Fig. 5.50 shows the PTS-OFDM signals at time domain using 4-QAM modulated subcarrier for N=64.

Fig. 5.50 Received signal in time domain for various V = 4,16,32,64 using 4-QAM


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Sub-Blocks (V) PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No Sub-Blocks ) 10.5 Proposed V = 4 7.5 Proposed V = 16 5.5 Proposed V = 32 4 Proposed V = 64 3.5


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Implementation Results Using 16-QAM Fig. 5.52 shows The PAPR characteristics of the PTS-OFDM signals which includes the distributions of the imaginary and real parts using various V for N=64 and 16-QAM .



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Fig. 5.53 Received signal in time domain for various V=4,16,32,64 using 16-QAM


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Sub-Blocks (V) PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No Sub-Blocks ) 11 Proposed V = 4 6 Proposed V = 16 5.5 Proposed V = 32 4.5 Proposed V = 64 4


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Implementation Results Using 64-QAM Fig. 5.55 shows The PAPR characteristics of the PTS-OFDM signals which includes the distributions of the imaginary and real parts using various V for N=64 and 64-QAM.



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Fig. 5.56 Received signal in time domain for various V=4,16,32,64 using 64-QAM


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Sub-Blocks (V) PAPR ( 10-4 PAPR level ) [dB] Basic OFDM ( No Sub-Blocks ) 11 Proposed V = 4 6.5 Proposed V = 16 5 Proposed V = 32 4.5 Proposed V = 64 4 As shown in figure 5.51, 5.54 and 5.57 : It is seen that the PAPR performance improves as the number of sub blocks increases with V = 4, 16, 32, and 64. The Basic OFDM curve has PAPR equals to 10.5 dB .After applying PTS technique , the value was significantly reduced to 3.5 dB using V=64 for 4-QAM , 4 dB using V=64 for 16 QAM and 4 dB using V=64 for 64-QAM.


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This proves that PTS gives better results which is superior performance in PAPR reduction. It is seen that the PAPR performance improves as the number o sub blocks increases with V = 4, 16, 32, and 64. Implementation results show that there is 0.5 - 1.5 dB offset in CCDF curves among simulation and implementation due to USRP performance.


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Conclusion The high PAPR (peak-to-average power ratio) is considered to be one of the major drawbacks of OFDM (orthogonal frequency division multiplexing) because the power efficiency decreases because of the large signal fluctuation. All the potential benefits of OFDM transmission are reduced by high PAPR value. The purpose of this thesis was to reduce the High (PAPR) of OFDM signals. The comprehensive research and comparison are put forward for a variety of currently promising PAPR reduction methods quoted in the literature in this research area. This thesis presented:

Selecting Level Mapping (SLM). Partial transmit sequence (PTS). Active Constellation Extension (ACE). Tone Injection (TI). Interleaving. Amplitude Clipping and Filtering (ACF).

These methods are used to reduce the PAPR of OFDM signals and these were successfully achieved using LABVIEW Simulation then Implement the technique with the best PAPR reduction result which is Partial transmit sequence using USRP. Simulation shows that :

PAPR for the original OFDM is 10-12 dB for 64-QAM. Using SLM technique PAPR reduced to 6.5 dB for 64-QAM with 8 parallel-block. Using PTS technique PAPR reduced to 3.5 dB for 64-QAM with 64 sub-block. Using ACE technique reduced to 7.5 dB for 64-QAM. Using TI technique reduced to 6.5 dB for 64-QAM. Using Interleaving technique reduced to 6 dB for 4-QAM. Using ACF technique reduced to 5 dB for 4-QAM.

Simulation results show that PTS technique gives the best PAPR reduction. PAPR result for PTS implementation on USRP is 4 dB. Implementation results show that there is 0.5 - 1.5 dB offset in CCDF curves among simulation and implementation due to USRP performance. .


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Future Work

1. Use another type of filter that does not effect on the PAPR or have little impact with clipping.

2. Find a new type of companding to reduce the PAPR with maintaining the BER performance.

3. The proposed companding PAPR reduction methods can be combined with Different PAPR reduction techniques such as PTS, SLM, TR and etc.

4. The proposed RFC can be combined with different PAPR reduction techniques such as coding, interleaving, TI and DSI etc.

5. The RCF, proposed RFC can be combined with different existing companding techniques such as airy companding, linear companding, Trapezoidal power companding and etc.

6. The proposed companding PAPR reduction methods can be combined with Zadoff-Chu matrix Transform precoding.

7. Analysis of the proposed techniques and find out its impact on the PAPR mathematically.

8. proposed new hybrid techniques by using the proposed method. 9. Study the impact of these proposed techniques on bandwidth, noise , distortion and

the ratio of power saving. 10. Study the impact of these proposed techniques on statistical distribution. 11. The proposed PAPR reduction methods can be used with MIMO OFDM. 12. The proposed PAPR reduction methods can be used with other multicarrier System.


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DISSEMINATION Mamdouh Gouda, Mohamed Hussien, Mohamed Rabie, Ahmed M.Anwar, Ahmed Gouda. “USRP Implementation of PTS Technique for PAPR Reduction in OFDM using LABVIEW”, IJSER International Journal of Scientific and Engineering Research 2016, Misr University for Science and Technology, Egypt, February 26, 2016 ISSN 2229-5518. (Accepted and Pending publication),

Reference

Reference

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Reference

[1] "Multi- Carrier Techniques for Broadband Wireless Communications A Signal Processing Perspective" Man-On Pun Princeton University, USA Michele Morelli C-C Jay Kuo University of Pisa, Italy University of Southern California, USA. [2] Mudit Ratana Bhalla " Generations of Mobile Wireless Technology , International Journal of Computer Applications (0975 – 8887) Volume 5– No.4, August2010. [3] Andrea,Goldsmith, “ Wireless Communications”, Cambridge University,2005, ISBN 978-0-521-83716-3 [4] E. Dahlman, S. Parkvall, J. Sköld, and P. Beming. 3G Evolution: HSPA and LTE for Mobile Broadband. Academic Press, July 2007 [5] A. Goel “Improved PAPR Reduction in OFDM Systems,” Ph.D. dissertation, JAYPEE Institute of Information Technology, India, April 2013. [6] R. V. Nee and R. Prasad, OFDM for Wireless Multimedia Communication. London: Artech house Publisher, 2000. [7] H. D. Joshi. “Performance augmentation of OFDM system,” Ph.D. dissertation, Jaypee Univ. of engineering and Technology, India, May 2012. [8] S. Khalid “Peak to Average Power Ratio Reduction in Orthogonal Frequency Division Multiplexing Systems.” Ph.D. dissertation, University of Engineering and Technology Taxila, Pakistan, Dec. 2009. [9] V. Vijayarangan, and R. Sukanesh “An Overview Of Techniques For Reducing Peak To Average Power Ratio And Its Selection Criteria For Orthogonal Frequency Division Multiplexing Radio Systems,” Journal of Theoretical and Applied Information Technology, vol. 5, no. 1, pp.25-36, January 2009. [10] Sanjeev Saini, and O.P. Sahu “Peak to Average Power Ratio Reduction in OFDM System by Clipping and Filtering,” International Journal of Electronics Communication and Computer Technology, vol. 2, pp.105-109, May 2012. [11] A.Tiwari, and K. Markam “An Overview: Peak-to-Average Power Ratio Reduction Techniques for OFDM Signals,” International Journal of Computer & Communication Engineering Research, vol. 2, pp.23-28, January 2014. [12] T. Jiang, and Y. Wu, “An Overview: Peak-to-Average Power Ratio Reduction Techniques for OFDM Signals,” IEEE Transactions on Broadcasting, vol. 54, no. 2, pp.257-268, June 2008. [13] National Instruments, “ Getting Started with LabVIEW,” http://www.ni.com/pdf/manuals/373427j.pdf. 2013.

Reference

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[14] National Instruments, “ NI USRP Hardware: Getting Started Guide,” http://www.ni.com/pdf/manuals/375717e.pdf. 2013. [15] Zainab Saad Hadi AL-Hashmi, An Overview: Peak to Average Power Ratio (PAPR) in OFDM system using some new PAPR techniques (with matlab code). Baghdad:jorden., pp. 57-59, 2015. [16] H. Taub, D. L. Schilling, G. Saha, “Taub’s Principles of Communication Systems”: Tata McGraw Hill, 2008. [17] T. S. Rappaport, “Wireless Communication: Principles and Practice”: 2nd Edition, Prentice Hall, 2002. [18] A. Sahu , and S. Behera “PAPR analysis and channel estimation techniques for 3GPP LTE system.” National Institute of Technology Rourkela , 2011 [19] A. Goel “Improved PAPR Reduction in OFDM Systems,” Ph.D. dissertation, JAYPEE Institute of Information Technology, India, April 2013. [20] V.R.Prakash, and P.Kumaraguru “Performance analysis of OFDM with QPSK using AWGN and Rayleigh Fading Channel,” Conference: ICIESP, 2012, pp.10 -15 [21] Shieh, William; Djordjevic, Ivan, “OFDM for Optical Communications”, Academic Press, Burlington, 2009,ISBN= 8790080952062.

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