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IGBT GATE-DRIVE WITH PCB ROGOWSKI COIL FOR IMPROVED SHORT CIRCUIT DETECTION AND CURRENT TURN-OFF CAPABILITY

D. Gerber, T. Guillod, and J. Biela

Laboratory for High Power Electronic Systems ETH Zurich, Physikstrasse 3, CH-8092 Zurich, Switzerland

Email: gerberdo@ethz.ch

IGBT GATE-DRIVE WITH PCB ROGOWSKI COIL FOR IMPROVEDSHORT CIRCUIT DETECTION AND CURRENT TURN-OFF CAPABILITY

D. Gerber, T. Guillod, and J. BielaLaboratory for High Power Electronic Systems

ETH Zurich, Physikstrasse 3, CH-8092 Zurich, SwitzerlandEmail: gerberdo@ethz.ch

Abstract

In this paper, a gate drive using gate boostingand double-stage turn off including voltage clamping aswell as with detection of overcurrent and a too high di/dtduring turn on is discussed in detail. Besides the gatedrive, also the design of a PCB-Rogowski coil, whichis used for measuring currents and for di/dt detection,is explained and different designs are compared. Thepresented coil has a bandwidth of more than 28 MHz anda propagation delay of 11 ns.

I. INTRODUCTION

Currently, a compact and cost effective X-ray freeelectron laser facility (SwissFEL) is planed at the PaulScherrer Institute (PSI) in Switzerland. For this lasersystem, high power modulators meeting the specificationsgiven in table I are required. There, an important featureof the modulator is the pulse repetition accuracy, whichmust be better than 10−5.

To fulfill these challenging requirements, a solid statemodulator based on a matrix (split core) transformer withsix cores as shown in Fig. 1a) is designed. In the modula-tor in total twelve 4.5 kV Press-Pack IGBTs manufacturedby ABB are used and each IGBT is connected to a separateprimary winding and always two primary windings arearound one core. Due to the matrix transformer, thecurrent balancing between the IGBTs on different coresis inherently provided as explained in [1], so that onlythe current balancing between the two IGBTs connectedto windings on the same core must be guaranteed by thegate drive circuit/control.

As the considered pulse length is relatively short, theswitches are operated at pulse currents much higher thanthe rated switch current, in order to minimize the numberof required IGBTs and fully utilize the semiconductors.This operation condition is not very critical with respect tothermal issues and/or thermal cycling due to the relativelylarge thermal capacitances and short pulses, but with thehigh pulse currents, the margin between currents duringnormal operating and maximum switch current (latchingcurrent [2]) becomes much smaller. Therefore, the current

TABLE ISPECIFICATIONS FOR THE SOLID STATE MODULATOR (VIn=3 KV).

Output Voltage 370 kVOutput Power 120 MW

Repetition Rate 100 HzRepetition Accuracy 10−5

Bouncer

Pulse GeneratorPulse Transformer

Load

Main Switch

Reset-Circuit

(a)

(b)

Fig. 1. a) Schematic of the solid state modulator with: generator circuit(incl. active reset), pulse transformer and bouncer circuit. b) Photo of onegenerator circuit consisting of 3 energy storage capacitors, Press-PackIGBT switch and active reset circuit.

through the switch could relatively quickly rise to levels,where the IGBT could not be turned off any more in caseof fault conditions or arcing of the klystron load.

A method to avoid this problem is to apply gate volt-ages, which are just large enough, so that the IGBT couldonly conduct the nominal current, but enters the linearoperation in case of an overcurrent. However, with thesmall gate voltage, the turn on behavior of the IGBT isslow, limiting the rise time of the pulse during turn on.

In order to limit the short circuit currents also in case ofhigh gate voltages, a fast and reliable overcurrent / shortcircuit detection is required. In [3], a di/dt short circuitdetection utilizing the parasitic inductance of the kelvincontact has been investigated and the reliability of themethod has been demonstrated.

Unfortunately the considered 4.5 kV press-pack IGBTsdo not provide a kelvin contact, so that in this papera di/dt short circuit detection method based on PCB-Rogowski coils is proposed. Such Rogowski coils offer ahigh bandwidth and can be used up to very high currentsdue to the lack of a ferromagnetic core which can saturate.Furthermore, the fabrication of a PCB-coil is simple andcheap and it is easy to integrate it in a press-pack stack.In the proposed system, each gate drive is able to detectovercurrents in both IGBTs connected to windings on thesame transformer core, in order to reduce the turn off timeof the two IGBTs.

In order to minimize the turn on time, a high gatevoltage is used at the beginning of the pulse, which isreduced after the IGBT is fully turned on. Additionally, thegate drive provides a 2-stage turn off and an overvoltageclamping [4] in order to reduce the pulse fall time duringturn off and lower the switching losses.

In the following, first, the gate drive circuit and the shortcircuit detection are described and thereafter, the designof the coil is presented in detail in section III.

II. GATE DRIVE

In the considered solid state modulator, a matrix (splitcore) transformer, which consists of six cores is applied.The two parallel connected secondary windings both en-close all six cores. Furthermore, each core carries twoprimary windings and only one IGBT is connected to aprimary winding. Consequently, there are twelve primarywindings and twelve IGBTs in the considered 120 MWmodulator. As explained in [1], the current distributionbetween windings mounted on different cores is inherentlyprovided by the matrix transformer. However, the currentbalancing between the two switches/primary windings onthe same core must be controlled by the gate drive/controlcircuit. This could be for example achieved by schedulingthe gate pulses as described in [4].

As already mentioned, for achieving a fast rise/shortpulse rise time, a high gate-emitter voltage is required dur-ing turn on. After the IGBT is turned on, the gate voltagecould be reduced to limit the maximal current throughthe IGBT before the IGBT enters the linear operationmode. Such a technique is known as gate boosting andis for example described in [5]. While the gate voltageis boosted, a too high current could flow through theIGBT in case of a short circuit, resulting in a latch-upof the parasitic thyristor in the IGBTs pnpn-structure. Ifthe parasitic thyristor is ignited by a too high current, theIGBT could not be turned off via the IGBT-gate and willbe destroyed in case of a short circuit. Such a situationmust be avoided by a fast short circuit detection and a fastturn off of the IGBT.

In case of a short circuit, turning off of one of twoIGBTs on the same core causes the current to commutateto the other IGBT on the same core. Since the pulsecurrents are high compared to the nominal current, thiscould lead to latching and/or destruction of the IGBT. Adetection and synchronous turn off via the main controlwould be relatively slow due to the optical link, which isused for transmitting the control signals.

Therefore, both IGBTs must be turned off at the same

time. For achieving a short delay time between short cir-cuit detection and IGBT turn off, the presented gate drivecircuit is capable to detect a short circuit in both IGBTsat the same time as. The implementation is relativelysimple because the current measurement with a Rogowskicoil is floating and two Rogowski coils providing twoindependent current measurements can be implementedon the same PCB.

A further feature implemented in the discussed gate-drive is a double-stage turn off circuitry with activeclamping as described in [6] and as could be also seen inthe block schematic of the gate drive given in Fig. 2. Inthe following, the operating blocks are shortly discussed.

a) Turn On Stage: The basic idea of the turn on circuitis using a higher gate voltage during turn on. To do so,two supply voltages are used. Because the gate voltagehas to be reduced after turn on, an additional switch isrequired to discharge the gate to the lower voltage railwhen the IGBT is turned on.

b) Turn Off Stage: At the beginning of the turn offaction, a low gate resistance is used to achieve fastswitching speeds. Due to parasitic inductances in thepower circuit, an overvoltage occurs across the IGBTduring turn off. To limit this overvoltage, the switchingspeed is reduced by using a higher gate resistance as soonas the voltage reaches a certain limit. This is done byusing two gate resistors – a high value resistor, which isused continuously during turn off, and a low value resistor,which is connected in parallel only at the beginning ofthe turn off action (cf. also [6]). Additionally, an activeclamping circuit is used to limit the over voltage at a fixlevel by slightly turning the IGBT on again (linear mode).

c) Status Detection: To properly control turn on andturn off process, the status of the IGBT, i.e. if it is fullyturned on or off, must be known. For detecting the status,the value of the DC link voltage, which could be variedduring operation to change the output power, must beknown. This detection is performed by a simple track andhold stage (THA) and two comparators, which track theDC link voltage VCE,m before the main IGBT turns onand compares the actual IGBT voltage with VCE,m.

With the detection three control signals are generated:Signal 1 indicates if the IGBT is turned off, signal 2indicates if the switch is turned on and signal 3 is used todetermine if the low or the high gate resistance is requiredduring turn off. The references for the first two signalsare adjusted from pulse to pulse by the track and holdstage. This is not necessary for the third signal because theovervoltage limit is independent of the DC link voltage.

d) Integrator & Short Circuit Detection: Since theRogowski coil only measures the di/dt, an integratorcircuit is required for obtaining the current amplitude. Asthe analog integrator circuit is sensitive to DC-offsets, thepresented gate drive uses an active offset compensation,which uses an LF analog PI-controller that feeds theintegrators output voltage back to its input (not shownin Fig. 2). Alternatively, the integrator could be reset atthe beginning of each pulse.

To detect short circuits two comparators are used toindicate an overcurrent and two comparators to indicate atoo high di/dt value.

Status Detection

Trigger for Double-Stage Turn-OFF

Vlim,I2

Vlim,dI2/dt

Integrator Integrator

Short-Circuit Detection

Turn On Stage

Active Clamping

Turn Off Stage

C-E Voltage Measurement

Vgate,1 Vgate,2

Ron,1 Ron,2

Roff,1 Roff,2

Vref,off

Vce

VCE,m

Rfb

C

E

GFPGA

Vlim,I1

Vlim,dI1/dt

VCoil1 VCoil2

VCE,m

VCE,mTHA

Fig. 2. Block schematic of the gate drive circuit where the functional blocks are summarized in the grey highlighted areas.

e) Control Circuitry: For obtaining a flexible and com-pact gate drive, the control circuitry is implemented ina FPGA. There, also a CPLD could be used in order toreduce costs, which, however, limits the flexibility duringthe development of the code.

To avoid ground loops, the status signals as well as theswitching signal are transmitted via plastic fibre optic linksand the power supply is galvanically isolated.

A. Gate Drive Operation

When a switching signal is received, the track and holdamplifier is put into the hold mode. As soon as thisis done, the turn on process begins and the MOSFETconnecting the higher gate voltage to the IGBT gate isturned on. Because the n-channel MOSFET connectingthe lower voltage rail to the gate has a diode in series, itis turned on at the same time, in order to reduce the risetime of the gate voltage.

As soon as the IGBT collector-emitter voltage hasdropped, the higher voltage rail is disconnected and aswitch is closed to discharge the IGBT gate capacitor tothe level of the lower gate voltage.

For safety, there is also a time-out circuit implementedon the FPGA which reduces the gate voltage after a certainamount of time in case the IGBT voltage does not drop

VGE

t

V

VCE

Vgate,1

Vgate,2

VCE or Time Controlled

t

V, R

VCE

Roff

Turn On Turn Off

Fig. 3. Gate drive operation during turn on and turn off.

due to a failure.When the IGBT is turned off, both turn on voltage

rails are disconnected from the gate and both turn offresistors are connected to reduce the gate voltage rapidly.As soon as the collector-emitter voltage reaches a givenlevel, one turn off resistor is disconnected. This reducesthe switching speed and therefore also the overvoltagecaused by parasitic inductances. Finally, the track and holdamplifier is set into the track state as soon as the switchis turned off.

During the switching action, the current status is trans-mitted to the main control unit. This can be used tosynchronize the edges and in case the higher gate voltage

Fig. 4. Photo of the gate drive.

VGE

IC

VCE

Fig. 5. Voltage and current waveforms for a 3µs pulse. The scale forVCE is 100 V per division, for IC 200 A per division and for VGE itis 5 V per division. The time scale is 500 ns per division.

can be adjusted by the main control also to synchronizethe switching speeds during turn on.

In Fig. 5, measured waveforms of the gate drive areshown nicely matching with the theoretical curves.

III. ROGOWSKI COIL

For detecting a short circuit, a current measurementsystem based on PCB-Rogowski coils is used. With thissystem not only the amplitude of the current could bemeasured, but also the di/dt of the current enablinga faster short circuit detection. The details of the coildesign, measurement accuracy and simulation results arepresented in the following.

A. Coil Design

The operation principle of a Rogowski coil is based oninduction, i.e. a flux variation induces a voltage in theturns of the Rogowski coil. The induced voltage at thecoil terminals is:

Vind = −N∑i=1

dφndt

= −M · didt

(1)

where N is the number of turns and M the mutualinductance between the loop conducting the current to bemeasured and the Rogowski coil.

Assuming that the magnetic field is constant inside thearea enclosed by the turns of the Rogowski coil, themutual inductance is given by

M =N

l·A · µ0, (2)

where l is the length of the coil and A the area enclosedby a single turn. This equation is only valid for currentsflowing in the middle of the coil. For currents not flowingin the middle, the mutual inductance has to be calculatedwith a more complex method which is described later.

This assumption is valid because the width of the areaenclosed by a single turn is small compared to the distanceof the current to be measured.

In the considered gate drive, two separate coils are usedfor a faster short circuit detection as described above. Inorder to obtain the same measurement result for both ofthe Rogowski coils, which are realized side by side on thesame PCB, the mutual inductance for both coils must bethe same. Considering the equation above this means that

N1

l1=N2

l2(3)

must be fulfilled if the area is the same for both coils.There, the number of turns is mainly determined by the

desired measurement accuracy and bandwidth. In the con-sidered case, the outer coil on the PCB has 146 windings,the inner coil has 126 windings. This corresponds to 2.43and 2.41 turns per cm, which results in only a deviationof the mutual inductance of 0.47 %.

The cross-sectional area of the turns is 11.2 mm2, sothat a mutual inductance of 3.4 nH results for both coils.

M didt

Lcoil

Ccoil Rd Vcoil

Fig. 6. Electrical model of a Rogowski coil valid up to the firstresonance frequency. The distributed inductance and capacitances aresummarized in two lumped components. Rd represents an additionaldamping resistor.

0.1 0.12 0.14 0.16 0.18 0.2 0.22 0.24 0.26 0.28−1

−0.5

0

0.5

1

1.5

2

Time (μs)

ζ = 0.1ζ = 0.5

ζ = 0.7

ζ = 1 M didt

Volta

ge (V

)

Fig. 7. Coil response on a square wave input voltage for differentdamping.

1) Coil Bandwidth

Besides the self-inductance each turn of the Rogowskicoil has, there are also parasitic capacitances betweenthe turns. These can be modeled by an equivalent circuitconsisting of many Ls and Cs, which result in a highnumber of resonances. However, in normal operation theRogowski coil is only used at frequencies below the firstresonance frequency, so that the electrical model of aRogowski coil could be simplified to a simple LC networkas shown in Fig. 6.

Based on the equivalent circuit given in Fig. 6, thesecond order transfer function of the coil up to the firstresonance frequency is given by:

Vout

M didt

=1

s2LC + s LRd + 1

=ω0

s2 + 2ζω0s+ ω20

(4)

The resonance frequencies of the two designed coils– the inner and the outer one on the PCB as shown inFig. 11 – are 33 MHz and 28 MHz. With these values, thetime domain response of the measurement system couldbe determined. In Fig. 7, the time domain response of theouter coil with a resonance frequency of 28 MHz for arectangular input is shown.

As could be seen in Fig. 7, the coil response stronglydepends on the damping ζ. Because the coil should bealso used for a di/dt short circuit detection, no overshootis allowed in order to avoid false short circuit detections.Therefore, the damping resistor Rd has to be selectedproperly to achieve ζ = 1.

The delay for a ramp voltage, which results due to thefinite resonance frequency fres, could be calculated by:

td =2ζ

2πfres. (5)

−15 −10 −5 0 5 10 15

−10

−5

0

5

10

(cm)

(cm)

0.5

1

1.5

2

2.5

3

3.5

e>3%

e>2%

Error e (%)

Fig. 8. Measurement error of the inner PCB Rogowski coil withconstant winding density. The given error is for perpendicular linecurrents flowing at the considered location of the display error.

For ζ = 1 and fres = 28 MHz this results in a time delayof 11 ns, which shows that the coils can be used to detecta short circuit very fast.

B. Measurement Accuracy

Besides the bandwidth, also the measurement accuracyof the coil is an important issue. There, two factorsinfluence the accuracy: first if the current, which shouldbe measured, is not exactly flowing in the middle of theRogowski coil, and second, other currents flowing in thevicinity of the Rogowski coil, which also could result inan induced voltage/measurement signal.

For determining the achievable accuracy for measuredcurrents not flowing in the middle of the Rogowski coil,first the mutual inductance between each turn of theRogowski coil and the loop, where the measured currentis flowing, is calculated for each turn for the consideredlocation of the measured current as described in [7]. Thetotal mutual inductance is obtained by adding up thevalues for the individual turns.

By comparing the calculated mutual inductance to thevalue for the ideal mutual inductance, which is obtained,when the measured current is flowing in the center ofthe Rogowski coil, the measurement accuracy is obtained.The ideal value for the considered coil given in Fig. 11 is3.4 nH.

The same method could be applied to determine theinfluence of currents flowing outside the coil, whichshould not result in a measurement signal, i.e. the idealmutual coupling for these currents is 0 nH.

In Fig. 8 the achievable accuracy for measured currentsflowing at different positions through the coil are given.The shown error value is for an ideal line current, whichis flowing perpendicular through the coil at the consideredposition.

In the figure it can be seen that the measurement errorfor currents flowing close to the corners of the coilsis relatively high. By shifting turns to the corners andtherefore increasing the number of turns in the cornerswith a constant total number of turns, the accuracy canbe improved significantly as shown in Fig. 9. To increasethe measurement accuracy further, more windings wouldbe required. However, this would reduce the bandwidth.

In a next step, the measurement accuracy of currentsflowing not perpendicular through or outside the coilis investigated. Unfortunately, this could not easily per-

−15 −10 −5 0 5 10 15

−10

−5

0

5

10

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

0.45

0.5

e<0.5%

e<0.5%

(cm)

(cm) Error e (%)

Increased Winding Density

Fig. 9. Measurement error of the inner coil with increased windingdensity in the corners.

formed by analytical calculations. Therefore, 3-D PEECsimulations are performed, which are discussed in thefollowing section.

C. Coil Simulation

For non-perpendicular currents, the induced volt-age/signal can be relatively large since the Rogowski coilhas one big turn in addition to the small turns, whichresults due to the large loop the Rogowski coil enclosesbetween the start and the end point (cf. ALoop in Fig. 11).

For coil (1) in table II, the total area enclosed by all turnsis 1635 mm2 and the area ALoop is 25230 mm2 resulting ina high sensitivity to noise generated by non-perpendicularcurrents.

For avoiding these disturbances, an additional returnconductor from the end of the coil to the start, so that bothconnection points of the coil are on the same Rogowskicoil point, can be used. An example is coil (2) in table II,where the area enclosed between the starting and the endpoint of the Rogowski coil is reduced to 73 mm2.

The second method is by continuing the turns at the endof the coil back to the starting point along the same pathinstead of just using a single wire. An example is coil (3)in table II, which has an area of approximately 1400 mm2

between the starting and the end point of the Rogowskicoil.

In table III the mutual inductance for the three differentdesigns for different locations of flowing currents aregiven (yellow boxes and red arrows). In row 1 the idealsituation with the ideal mutual inductance is given. Forcurrents flowing outside the Rogowski coil, the mutualinductance should be 0 as mentioned above. These valueshave been obtained with GeckoEMC, which uses thePEEC method.

The coil without compensation shows the best frequencyperformance, but the noise immunity is very poor. Withthe compensation, the bandwidth reduces to approximately30 MHz, which is relatively independent of the compen-sation method. Since the noise immunity of the coil (2)is slightly better, this geometry is used for the Rogowskicoil.

In Fig. 10, the real test setup is shown. As could beseen, the current is not flowing only through the middleof the coil, but also on the outside in order to achieve alow inductive setup for the IGBT. Therefore, the mutualinductance in the test setup is a combination of theinvestigated cases in table III. This leads to an increased

TABLE IIDIFFERENT WINDING TOPOLOGIES FOR THE ROGOWSKI COIL: (1)

WITHOUT COMPENSATION, (2) WITH A SINGLE RETURN CONDUCTORAND (3) WITH TURNS FROM THE STARTING POINT TO THE END AND

BACK.1 2 3

TABLE IIICOMPARISON OF DIFFERENT COMPENSATION METHODS.

Windingtopology

1 2 3

L (µH) 2.369 2.105 1.849

fres (MHz) 50.0 29.95 31.3

1 3.450 nH 3.453 nH 3.359 nH

2

10.6

0.104 nH 0.103 nH 0.107 nH

3 13.5 0.092 nH 0.094 nH 0.089 nH

413.5

22.940 nH 0.087 nH 1.166 nH

5

10.6

27.795 nH 0.497 nH 2.213 nH

6 0.0017 nH 0.0104 nH 0.0728 nH

7 1.193 nH 1.192 nH 1.191 nH

total mutual inductance in the real circuit which is around32 % higher than the ideal inductance. A picture of thePCB-Rogowski coil is shown in Fig. 11.

IV. CONCLUSION

In this paper, a gate driver circuit designed for 4.5 kVpress-pack IGBTs used in a 120 MW/370 kV solid statemodulator is presented. The gate drive implements gateboosting for achieving faster turn on and a double-stageturn off with additional voltage clamping to minimize turnof time and limit the occurring overvoltage.

For a fast over-current detection, which uses not only

CapacitorLoad

+

++

+

−−

−−

IGBTEmitter

Collector

Fig. 10. 3D model used for the simulation of the IGBT test setup.

Fig. 11. Picture of the presented Rogowski coil.

the current amplitude but also the di/dt for detection,PCB-Rogowski coils are used that could be easily inte-grated in the press-stack. The design of these Rogowskicoil is investigated in detail in the paper by analyticalcalculations and by 3D simulations based on the PEECmethod. Furthermore, different designs for the coil arecompared. There, a bandwidth of more than 28 MHz anda propagation delay of 11 ns has been achieved for thecoil.

ACKNOWLEDGMENT

The Authors would like to acknowledge the supportof ABB Semiconductor, who provided IGBTs for theexperimental system, and the strong support of PPT inrelation with the practical realization of the project.

References

[1] D. Bortis, J. Biela and J.W. Kolar, ”Transient behaviour of solid statemodulators with Matrix Transformers”, 17th IEEE InternationalPulsed Power Conference (PPC), 2009, pp. 1396-1401.

[2] N. Mohan, T. M. Undeland and W. P. Robbins, ”Power Electronics”,Third Edition, John Wiley & Sons, United States of America, 2003,pp. 631-634

[3] M.N. Nguyen, R.L. Cassel, J.E. deLamare and G.C. Pappas, ”Gatedrive for high speed, high power IGBTs”, Digest of Technical PapersPulsed Power Plasma Science, 2001, pp. 1039 - 1042.

[4] D. Bortis, J. Biela and J.W. Kolar, ”Active Gate Control for CurrentBalancing in parallel connected IGBT Modules in Solid StateModulators”, 16th IEEE International Pulsed Power Conference(PPC), 2007, pp. 1323-1326.

[5] A. Volke and M. Hornkamp, ”IGBT Modules, Technologies, Driverand Application”, Infinieon Technologies AG, Munich, 2011, pp.226 - 227.

[6] D. Bortis, P. Steiner, J. Biela, and J.W. Kolar, ”Double-Stage GateDrive Circuit for Parallel Connected IGBT Modules”, Proceedingsof the IEEE International Power Modulators and High VoltageConference, 2008, pp. 388 - 391.

[7] N.Karrer, P. Hofer-Noser and D. Henrard, ”HOKA: A New IsolatedCurrent Measuring Principle and its Features”, Conference Recordof the IEEE Industry Applications Conference, 1999, pp. 2121 -2128.