Load-Resonant Converter with Changing Resonant Tanl< Topology for Welding Applications

Jevgeni Shklovski, Kuno Janson, Ants Kallaste Department of Electrical Engineering and Electrical Machines

Tallinn University of Technology, Tallinn, Estonia

jevgeni.sklovski@ttu.ee, kuno.janson@ttu.ee, ants.kallaste@ttu.ee

Abstract-In this paper a new approach and corresponding

converter based on varying resonant tank topology for supplying

DC loads like welding arc is presented. In order to describe

suggested topology the circuitry of proposed converter and some

design aspects are discussed in relation to typical load conditions

during welding process. Simulation and first experimental data

are also provided and analyzed. The obtained natural

characteristic of this converter has a constant power mode in the

range of welding currents and inherent parametrical short­

circuit limitation. Also, its fast "natural" response to change of

load conditions could probably yield in better arc stability and

more even weld bead quality.

Keywords: resonant converter; resonant tank topolog;, welding power supply

I. INTRODUCTION

Load-resonant converters are the class of resonant converters in which load forms a part of the resonant circuit [9]. Switch-mode resonant converters become more and more popular due to the soft-switching possibilities (ZVS and ZCS) and increased frequency of operation and, consequently, increased power density of converters.

The resonant converter proposed and discussed in the paper belongs to the class of switch-mode load-resonant DC­DC converters. As a specific feature, its resonant tank topology is parametrically changing and could form parallel and/or series resonance circuits. The topology and duration of each resonance circuit depends on load parameters and discussed more thoroughly in [10]. Its predecessor a resonant converter operating at mains frequency and known as PSA (Parallel and Series resonance Alternating) converter was successfully applied for supplying loads like DC arc furnaces in MW range [5]. The idea to adopt this converter for welding arc supply has yielded in Switch-Mode PSA converter (SM­PSA). Increasing of supply voltage frequency resulted in smaller weight and size of converter that were the main drawbacks at mains frequency.

Arc welding is a widely used process of metal joining by means of electric arc burning between electrode and object being welded. There are different arc welding processes characterized by current type, protective environment, level of automation. In given paper, we only consider manual arc welding (SMA W or MMA) widely used process in

978-1-4673-2421-2/12/$31.00 ©2012 IEEE 518

construction and repairs. Typically it is represented by DC current electric arc burning in the air between stick electrode and welded product. The voltage of such steady arc is in between 24 and 28 V (at 1007200 A current) and this voltage is slightly dependent on the current. However, electric arc is very unstable and deeply varying type of load that depends on various conditions.

Fluctuations in electric arc are followed by current and voltage that make supplied power to change chaotically too. Commonly, to provide a stable manual arc the welding current is kept constant. Thus, power sources for manual arc welding are mainly represented by regulated DC current sources. The typical current sources have a feedback circuit to compare real welding current with its reference value [2--4]. However, voltage of welding arc depends also on conditions in plasma cloud. As a result, the total thermal power of arc is fluctuating. Small mass of welding plasma cloud can cause oscillation of arc's resistance that reaches some hundreds of kilohertz. Thus, power supply unit and its feedback circuitry are operating in permanent transient mode. Usually, current stability is not the best in transient mode. Consequently, undefined fluctuations of thermal power exist and welding quality could suffer with this type of welding.

In a view of welding process efficiency it is desirable to provide weld pool with stable thermal power by mean of electric arc [I]. However, voltage fluctuations in electric arc lead to chaotically changing power of the arc. In the given research we have assumed that power source providing constant power to a welding bead and having relatively stable welding current could make welding process better. A modified SM-PSA converter having constant power range in output characteristic is suggested for this purpose and analyzed as a power supply for manual arc welding.

So far we have assumed that the thermal power of welding arc could be stabilized much better if it is supplied with constant-power source. The current supplied by a constant­power source is defined according to relation

i = P lu

where P - is the power supplied to welding arc; u - arc voltage; i- current supplied to welding arc.

(1)

Herein, load current changes together with load voltage in such way that product of voltage and current remains constant. The typical switch-mode power supply can be transformed into constant-power source if its control circuit is modified to operate in accordance with (I). However, slow feedback and control circuits could be obstacles here and a fast converter without feedback control could be some better option. Also, output current limitation is needed in the range close to short­circuit or in other words converter must tum into constant­current operation mode. These features are inherent to PSA converters family that is discussed further.

II. CIRCUIT CONFIGURATION

A. Mains Frequency PSA Converter

To give some backgrounds of the proposed converter a brief description of its origin, PSA converter, is given. This PSA converter operates at mains frequency (50 or 60 Hz) and includes capacitor C and inductor L 1 in resonant circuit (see in Fig. 1). The PSA converter has transformer T with tapped secondary winding. Resonant circuit components are placed between the secondary winding W2& W3 and uncontrolled three-phase full-bridge output rectifier VI-V6.

Uy

tliJ T

C

1L2 V1 a V4 V3 b V6 V5 c V2

+ Ud

Figure 1. Single-phase PSA converter basic topology

The most distinctive properties of such PSA converter are listed below

• Wide operating range from no-load to short-circuit without overloads

• High power factor in the whole operating range (THD:::;;O,I; power factor 0,99-1).

• Short transient times (2-3 half-cycles of supply voltage) and low transient overshoots.

• Output voltage is "parametrically adjusting" to the

load condition without feedback control. "Transition" from series to parallel resonance topology can

take place each half-cycle and depends on load condition. When the load voltage is low the interval of parallel operation increases, resulting in increasing of average turns ratio for one half-cycle. And it can be represented as a smooth variation of the transformer turns ratio. As compared to the parallel and series resonant converters, the PSA converter can form both types of resonant circuits: parallel and/or series, that occurs parametrically according to the load condition [5].

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However, if we compare both loads: welding arc and arc in furnace the voltage of welding arc is much lower that yields in larger resonant and reactive components in PSA converter.

B. Switch-Mode PSA Converter

To overcome this drawback a frequency increasing of supply voltage is usually applied, and, there is a corresponding Switch-Mode PSA converter suggested [6]. One of its simplest circuit embodiments is given in Fig. 2.

Due to the higher frequency of supply voltage the capacitance of capacitor C 1 and inductance of inductor L 1 become less and given topology is more applicable for welding purposes. To change the supply voltage frequency an inverter circuit is commonly used. In proposed converter a full-bridge inverter circuit is implemented due to relatively high power level of the converter (about 4 kW). Switches of inverter are transistors operating with fixed frequency and predefmed pulse width. Switching occurs in pairs Tl&T2 and T3&T4, resulting in rectangular AC output voltage. On the one hand it is a very simple control method on the other hand the supply voltage is far from sinusoidal as it was with mains frequency PSA converter. Such a distorted voltage can result in some specific behavior of the converter due to the presence of higher harmonics [6].

T1

T4

09 011 013

010 012 014

Figure 2. Switch-mode PSA converter base topology

C. Simulation o/Output Characteristics o{SM-PSA

The constructed output characteristics for load current and power represented in Fig. 3 are given for the circuit shown in Fig. 2. having output no-load voltage about 80 V and rated current about 150 A while gating pulse width is close to supply voltage half-period or D = 0.45.

These curves are build up with data obtained by computer simulation with PSIM software, wherein load is represented by reverse DC voltage source imitating the electric arc. By changing its voltage from 0 to 80 V the whole operation range from no-load to short-circuit was covered.

This range can be conditionally divided into three regions. The first one at output voltage 0-30 V, where converter behaves almost as a current-source and load current changes from 200 A at short-circuit to 135 A at 30 V. Then, it is the second part at 30-58 V. where SM-PSA converter resembles power-source having little power variation in the range between 4.3 kW-5.2 kW--4.1 kW. And finally, at 58-80 V load voltage it is to some extent similar to the voltage-source mode, where current changes from 70 A to zero.

Current source mode

I,A

200 .... IpSA

.... 150 Icanst = 160 A

100

50

a 20

Power source mode Voltage source mode

.... .... .... .... ....

40

"

60 UPSA. V 80

P, kW

5

4

3

2

Figure 3. Output current (IPSA) and power (PPSA) of Switch-Mode PSA converter depending on load voltage (UPSA) (computer simulation)

Curves shown in Fig.3 are natural output characteristics (without feedback and regulation) of Switch-Mode PSA converter topology embodiment given in Fig. 2 and it is, in principle, fits to supply electric arc. It has an inherent short­circuit current limitation and relatively constant current in welding range.

The computer simulation also proved that feedback circuitry replaced by the parametrical variation of resonant tanl<­topology can reduce current oscillations during transients. Resonant circuit is permanently "adopting" its topology to the load conditions and does it rapidly, in principle, up to six times of supply voltage frequency [5]. That corresponds well with high-performance welding arc power source requiring the fast alteration between different load modes [7]. Nevertheless, there are some specific and inherent to SM-PSA converter issues described further.

D. Developing Circuit Topology

First of all, the voltage range of welding arc for MMA welding is usually 24-28 V, while as it is seen in Fig. 3 the converter with circuit given in Fig, 2, operates as a current source, and, consequently, to move into power-source range we need to "shift" this curve. Additionally, welding arc of power source being developed has an output power about 4 k W that is lower than the maximal and in the same time rated power of simulated one shown in Fig. 3. The rated point of welding arc and rated power of converter could be shifted to one point by decreasing secondary voltage of transformer (about l.5 times). But this, in its turn, decreases no-load voltage to 53 V, while it is preferable to have higher no-load voltage for comfort arc ignition and stable operation with different types of electrodes [8].

This issue could be simply resolved by adding auxiliary voltage source to the resonant circuitry (additional secondary winding W4) with two rectifying diodes VI5, VI6 and one current limiting inductor L4. The new modified circuit is given in Fig. 4.

520

Vl-V4

230V (2 50Hz o--+----1

Figure 4. Modified circuit of SM-PSA converter

Another possible problem for topology depicted in Fig.2. is a significant current harmonics in the capacitor C2 and corresponding transformer winding W3 [5]. Tn the case of rectangular voltage waveform the 3rd harmonic component is about 30% that also facilitates increasing of 3rd current harmonic. These current harmonics appear in the range close to load short-circuit. The equivalent circuits for short-circuit mode are given in Fig. 5. The secondary windings here are replaced with rectangular voltage sources Uc and UL, having corresponding leakage inductances Lsc and Lsr connected in series. Short-circuit at rectifier output is replaced here, for the sake of simplicity, by short-circuit at rectifier input. In a short­circuit mode, there is a series resonant tank topology forms in C2 branch.

� l�r a)

b)

Figure 5. Equivalent circuits of PSA converter in short-circuit mode: a) without additional inductor; b) with additional inductors L2 and L3

Circuit in the Fig. 5.b with one more inductor L2 (connected to the transformer tap) and inductor L3 connected in series with capacitor C2 is used to reduce harmonic effect. In its turn, to avoid undesirable resonances the self-oscillation frequency to of this circuit must be defined beforehand. The relation of self-oscillation frequency fo to supply voltage frequency fm can be determined as

(2)

where Xc, Xu, Xn, and Xsc are reactances for fundamental component at supply frequency (i.e. 40 kHz in given study). If self-oscillation frequency is equal to the frequency of third harmonic component, there is a series resonance condition in

C2 branch. And, in this case, the current of 3rd harmonic appears higher than current of main harmonic component /,n. that is extremely undesirable and must be avoided.

The effect of 3rd harmonic changes miserable when .fa /f,n < l.5. Thus, it is a good practice to implement L2 or L3 or both and calculate them so, that

or

Xc /------'''---- < 1,5 xsc +xu +xn

(3)

(4)

The third issue is a current increasing during transition to short-circuit mode. The current grows from 145 A to 200 A or about 1.4 times (Fig. 3). Large current at short-circuit causes more spatter of welded metal that is undesirable [I]. The load short-circuit current can be limited if use a resonant circuit, which includes inductors L2 and L3 (Fig. 4), but inductor L I is missing. The phasor diagram for main harmonic component corresponding to this circuit is given in Fig. 6.

ULl

Ie

Uc

Uu /

Figure 6. Phasor diagram for short-circuit mode without Ll

During short-circuit mode in such a circuit (Fig. 4.) current appears mainly through the one secondary winding W3. Current in winding W2 does not appear due to the voltage on inductor L2 (Un in Fig. 6) equal to voltage on winding W2 (UW2 in Fig. 6). However, the exact equality of the voltages is hardly achievable in practice due to the phase shifts and harmonics. Nevertheless, it was experimentally controlled that current in winding W2 remains low.

The main parameters of circuit components used for final simulation, laboratory prototype and their comparative study are given in Table 1.

E. Parameters of Developed Power Converter

Based on the results of computer simulation the laboratory prototype of converter was developed. Along with power circuits it has a control module to modify the switching frequency and gating pulse width manually. This module is constructed on simple ATmega644 microcontroller. The parameters and data of main power components in the converter experimental setup are given in Table 2.

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TABLE I. SM-PSA MODEL PARAMETERS Switching frequency 40 kHz (pw-I 0 J.ls)

Transformer turns ratio 11:1:1:1

C I capacitance 9 J.lF

L2 inductance O.S J.lH

L3 inductance O.S J.lH

L4 inductance 10 J.lH

LS inductance 4 J.lH

C2 capacitance 2000 J.lF

TABLE II. LABORATORY PROTOTYPE DATA Inverter circuit VKM40-06Pl H-bridge (MOSFET 600V, 38A) LS inductor single layer coil 4 uH, air core

Power transformer Two ElOO ferrite core and litz-wire for primary and secondary

C 1 resonant capacitor FKP4, IJ.lF, 400V, 9 capacitors in parallel

C2 smoothing filter 5x400 J.lF, 400V capacitor L2 resonant inductor 0.5 J.lH air core, single layer L3 resonant inductor O.S JIH air core, single layer

L4 auxiliary inductor 10 J.lH, iron powder torroid core, single layer

L4 current limiting 4 J.lH, air core, single layer inductor Main output rectifier M6W-2S0-2 (Schotky 3 phase full-V9-V14 bridge rectifier 2S0A, 200V) Auxiliary rectifier BYT200PIV -400

VIS&VI6 (ultrafast rectifier)

Figure 7. Photo of the experimental setup

The circuit of converter given in Fig. 4 has been implemented in laboratory prototype. If compare it to the original PSA circuit (Fig. 1), there are added mains rectifier VI-V4 and smoothing capacitor Cl. Inductor Ll is missing while there are added inductors L2, L3, L4 and rectifying diodes V 15, V 16 at output. One additional secondary winding W4 is placed to increase output no-load voltage. And finally one inductor L5 in DC supply circuit is added. It is intended to reduce high current rate during load short-circuit caused by slow reverse recovery of inverter in-body antiparallel diodes that is a drawback of given element in our topology. That last sentence requires some explanations.

The load short-circuit mode is different here due to the energy saved in resonant tank elements (Ll, L2, L3, L4) that goes partly through the transformer and inverter back to the inverter OC supply side at each half-cycle of supply voltage. Wherein, during switching-on of transistor Tl (T2) the current initially flows through transistor and then changes its direc­tion, due to resonance nature, and flows through reverse diode Dl(02) while gating signal is still on (Fig. 11). And when the next half-cycle starts with transistor T3(T4) switching-on this diode 01(02) is still conducting. Because of that, there is a temporary short circuit in DI-T4 (D2-T3) connection. This short-circuit lasts as long as diode 01(02) remains open (depends on its reverse recovery time).

III. SIMULATION AND EXPERIMENTAL RESULTS

The first diagram depicting curves of output current and power measured on laboratory prototype is given in Fig. 8.

P,W I,A

4000 250

200 3000

150

2000

100

1000 50

0 0

0 20 40 60 80 U,V 100

Figure 8. Relation of output current Id (dashed) and output power Pd (solid) to load voltage Uacc (measurements)

As it was mentioned above, the curves of output current and power are shaped out in accordance to current circuit topologies that are forming through rectifier bridge V9-V16. The parametrical "selection" of circuit topology is dependent on arc voltage value Uarc•

The characteristics shown in Fig. 8 are obtained for experimental setup without any feedback and control of inverter switches TI-T4 (operating with constant frequency and gating pulse width). Load of the converter was a variable high power resistor. As one can see, the maximal output power occurs when output voltage is in the range 16-30 V that covers well welding arc range.

The levels of current and power are defined by values of components selected in circuit. The waveforms of current in inverter transformer primary winding and voltage on inverter switch during welding process are given in Fig. 9.

For comparison, the simulation results for the same circuit, parameters and conditions are given in Fig. 10. As it is seen the pictures in Fig. 9 and 10 are quite similar. Also, it could be noted that current through transformer primary winding is close to sinusoidal and on both oscillograms the ZCS

condition in inverter switch is present.

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Figure 9. Waveforms of voltage over inverter switch (1) and current in transformer primary winding (3) during welding (measured)

VT1, V

600 500 400 300 200 100

0 11,A

40 �i l i 2� -�- -=�F. -----=---+�-=�- - - - -J.-

-=== ·20 - - r- - - � - - - - - - - -� I

0.0032 0.00322 0.00324 0.00326 0.00328 0.0033 Time (s)

Figure 10. Simulated voltage over inverter switch (VTI) and current in transformer primary winding (II) during welding

Another point of interest is a short-circuit mode that is typical in case of arc welding. The oscillograms during short­circuit are presented on Fig. I I.

Figure II. Waveforms of voltage on inverter switch (I), current in transformer primary winding (3), and load current at short-circuit (4)

Here, one can observe that current through transformer primary becomes even less in a short-circuit mode that is true for PSA converter breed.

Another diagram (Fig. 12) is given to show behavior of suggested power converter in case of typical transient process short-circuit - welding - short-circuit, that also present all the time during welding. And here become obvious that transition from one mode to another is relatively short and takes only 100 fls while no significant overcurrents is observed that is a favorable condition for good quality welding

• PreVu M 2,00ms oom Factor: 20 NOISe Filter Off

<10Hz . 216A

17.7 A 15:07:23

Figure 12. Measured voltage over inverter switch (2), current in transformer primary winding (I), load current at short-circuit (4), output voltage (3) and

computed power (M)

Also, there is one unusual situation in Fig.12 can be observed in output voltage waveform, namely output voltage is represented by pulse train having higher than switching frequency. Voltage here is pulsating from zero to some amplitude value and has relatively long pauses between pulse packets while current remains relatively steady all the time. Data obtained in previous measurement (Fig. 12) are used to build up a detailed graphics representing welding voltage and current (Fig. 13) and output power during welding (Fig. 14)

200

V,A

175 � � � � <:�

150

125

100

75

50

25 I� ,\� �l �\ ...,J

3,20E-03 3,21E-03 3,22E-03

I� I� ,

3,23E-03

weld voltage

, �\ 3,24E-03 3,25E-03

t,5

Figure 13. Load current and load voltage during welding based on measurement data

P,W 2,5E+04

2,OE+04

1,5E+04

l,OE+04

5,0£+03

j�I\�

0,0£+00 3,20E-03

I\, II 3,21E-03

Output power

...... 3,22E-03 3,23E-03

�I 3,24E-03 3,25E-03

t,5

Figure 14. Output power during welding based on measurement data

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These pulses of voltage are forcing the power to change accordingly. During welding the pulses of supplied power reach up in amplitude up to 25 kW, while average power remains relatively steady and it is about 4 kW that is in accordance with designed power level. However, the welding process itself has not suffered from such an untypical behavior of output and welding quality study is being conducted now.

The corresponding computer simulation has yielded in different oscillograms (Fig. 15). The voltage here is relatively stable and changes slightly with current that could be explained by simplified model of welding arc.

I load, A 200

175

150

125

100

75

50

25

0

V load

'�! 1 ��t���! ��1� 0.0032 0.00321 0.00322 0.00323

Time (s) 0.00324 0.00325

Figure 15. Load current and load voltage during welding (simulation)

The reasons for such unusual voltage behavior obtained during real prototype tests are being studied too. Main hypothesis are the appearance of second harmonic in resonant circuit and parasitic capacitance between output rectifier module and heatsink.

To get some more complete overview of converter applicability its regulation properties were also studied. Output regulation of proposed SM-PSA topology was realized by means of changing its gating pulse width while keeping switching frequency constant. The diagram in Fig. 16 was plotted with acquired measurement data, where time scale corresponds to pause between gating signals.

A

250

225

200

175

150

125

100

75

50

25

o

L

2

' ... , ,

4

�Short-circuit current

� \

\. �

� ..... 1' ........

......... -

8

- 'Load current

� ........

9

..... � 10 11 12 13

I'S

Figure 16. Welding (dashed) and short-circuit (solid) current regulation curves (measured)

The welding current depicted in Fig. 16. decreases almost proportionally to pulse width reduction while short-circuit current behavior is different. Short-circuit current is only slightly changing with pulse width decreasing from 10 fls up to 6 flS and then it has almost linear decrease slope. This insensitivity range is probably caused by frequency response of resonant circuit in SM-PSA converter along with changing of higher harmonic spectrum while decreasing pulse width that was also discussed in [6].

IV. CONCLUSION

Even supply of thermal power to welding spot is a problem in a case of manual arc welding due to the unpredictable change of arc voltage yielding in changing of power of welding arc. The power of arc could be stabilized if current is changed in that way so the product of voltage and current remains constant. If feedback circuit is used to control arc current it is a difficult to achieve needed speed of reaction because of the fast voltage oscillation in welding arc. Some better results can be provided with suggested switch-mode converter with parallel and series resonance alternation (SM­PSA) that has a natural constant power characteristic in welding range. In this range with voltage drop the current increases parametrically without any external control and vice versa. This way we can get a stabilized power at welding bead for nominal arc voltages ranging typically 24-30 V. "Cooperation" of varying resonant tank topologies with output rectifier yields in the required output characteristic.

This characteristic is obtained by utilizing full-bridge rectifier with four-phase input, where one input is connected through capacitor, second and forth through inductor and third input connected directly to transformer secondary. Resonant circuits are selected in a way to have near sinusoidal currents with low harmonic content during welding.

The proposed circuit has been studied by using PSIM simulation software and verified by measurements on converter's prototype experimental setup. The experimental and simulation results were corresponded well, despite the unusual load voltage behavior that will be studied additionally. Nowadays, there are studies running also in output regulation

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options, widening soft-switching range and welding bead quality control.

We also hope that suggested converter topology could provide some new ideas on improvement of manual arc welding power sources.

ACKNOWLEDGMENT

The authors thank Mr. Lauri Kiitt for his support in design and implementation of gating control module.

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[4] C. Wang, Z. Wang, Q. XU, "Study on Dynamic Characteristics of Inverter Arc Welding Power Supply Based on Double-Loop Control," IPEMC 2009. IEEE 6th International. - P. 1609-1612.

[5] K. Janson, 1. Jarvik, "Load Adapting Mains Frequency Resonant Converters for Supplying Electric Arc - a new Way in Power Electronics," PESC 98 Record. 29th Annual IEEE: 1998. - P. 2090-2096.

[6] J. Shklovski, PhD Thesis: "LC Circuit with Parallel and Series Resonance Alternation in Switch-Mode Converters," Tallinn University of Technology, 2007. - 117 p.

[7] 1.. Zhao, X. Bai-Iu, W. Shu-hui, Z. Guo-rong, D. Shan-xu, K. Yong, 'The Sliding Mode Control for Arc Welding Inverter Power Source," Industrial Electronics and Applications, ICIEA, 2008. - P. 1100--1104.

[8] J. Dudrik, N.-D. Trip, "Soft-switching PS-PWM DC-DC Converter for Full-load Range Applications," IEEE Transactions on Industrial Electronics. Vol. 57, No. 8, August 2010. - P. 2807-2814.

[9] H. Pollock, J. O. Flower, "Properties and design of series-parallel load­resonant converters: their potential in marine and marine-related applications," Trans IMarE, Vol. 110, Part 2, November 1997, pp. 95-118.

[10] K. Janson, J. Jarvik and T. Vinnal, "Circuit diagrams of DC-DC line­frequency resonant converters with alternation of parallel and series resonant mode," Proceedings of the 5th International Conference on Electrical Power Quality and Utilization, 15-17 September 1999. Krakow, Poland. pp. 479-485.