[IEEE 2014 International Power Electronics Conference (IPEC-Hiroshima 2014 ECCE-ASIA) - Hiroshima,...
Transcript of [IEEE 2014 International Power Electronics Conference (IPEC-Hiroshima 2014 ECCE-ASIA) - Hiroshima,...
The 2014 International Power Electronics Conference
Load-Independent Current Output of Inductive Power Transfer Converters with Optimized
Efficiency Wei Zhang, Siu-Chung Wong and Chi K. Tse
Department of Electronic and Information Engineering
The Hong Kong Polytechnic University, Kowloon, Hong Kong
Qianhong Chen Nanjing University of Aeronautics and Astronautics
Nanjing 210016, China
Abstract-Inductive power transfer (IPT) systems are resonant converters whose output characteristics are mostly load dependent. Conditions for ensuring a load-independent voltagetransfer ratio of an IPT system have been studied. However, load-independent current output can be more desirable for applications such as battery charging and LED driving. This paper studies the characteristics of IPT systems operating at different frequencies as transconductance converters. Operating frequencies for load-independent transconductance are explored, looking for optimal efficiency. The frequencies interested are found to facilitate the design of a current-output IPT converter with efficient power conversion. The analysis is supported by experimental results.
Index Terms-Inductive power transfer, wireless power transmission, compensation topology, efficiency, load-independent transconductance.
I. INTRODUCTION
Wireless power transmission (WPT) technologies are nor
mally classified according to short-range, mid-range, and long
range applications, which are mainly distinguished by the
operating principle and the frequency band. The short-range
power transmission technology has a typical transmission
distance from a few millimeters to hundreds of millimeters. It
normally works by means of inductive magnetic coupling at a
relatively low frequency (20kHz to IMHz) [1]. In the recent
decade, this short distance wireless power transmission or IPT
technology has emerged as an attractive and user-friendly solu
tion to constructing charging platforms for portable electronic
products [2] and electric vehicles [3], [4].
Major research work of IPT focuses on voltage to volt
age conversion, to select the most appropriate compensation
techniques, isolated to specific application requirements [4]
[7]. However, in some applications, a load-independent current
output is more desirable. For instance, a constant current
output is preferred to drive an LED or to charge battery packs
of electric vehicles. An extra stage of current regulator which
incurs extra power loss is needed if the IPT converter is an
output voltage power source.
Therefore, voltage to current conversion or transconduc
tance IPT systems will be studied in this paper. General calcu
lation and simulation are conducted for the IPT converter using
the secondary series- and parallel-compensations, looking for
This work is supported by Hong Kong RGC GRF project PolyU5258/13E.
[::::::::::::::::::c:,::] Parallel compensation
Primary loop Secondary loop
Fig. I. Series- and parallel-compensation topologies. In this figure, the primary inductance L p is compensated with a series connected capacitor Cp, while secondary inductance Ls is compensated with either a series or parallel connected capacitor Cs. !vI is the mutual inductance of the loosely coupled transformer. Rp and Rs are resistances of primary and secondary coils, and RL is the equivalent loading resistance.
their operating frequencies to achieve maximum efficiency
and load-independent current transfer ratio. The maximum
efficiencies of the two compensation topologies are compared
under the condition of load-independent current output.
II. CIRCUIT MODEL AND EFFICIENCY ANALYSIS
The coils of IPT systems are operating at a frequency
well below their self-resonant frequencies [8], and therefore,
additional compensation capacitors are needed to form the
resonant tanks in both primary and secondary sides of the
loosely coupled transformer. The tuned capacitors of the IPT
system can be series or parallel connected with the transformer
coils. Fig. 1 shows the circuit model for the analysis of steady
state transfer functions. The primary loop of the circuit is
driven by a modulated AC voltage source which is an equiv
alent voltage readily generated from a pulse-width-modulated
DC voltage source using either a simple full- or half-bridge
switching circuit. For a parallel resonant compensated primary,
an equivalent current source is needed. Due to the difficulty
of energy storage in the form of a simple current source, extra
components will be needed to transform it on demand from a
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voltage source, creating extra loss. Therefore, series resonant
primary compensation will be used in this paper [5], [6]. The
secondary loop is compensated with either series connected or
parallel connected capacitor to form resonant tanks as shown
on the right-hand-side of Fig. 1.
Coupling coefficient
TABLE I DEFINITION OF VARIABLES.
Primary winding quality factor
Secondary winding quality factor
k= � LpLs Qp = wR: Qs = *,S
Circuit quality factor of series compensation Q fLs 1 L-S = ds RL Circuit quality factor of parallel compensation QL-P = VC8 RL L8 Circuit quality factor of series compensation Q _ CJ8Ch-8
including RL 0- QS+QL-S
Resonant frequency of primary loop wp= � LpCp Resonant frequency of secondary loop w s = -:rt;c;; v L8C8
A thorough efficiency analysis of both secondary series
and parallel compensations has been carried out [5], [6]. The
system efficiency derived is given as
�(Zr) �(Zs) - Rs T) = T)pT)s = Rp + �(Zr) �(ZS) , (1)
where rip and rls are the efficiencies of primary and sec
ondary loops. The parameters and variables are summarized
in Tables I and II. The tailing subscript -S or -P of Q L indicates that it is of secondary series or parallel compensation
respectively.
For the secondary series compensation, there exists a local
maximum of 'T) at W = W M which is load-dependent.
(2)
Variable operating frequency is needed for achieving this
theoretical maximum efficiency. Fig. 2 shows two types of
power transfer efficiency and the corresponding W M versus
Qo [5], where
Qos = J 2\ (1 + )1 - 3),), and (3)
The two types of efficiency curves are identified as follows:
TABLE IT PARAMETERS NEEDED FOR EFFICIENCY AND Gi CALCULATION,
Series topology Parallel topology
Zs jwLs + jw �s + Rs + RL jwLs + Rs + ' Ie_ I IRL 7W s J'wLp + ,IC + Rp 7f...1..,' P
Zr
iout
rT----,-------------,--------, 2,5ws
0, 91-_____ -+----------*"">=<>A-- 1 2ws �
0.7 e-e--e Efficiency ofii<1I3 -a--e--e Efficiency of A2:1/3
0.6 fr-ts--t:s: Frequency
QO,critical 0.9 12 1,9 3,7
§ g.
1.5OJs I " "
0,5ws Qs
" o '<
Fig, 2, Simulated efficiency and optimal frequency (w AI) versus Qo,
CD For)' < i, the optimal efficiency peaks at Qos' The
value of WM is just slightly higher than Ws in a reasonable
range of loading (close to Qos) to achieve high efficiency.
Therefore, Ws, which is load-independent, is an ideal constant
operating frequency for the best efficiency operation of the
converter.
@ For ). :;0. i, the optimal efficiency increases monotoni
cally with decreasing Qo by increasing WM to 00. However,
the operating frequency is affected by the self-resonant fre
quency of the coils, switching losses and voltage stresses. In
addition, it can be seen from Fig. 2, when). :;0. i, the effi
ciency improvement with increasing frequency is insignificant.
Therefore, Ws can be a constant operating frequency to achieve
high efficiency.
As a result, operating at constant Ws , rl maximizes at
a 2 T)max = --------;>2 ' where a = k QpQs· (l+�)
(4)
rlmax is only decided by the value of a, which is the product
of k2 and the two winding quality factors. Therefore, for an
IPT system whose coupling is usually higher than 0.1, the
requirement for high Qp and Qs is not demanding in order
to achieve a high efficiency.
For the secondary parallel compensation, the efficiency
maximizes at W N, which is load independent.
Ws WN= (1+k2)�' (5)
Fig. 3 gives a comparison of maximum achievable efficien
cies of secondary series and parallel compensations versus
the loading condition QL. In the figure, Qp = Qs = 100. It shows that the two compensation topologies can achieve
nearly identical theoretical maximum efficiency except when
). :;0. i (i.e. k > 0.577 if Qp = Qs). For instance, if k = 0.8, the theoretical maximum efficiency of the converter with
secondary series compensation operating at a much higher
operating frequency can be slightly better than that of the
secondary parallel compensation converter.
III. TRANSCONDUCTANCE
The output current iout through RL of the series- and
parallel-compensated secondary are calculated with parameters
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G' .: (1)
·0 � 0+-W
0.8
0.6
0.4
0.2 1--++-!'"++----f.'l+ I-I-I1
.. .. o . .... . 0.01 0.1
Series @WM ---
Parallel @WN .... .... ....
10
,00=0.1 II--++'IH--+ I ,00=0.2 1\, ,00=0.8
100 1000
Fig. 3. Efficiency comparison of series and parallel compensations.
C '-' '"
r5
0.4,--------------;---------,
0.3
0.2
0.1
Frequency (kHz)
(a)
O.081--I---�==;�==;�--�
.-.. 0.06 1-- --=+----t-+-\-------t----1
g 0;: 0.041----+---+---+-----\;------t---- 1 c5
Frequency (kHz)
(b)
Fig. 4. Transconductances of (a) series and (b) parallel compensations.
given in Table II. The transconductances Gi-S and Gi-P of
secondary series- and parallel-compensations are expressed in
(6) and (7).
G _ iout _ wl'vl i-S - Vin - ZpZS-S + W2 lvI2'
Gi-P = iout = wlvI
Vin ZpZS-P + w2l'v12
(6)
1 1 + jWCSRL' (7)
where Z p and Z s are the impedance of the primary and
secondary networks. In this section, Rp and Rs are neglected
for their small values and insignificant impact on the transcon-
ductance [6]. From (6), Gi-S can be RL-independent when
Zp equals zero, i.e. the operating frequency W equals wp. This frequency can also be found by solving for w from the
equation 8Gi-.s = o. 8RL
The RL-independent transconductance is a very desirable
feature for power converters. It guarantees constant output
current while the loading changes. If Rp is neglected, Zp = 0 requires w = wp. Hence,
1 1 IGi-S(wp)1 =
------'--1 = k JLPLS ' (8)
Wpl> Wp p S which is load-independent.
Fig. 4 (a) depicts the SPICE simulation results of transcon
ductance at various loading conditions of the secondary series
compensated circuit with zero Rp and Rs. In the simulation,
k is set as 0.2 and Cp is selected to resonate with Lp at
wp/(27r) = 200 kHz and Wp = Ws. The QLmin, QLs and
QLmax represent different loading conditions of 1.19, 5.00
and 21.05 respectively, which are calculated by the method
introduced in [5].
Similarly, for the secondary-parallel-compensated converter,
the transconductance in (7) is simplified as
1 Gi-P = . L Z ' where wl'vl + JW w'ivI P + 5RL
5 . 2MC + Zp(1-w2LsCs)
=JW ' s wM .
(9)
(10)
From (9), Gi-P is RL-independent when 5 = o. Solving
for the roots of (10), the frequencies at which Gi-P is RL-
independent can be obtained as
WL = w� +w� -�
2(1 _ k2) and WH = W� +w� +�
2(1 -k2) (11)
(12)
The load-independent transconductances of the converter op
erating at frequencies WL and WH are usually not equal to each
other except when
Wp=Ws�.
Under this condition, we have
1 IGt-P(WL)1 = IGt-P(WH)1 =
JLPLS Wp LpLs
(13)
(14)
The simulated I G i-P I curves are shown in Fig. 4 (b)
by using the parameters of series compensation except the
condition (13).
IV. COMPARISON OF SERIES AND PARALLEL
COMPENSATIONS
The foregoing analyses on the frequencies operating for
load-independent transconductance and maximal efficiency are
derived independently on Sections II and III, and summarized
in Table III.
For a secondary series compensated transconductance con
verter, the Wp and Ws that realize load-independent IGil and
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1�--II--��=hCTffiI�TTI�TTITI �. I " � ' " � 0.8 .' 1,.('- . -.
� 06
W, " I v/ . .. �� ;g : / " . 1\ � 0.4 • I. \ Senes Parallel
.
@OJs @OJH '" 1\ 0.2 / -- .... k�O.l " :,. r'\ J4 = :::: :�� · · · ··f:l> � . 0.01 0.1 10 100 1000
Fig. 5. Efficiency comparison of secondary series compensation converter working at w s and W H versus loading quality factor.
...... QL�3.44 0.24�-
J � 0.1 ...... QL�6.38
C � 0.12 !-J-,.r�-� r.)
0.06
O+----+----f------I------l
0.08
160 180
-+- QL�2.82 ...... QL�12.4
180
200 220 Frequency (kHz)
(a)
200 220 Frequency (kHz)
(b)
240
240
Fig. 6. Transconductances of (a) series compensation and (b) parallel compensation.
maximum efficiency are independent. They can be adjusted
individually by changing values of compensation capacitors
in each side of resonant tank according to their definitions
shown in Table I. For a secondary parallel compensated
transconductance converter, there is no single frequency that
achieves load-independent IGi I and maximum efficiency as the
frequencies never align.
If wp is selected equal to Ws as in [4], [7], the efficiency
comparison of operation at Ws of series compensation and
that at W H of parallel compensation is depicted in Fig. 5. It is
obvious that series compensation has higher efficiency at full
load.
» u �
0.9
·C 0.8 �
L ...
r 'I�� � ...... � Series co /npensation
f-- Parallel ompensatlon
'-H u.l 0.7
06 o 5 20 25
(a)
0.91 ------:;r ...... -e-te-��=--+--- 1 G' � ·CO.8r-����----r-------------T------ 1 S u.l
0.7
5 10 15 25
(b)
Fig. 7. Efficiencies obtain from (a) measurement and (b) calculation.
TABLE TIT FREQUENCIES TO ACHIEVE LOAD-INDEPENDENT IGil AND MAXIMUM
EFFICIENCY.
Compensation Load-independent I Gi I Max. efficiency Series Wp Parallel WL or wH
Therefore, by adjusting the values of Cp and Cs, a sec
ondary series compensated current output IPT converter can
have load-independent output and maximum efficiency by
operating at Ws = wp. V. EXPERIMENTAL EVALUATION
Full-bridge prototype circuits are built to verify the results
from the analyses. The prototypes are constructed with a
loosely coupled transformer whose parameters are shown in
Table IV.
Fig. 6 gives the experimental trans conductances of the
secondary series and parallel converters versus operating fre
quency with three output loading conditions.
The efficiencies versus Q L (adjusted using RL) at a constant
output power of 24 W (adjusted using Yin) are measured
for the two prototype converters. A constant power operation
can maintain a smaller converter temperature variation and
thus a smaller variation of converter component parameters,
leading to a more consistent efficiency measurement. The
calculated and measured efficiencies of the secondary series
compensated converter operating at ws, as well as secondary
parallel compensated converter operating at w H, are compared
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TABLE TV COMPONENTS AND PARAMETERS USED IN THE EXPERIMENTAL
CONVERTERS.
Transformer and compensation k Lp Ls Cp Cs
Value 0.18 32.77 JiB @200kHz 31.77 JiH @200kHz 19.32 nF / 630 V 19.93 nF / 630 V
as shown in Fig. 7. All the efficiencies are measured with
soft switching conditions. From the comparison of efficiency
curves shown in Fig. 7, and some other measurements with
stronger and weaker coupling coefficients, the series compen
sated IPT current output converter always performs with better
efficiency.
VI. CONCLUSION
The operating frequencies of an inductive power trans
fer output current converter are studied with primary series
compensation and secondary series or parallel compensation.
Operating frequencies for maximum efficiency and load
independent transconductance are identified and compared
with the two secondary compensation techniques. Operating
at Ws of the secondary series compensation and operating
at W N of the secondary parallel compensation have a similar
maximum efficiency. However, for the secondary series com
pensation, the load-independent transconductance operating
frequency Wp and maximum efficiency operating frequency
Ws can be adjusted independently. Thus, the two desired fea
tures can be achieved simultaneously. While for the secondary
parallel compensation, the two frequencies never align. The
efficiency will be lower than the maximum when it has load
independent current output. Two experimental prototypes are
built and the experimental results confirm the analysis.
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