Quadrature VCO Using the Composite Right-/Left-Handed

Dual-Resonance Resonator Sheng-Lyang Jang#, Heng-Fa Teng*, Chia-Wei Chang#, and Chao-Wei Hsieh#

#Department of Electronic Engineering, National Taiwan University of Science and Technology, Taipei City, Taiwan

* Department of Computer and Communication Engineering, Nan Kai University of Technology, NanTou, Taiwan

Abstract — A dual-band quadrature voltage-controlled

oscillator (QVCO) has been studied and implemented in the TSMC 90 nm 1P9M CMOS technology. The proposed QVCO comprises two complementary cross-coupled dual-resonance VCOs and 4 capacitors in a ring for coupling the two differential VCOs. The QVCO actually uses an LC-based composite transmission-line resonator. The die area of the QVCO is 0.831×0.97 mm2. At the supply voltage of 1.1 V, the total power consumption is 2.75 mW. The free-running frequency of the QVCO is tunable from 8.77/3.68 GHz to 9.12/4.16 GHz as the tuning voltage is varied from 0.0/0.6 V to 0.5/1.1 V. The measured phase noise at 1 MHz frequency offset is -120.39/-118.75 dBc/Hz at the oscillation frequency of 8.79/3.847 GHz and the figure of merit (FOM) of the proposed QVCO is -194.86/-186.05 dBc/Hz.

Index Terms —CMOS, dual-band quadrature voltage-controlled oscillator (QVCO), dual-resonance resonator, capacitor coupling.

I. INTRODUCTION

The wireless communications have been rapidly progressing, and demands for wireless radios to support multi-band and multi-standard are rapidly increasing. One of the major difficulties in implementing a dual-band transceiver is a dual-band voltage-controlled oscillator, because most VCO’s are not capable of providing an agile reconfiguration between dual bands. Dual-band quadrature voltage controlled oscillators (QVCOs) can be an integral part of homodyne dual-band transceivers, and fully-integrated dual band QVCOs can play an important role in the transceivers to provide multifunction services. There are two straightforward design methods can be employed to design dual band QVCOs. Firstly, dual-band QVCOs can be formed by modifying the existing single-band QVCOs to have the capacitance-switching and inductance switching resonators [1]. Secondly, two dual-band differential VCOs [2] are coupled via a passive and/or active coupling network to form the QVCO. Single-band QVCOs formed with capacitor coupling have been widely studied [3-6] and they use an LC-based composite right-/left-handed transmission-line (T-line) resonator. The aim of this letter is to design a dual-band CMOS QVCO for multi-band application.

Q-

Vtune L2

VDD

K

L1

M3 M4

CV1 CV2

L3

M1 M2Q+

I+I- Q-Vtune L5

VDD

K

L4

M7 M8

CV3 CV4

L6

M5 M6

Q+

I+ I-

V(LC)

C1 C2 C3 C4

(a)

Vtune

L5

K

L4

CV1

CV2

L6IP2 Vtune

L2

L1

CV1

CV2

L3IP1

C3

C4

C1

C2

VO(Q-)

-VO(Q+)

jVO(I-)

-jVO(I+)

K

(b)

Fig. 1. (a) The proposed dual-resonance dual band QVCO. (b) Equivalent block diagram of the QVCO. The proposed QVCO uses an LC-based composite right-/left-handed T-line dual-resonance resonator. The dual-resonance LC-based T-line resonator is consisted of two dual-resonance LC resonators coupled via 4 high-Q MIM capacitors configured in a ring to provide the quadrature phase. Consequently a high-figure-of-merit QVCO can be designed in the 90nm CMOS technology.

II. CIRCUIT DESIGN

Figure 1(a) shows the proposed dual-band cross-coupled QVCO consisted of two differential dual-band VCOs [2] coupled by a MIM capacitor ring, the common nodes of MIM capacitors are wired to the outputs of two differential VCOs. The first dual-band differential cross-coupled VCO consists of a dual-resonance resonator and a pair of cross-coupled

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(a)

(b) Fig. 2. Simulated voltage waveforms at the common node of L1 and Cv1, I+, I- in Fig. 1(a).VDD = 0.9V. (a)Vtune = 0 V, and (b) Vtune = 1 V.

transistors. The inductors L1, L2, configured as a transformer, and the accumulation-mode varactors (Cv1, Cv2) form the series-tuned LC-tank and the inductor L3 and the parasitic capacitors form the parallel LC-tank. Both the LC tanks are combined in one dual-resonance LC resonator. pMOSFETs (M1, M2) and nMOSFETs (M3, M4) are used to generate the negative resistance to compensate the resistive tank loss. The control voltage Vtune is used to tune the oscillation frequency. The second differential VCO comprises a dual-resonance resonator, and pMOSFETs (M5, M6) and nMOSFETs (M7, M8) to generate the negative resistance. The transformer (L4, L5) and the accumulation-mode varactors (Cv3, Cv4) form the series-tuned LC-tank and the inductor L6 and the parasitic capacitors and varactors form the parallel LC-tank. Fig. 1(b) shows the simplified block diagram of the T-line QVCO, the current sources are due to the two cross-coupled pairs to generate the equivalent negative resistance. The 4-coupling capacitors (C1~C4) form a ring to provide 360 degrees of phase shift for the loop and the phase shift across each MIM capacitor is 90 degrees. A stop band [7] is found in the frequency region of 1/[L1mCv1+L3xCv1] < ω2 < 1/( L1mCv1), where L1m = L1(1+M), L3x = L3/2, M is the mutual inductance. The oscillation frequency is determined by the Barkhausen criteria.

The QVCO uses the complementary core, so the output voltage of QVCO will be clipped at VDD by the pMOS transistor and at ground by the nMOS transistor in case the voltage swing is large enough. Figure 2 shows the simulated steady-state voltage waveforms V(LC), V(I+) and V(I-) at Vtune = 0 V (high-frequency band) and Vtune = 1 V (low-frequency band). The output voltages V(I+) and V(I-) in Fig. 2(a) and Fig. 2(b) are differential and have similar output voltage swing. This will lead to easy circuit design of the dual-band circuit following the VCO. The voltage V(LC) at

Fig. 3. Chip photograph of the proposed DB VCO. Fig. 4. Measured tuning range by varying Vtune of varactor. VDD = 1.1 V. Vtune = 0 to 1.1 V. the common node of L1 and Cv1 can be below the ground level, and above the supply level and it is smaller/larger than the voltage V(I+) in Fig. 2(a)/Fig. 2(b). In the series-tuned mode the voltage across the varactor is larger than that in the parallel-tuned mode; as a result, the larger RF modulation effect leads to large AM-to-PM noise conversion, which can be reduced by replacing the varactors with the BBS varactor configuration [8] to reduce the voltage swing across each varactor through the capacitive voltage divider.

III. MEASUREMENT RESULTS

The QVCO was designed and fabricated in the TSMC 90nm 1P9M CMOS technology. Figure 3 shows the micrograph of the proposed dual-band QVCO with a chip area of 0.831×0.97 mm2 including all test pads and dummy metal. The passive L3, L6 are designed as a 4-turn octagonal inductors, and the inductors (L1, L2) and (L4, L5) are designed as a 4-turn octagonal transformer. Accumulation-mode nMOSFETs are used as varactors for frequency tuning. The output spectrum was measured using the Agilent E4407B spectrum analyzer. The oscillating frequency of the high-band/low-band QVCO can be tuned 8.77/3.68 GHz to 9.12/4.16 GHz as shown in Fig. 4 as the tuning voltage is varied from 0.0/0.6 V to 0.5/1.1 V. As the control voltage is increased, the varactor’s capacitance decreases, therefore the output frequency increases. The power consumption of the QVCO core is 2.75 mW at the supply voltage of 1.1 V.

Figure 5(a) shows the measured high-band frequency spectrum of the proposed QVCO. The output power is -1.75 dBm. Figure 5(b) shows the measured low-band frequency spectrum of the proposed QVCO. The output power is -2.17 dBm. The measurement data shows that the QVCO can

time (ns)63.3

V (V

)

063.4 63.5 63.6

0.25

0.5

0.75

1.0V(I+)V(I-)V(LC)

time (ns)67.2 67.4 67.6 67.8

-0.25

0.25

0.5

0.75

1.0

0

V (V

)

V(I+)V(I-)V(LC)

Vtune (V)0.0 0.2 0.4 0.6 0.8 1.0

Freq

uenc

y(G

Hz)

4

5

6

7

8

9

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Fig. 5(a). Measured spectrum of the quadrature VCO at 8.79 GHz. VDD = 1.1 V, and Vtune = 0 V. Fig. 5(b). Measured spectrum of the quadrature VCO at 3.85 GHz. VDD = 1.1 V, and Vtune = 0.8 V. Fig. 6. Measured output waveforms of the I and Q channels. VDD = 1.1 V, Vtune = 0.46 V and fosc = 9.1 GHz.

provide large output power either in the high band or low-band, this is crucially important for the application of the QVCO to the RF transceiver. The measured phase noise at the oscillation frequency of 8.79/3.847 GHz is -120.39/-118.75 dBc/Hz at 1 MHz offset frequency. The figure of merit (FOM) is defined as where ωo is the oscillating frequency, Δω is the offset frequency, L{Δω} is the phase noise at Δω, PDC is the DC power consumption of the measured QVCO core. The FOM of this proposed QVCO is -194.86/-186.05 dBc/Hz. Figure 6 shows the measured time-domain output waveforms of two quadrature VCO output buffers by using the Agilent 54855A Infiniium oscilloscope. The average output phase error is 2.8° introduced partly by cables and connectors. TABLE I summarizes the performance comparison of QVCOs.

IV. CONCLUSION

This letter has proposed an LC-based dual-resonance right-/left-handed transmission-line resonator to design a dual-band QVCO. The QVCO uses a MIM capacitor ring, which introduces no flicker noise to the resonators, while it provides accurate quadrature phase. The proposed QVCO has been

TABLE I PERFORMANCE COMPARISON OF DUAL-BAND QVCOs.

successfully implemented in 90nm CMOS technology, and it provides dual-band frequency tuning via a single voltage control to select the high-band or low-band operation. Experimental data shows that the proposed QVCO can have high performance and is useful for radio-frequency circuit application.

ACKNOWLEDGEMENT

The authors would like to thank the Staff of the CIC for the chip fabrication and technical supports.

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quadrature VCO for low power and low phase noise application,” IEEE Conf. Radio Frequency Integration Technology (RFIT), December 2007, pp 310-313.

[2] S.-L. Jang, Y.-K. Wu, C.-C. Liu and J.-F. Huang, ” A dual-band CMOS voltage-controlled oscillator implemented with dual-resonance LC tank,” IEEE Microw. Wireless Compon. Lett., vol. 19, no. 12, pp.816-818, Dec. 2009.

[3] S.-L. Jang, C.-W. Chang, C.-C. Shih, and C.-W. Hsue, ” Quadrature VCO based on an LC-ring in 0.18 μm CMOS Technology,” Microwave and Optical Technology Lett., vol. 54, no. 2, pp. 474-477, Feb., 2012.

[4] S.-L. Jang, W.-C. Liu, J.-F. Huang, and C.-W. Chang, ” A CMOS transconductance-enhanced Colpitts-like quadrature VCO,” in press, Microwave and Optical Technology Lett., 2011.

[5] S.-L. Jang, S.-S. Lin, C.-W. Chang, and S.-H. Hsu, ” Quadrature VCO formed with two Colpitts VCO coupled via an LC-ring resonator,” PIER C, vol. 24, page 185-196, 2011.

[6] G. Li and E. Afshari, “A low-phase-noise multi-phase oscillator based on left-handed LC-ring,” IEEE J. Solid-State Circuits, vol. 45, no. 9, pp. 1822–1833, Sep. 2010.

[7] A. Lai, T. Itoh, and C. Caloz, “Composite right/left-handed transmission line metamaterials,” IEEE Microw. Mag., vol. 5, no. 3, pp. 34–50, Sep. 2004.

[8] A. Bonfanti, S. Levantino, C. Samori, and A. L. Lacaita, “A varactor configuration minimizing the amplitude-to-phase noise conversion in VCOs,” IEEE Trans. Circuits Syst. I, vol. 53, no. 3, pp. 481–488, Mar. 2006.

[9] S. Rong and H. C. Luong, “A 1V 4 GHz-and-10 GHz transformer-based dual-band quadrature VCO in 0.18 μm CMOS,” CICC2007, pp.817-820, Sept. 2007.

[10] H. Shin, Z. Xu, and M. F. Chang, “A 1.8-V 6/9-GHz switchable dual-band quadrature LC VCO in SiGe BiCMOS technology,” in IEEE RFIC Symp. Jun. 2002, pp. 71–74.

olog201mW

log10}{ΔFOM DCPL (1)

Tech. VDD(V)/ Pd.(mW)

Freq. (GHz)

Tuning range (GHz) (%)

FOM (dBc/Hz)

[1] 0.18μm CMOS

1.7/8.6 - -

4.12~ 4.89 1.8 ~ 2.06

-180.5 -181.8

[9] 0.18μm CMOS

1.0/6 10.0 4.2

9.48~ 11.36 3.27 ~ 5.02

-182 -181

[10] 0.18μm BiCMOS

1.0 9 6

9.45~ 9.9 6.3 ~ 6.6

-178 -177

This 90nm CMOS

1.1/2.75 8.79 3.85

8.77~ 9.12 3.68 ~ 4.16

-194.86 -186.05

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