HYBRID FIELD GENERATOR

CONTROLLER FOR OPTIMISED

PERFORMANCE

by

Christopher Teboho ‘Moleli

B-Tech: Electrical Engineering

A research dissertation submitted in compliance with the

requirements for the degree

Magister Technologiae: Electrical Engineering

in the

Faculty of Engineering

Port Elizabeth Technikon

Promoter: Mr. A. Roberts

M Dip Electrical Engineering (Heavy Current)

Co-promoter: Dr. H. A. Van der Linde

PhD: Electrical Engineering

Submission Date: 10/12/2003

i

Acknowledgements

The following persons are sincerely acknowledged for their support that

contributed to the successful completion of the research project:

Mr. A. Roberts and Dr. H. A. Van der Linde for their continued

academic guidance.

Faculty of Engineering, for providing test instruments and

equipment to meet the project requirements.

ii

Declaration

I Christopher Teboho ‘Moleli hereby declare that:

The work in this dissertation is my own original work;

All sources used or referred to have been documented and

recognized; and

This dissertation has not been previously submitted in full or partial

fulfillment of the requirements for an equivalent or higher

qualification at any other recognized education institution.

___________

C. T. ‘Moleli Date 10/12/2003

iii

Abstract

Battery charging wind turbines like, Hybrid Field Generator, have become

more popular in the growing renewable energy market. With wind energy,

voltage and current control is generally provided by means of power

electronics. The paper describes the analytical investigation in to control

aspects of a hybrid field generator controller for optimized performance.

The project objective is about maintaining the generated voltage at 28V

through out a generator speed range, between 149 rpm and 598 rpm. The

over voltage load, known as dump load , is connected to the control circuit

to reduce stress on the bypass transistor for speeds above 598 rpm.

Maintaining a stable voltage through out the speed range, between 149rpm

and 598rpm, is achieved by employing power electronics techniques. This

is done by using power converters and inverters to vary the generator

armature excitation levels hence varying its air gap flux density. All these

take place during each of the three modes of generator operation, which

are: buck, boost and permanent magnet modes.

Although the generator controller is power electronics based, it also uses

software to optimize its performance. In this case, a PIC16F877 micro-

controller development system has been used to test the controller

function blocks.

iv

Table of contents

Page

ACKNOWLEDGEMENTS .............................................................. I

DECLARATION .......................................................................... II

ABSTRACT............................................................................... III

TABLE OF CONTENTS .............................................................. IV

LIST OF FIGURES .................................................................... IX

LIST OF SYMBOLS ................................................................. XIII

ABBREVIATIONS ..................................................................... XV

DEFINITION OF CONCEPTS ................................................... XVII

CHAPTER 1 ............................................................................... 1

Introduction ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.1 Controller overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.2 Problem Statement .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

1.3 Sub-problem Statement .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

1.4 Hypothesis.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.5 Project Delimitations... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.6 Project Outline ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.7 Project Significance ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

1.8 Methodology ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

v

CHAPTER 2 ............................................................................... 8

Project literature ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.1 Literature overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.2 Hybrid Field Generator characteristic background ... . . . . . . . . . . . . . . . . . . 9

2.2.1 Permanent magnet mode ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

2.2.2 Boost mode... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

2.2.3 Buck mode ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

2.3 Charge control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.3.1 Rectification ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

2.3.2 DC voltage and current regulator .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

2.3.3 Charge load protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.4 Excitation control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.4.1 DC/DC converter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

2.4.2 Exciting current control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

2.4.3 Current protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

2.5 Charge and excitation control system .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

2.5.1 Micro-controller selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

2.5.2 Signal conditioning ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

2.5.3 Interfacing... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

2.6 General aspects of the project literature ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

CHAPTER 3 .............................................................................. 39

Controller specifications and considerations ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

3.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

3.2 Generator specifications ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

3.2.1 Generator tests .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

3.3 Battery bank specifications ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43

3.4 Components selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43

3.4.1 Discrete components .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

3.4.2 Passive components .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

vi

3.4.3 Integrated circuits .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

3.5 Transformer and choke ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51

3.6 Sensors .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

3.6.1 Hall-effect sensors .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

3.6.2 Optical and electromagnetic isolators .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

CHAPTER 4 .............................................................................. 57

Charge voltage regulation design ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

4.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

4.2 Uncontrolled rectification ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

4.2.1 Controller protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

4.3 Over voltage load ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

4.4 Charging voltage regulation ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

4.4.1 Overload protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

4.5 Battery protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

CHAPTER 5 .............................................................................. 74

Excitation circuit design ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

5.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

5.2 DC/DC converter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

5.2.1 Transformer Selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92

5.3 Exciting current control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94

5.3.1 Exciting current polarity changer .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98

CHAPTER 6 ............................................................................ 100

Control Circuit design ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100

6.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100

6.2 I/O circuits .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100

6.3 Switching circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111

6.4 Micro-controller .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114

vii

6.5 Operation control modes ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115

6.5.1 Analog and digital input signals execution ... . . . . . . . . . . . . . . . . . . . . 117

6.5.2 Operation mode selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118

CHAPTER 7 ............................................................................ 122

Conclusion ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122

7.1 Conclusion ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122

7.2 Recommendations ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 126

7.3 Future work ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127

List of references ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129

Application Notes ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133

Abstracts .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133

CD-ROMs .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134

APPENDIX A .......................................................................... 135

Flow charts and blocks code ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 135

APPENDIX B .......................................................................... 158

Hybrid field generator performance characteristics tests results .. . . 158

Generator test results .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 158

APPENDIX C .......................................................................... 176

Tests boards-data ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 176

Line voltage harmonics ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 176

Load current influence on ripple voltage at zero exciting current ... 177

Switched exciting voltage and current waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . 185

viii

APPENDIX D .......................................................................... 186

Schematics test results .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186

APPENDIX E........................................................................... 204

Project Test boards and schematics .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 204

ix

List of figures Page

Figure 1.1 Project Block Diagram.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

Figure 2.1 Permanent magnet mode generator speed vs generated voltage

... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Figure 2.2 Average permanent magnet air gap flux density distribution

over a five teeth pole. .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Figure 2.3 Boost mode speed vs voltage at different exciting current

levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

Figure 2.4 Boost mode average air gap flux density at different exciting

current levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

Figure 2.5 Buck mode speed vs voltage at different exciting current levels

.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

Figure 2.6 Buck mode average air gap flux density at different exciting

current levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

Figure 2.7 Charge control block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

Figure 2.8 Three-phase AC/DC converter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

Figure 2.9 Six-pulse bridge rectifier .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Figure 2.10 Excitation control block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Figure 2.11 Gapped and un-gapped core magnetic characteristic curves23

Figure 2.12 Line noise filtering ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

Figure 2.13 Screened transformer ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Figure 2.14 Continuous waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Figure 2.15 Discontinuous waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

Figure 2.16 Frequency response of three common types of fi lters .. . . . . . . . . 30

Figure 2.17 Charge and excitation control system .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Figure 2.18 Line current sensor block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

Figure 2.19 Speed sensor block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 2.20 Opto mechanical speed sensor .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

Figure 2.21 Speed disk mounting ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

x

Figure 2.22 Line currents and speed sensors power supply and rotor

excitation current control power supply ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

Figure 3.1 Generator tests setup ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

Figure 3.2 IR2153 functional blocks ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

Figure 3.3 (a) Inverting op-amp (b) non-inverting op-amp ... . . . . . . . . . . . . . . . . . . 47

Figure 3.4 (a) Differential op-amp (b) Voltage follower ... . . . . . . . . . . . . . . . . . . . . . 48

Figure 3.5 Instrumentation op-amp ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50

Figure 3.6 LOC11X .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56

Figure 3.7 LOC11X V i n vs Vou t . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56

Figure 4.1a Line voltage (Ch 1) and line current (Ch 2) waveforms ... . . . . 58

Figure 4.1b Line current 5A waveform.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

Figure 4.2 Open circuit excitation characteristics curves at 24V line ... . . 61

Figure 4.3 Ripple voltage waveform at 0A load current .. . . . . . . . . . . . . . . . . . . . . . . . 61

Figure 4.4 Ripple voltage at 2A load current .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61

Figure 4.5 Ripple voltage waveform at 2A load current and 2A excitation

current .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61

Figure 4.6 Ripple voltage waveform at 2A load current and 4A excitation

current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

Figure 4.7 Voltage protection circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

Figure 4.8 MOSFET switching voltage levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

Figure 4.10 Battery bank voltage regulator.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

Figure 4.11 Battery bank voltage monitor .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73

Figure 5.1 Static air gap flux density ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

Figure 5.2 Dynamic air gap flux density ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

Figure 5.3 Rotor (armature) excitat ion schematic diagram .... . . . . . . . . . . . . . . . . . 78

Figure 5.4 IR2153 RT vs frequency curves ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79

Figure 5.5 DC/DC converter drive signals, signal 1 for Q1 signal 2 for Q2

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79

Figure 5.6 DC/DC converter transformer primary signals with respect to

battery bank positive terminals. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

Figure 5.7 DC/DC converter transformer secondary signal .. . . . . . . . . . . . . . . . . . . 80

xi

Figure 5.8 DC/DC signal overshooting and ringing ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81

Figure 5.9 Open circuited rotor back emf waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . 82

Figure 5.10 Freewheeled back emf waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82

Figure 5.11 Zener clamped freewheeled back emf waveform.... . . . . . . . . . . . . . . 83

Figure 5.12 rotor circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

Figure 5.13 Exciting current control circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

Figure 5.14 Continuous mode waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87

Figure 5.15 Discontinuous mode waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

Figure 5.16 Continuous mode waveform at 30% duty cycle ... . . . . . . . . . . . . . . . . 91

Figure 5.17 PWM generator block diagram.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94

Figure 5.18 Characteristic behavior of the PWM generator .. . . . . . . . . . . . . . . . . . . 95

Figure 5.19 PWM duty cycle levels at different DC levels. . . . . . . . . . . . . . . . . . . . . 96

Figure 5.20 PWM generator schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96

Figure 5.21 PWM generator signal waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98

Figure 5.22 Exciting current polari ty changer .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99

Figure 6.1 Three terminal voltage regulator with by pass transistor .. . . . 102

Figure 6.2 ADC voltage protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102

Figure 6.3 Active low pass filter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103

Figure 6.4 60Hz active low pass fi lter characteristics. . . . . . . . . . . . . . . . . . . . . . . . . 104

Figure 6.8 Speed sensor schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106

Figure 6.9 Speed pulses waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106

Figure 6.10 Line current sensor characteristics curve ... . . . . . . . . . . . . . . . . . . . . . . . 108

Figure 6.11 Line current sensor schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . 109

Figure 6.12 4N25 current and voltage characteristic curves ... . . . . . . . . . . . . . . 109

Figure 6.13 Hall-effect current sensor characteristic curves ... . . . . . . . . . . . . . 110

Figure 6.14 Voltage detector schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111

Figure 6.15 Load disconnect switch ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 112

Figure 6.16 2N3904 VB E vs VC E . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113

Figure 6.17 Relay drive circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114

Figure 7.1 On load excitation curves at 24V line voltage and at 600rpm

.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124

xii

Figure 7.3 Hybrid field control station network ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 128

xiii

List of Symbols

a : armature

A : ampere

Ac : core cross sectional area

Av : amplifier gain

Cp : pulse count per second

CH : speed disk holes

D : duty cycle

DX : diode ( note)

e : instantaneous voltage

E : maximum voltage

f : frequency

G : gate

Hz : hertz

i : instantaneous current

I : current

k : kilo

L : inductance

mT : milli tesla

M : mega

N : revolutions per minute

p : pole pair

P : power

Ph : phase

Q : transistor

R : resistor

Sp : speed

t : t ime

T : period

V : volt

xiv

W : watt

Wa : window area

X : reactance

Z : impedance

π : PI

τ : t ime constant

°c : degree celcius

Φ ex : excitation flux

Φ F : field coil flux

Φ L : leakage flux

Φ m : permanent magnet flux

dT/dt : temperature rate of change

di/dt : current rate of change

Note: x refers to a subscript. Example D1, D2

xv

Abbreviations

AC : alternating current

AC/DC : alternating current to direct current converter

BJT : bipolar junction transistor

CMRR : common mode rejection ratio

CT : t iming capacitor

DC : direct current

DCX : direct current voltage level (note)

D/A : digital to analog

DC/DC : direct current to direct current converter

emf : electromagnetic force

EMI : electromagnetic interference

FET : field effect transistor

Ho : high output

Icg : charge current

IC L : controlled load current

IcTot a l : total input charge current

IC : integrated circuit

IO : input output

IR : international rectifier

Isc : short circuit current

IV : current voltage

L-disc : load disconnect

LED : light emitting diode

Lo : low output

LOC : linear optocoupler

MOSFET : metal oxide field effect transistor

Neg : negative

O.C. : open circuit

xvi

Ov : over voltage load

pfm : peak frequency modulation

pwm : pulse width modulation

Pos : positive

Pot : potentiometer

rpm : revolution per minute

RFI : radio frequency interference

Rlyx : relay (note)

RT : t iming resistor

sd : shutdown

UVLO : under voltage lockout

Vb a t : battery bank voltage

Vb mi n : minimum battery bank voltage

Vb mx : maximum battery bank voltage

VC E(SAT) : collector to emitter saturation voltage

Vcg : charge voltage

V i n : input voltage

VGS :gate to source voltage

Vline : l ine voltage

Vo : output voltage

V re f : reference voltage

V ro t or : rotor voltage

V t r c : tracking voltage

VSD : variable speed drive

Note: x refers to the position of the relay. For an example: Rly1, first

relay.

xvii

Definition of concepts

Hybrid Field Generator: Combination of a permanent magnet field

generator and wound field generator.

Hybrid Field Generator: The power controller that stabilizes the

power controller generated power by controlling the

generator excitation.

Performance data sheets: Documentation of tests results.

Bit addressable : A micro-controller whose single bit can be

micro-controller addressed without bit masking.

Low power components : Components that consumes less power.

On board power losses : Power that is dissipated by power

components on-board.

Region of operation : Operation region of high efficiency.

Safe region : Region where the controller is operated

safely.

MOSFET driver : MOSFET switching integrated circuit

MPLAB : It is a software-developing tool used to

write PIC micro-controller programs.

xviii

Uncontrolled rectifier : One type of AC-to-DC converter that does

not need a triggering pulse in order to

operate.

Controlled rectifier : Another type of AC-to-DC converter that

needs to be triggered before it operates.

1

Chapter 1

Introduction

1.1 Controller overview

Wind turbines have become widely used for the generation of electricity.

A disadvantage of wind turbines is that they do not generate constant

voltage as required by most electrical appliances. This is essential

because the generated voltage is directly proportional to the wind speed.

The generated voltage is maintained at the rated voltage over a wide range

of wind speeds with the aid of an electronic power controller.

A novel generator development for a wind turbine application is the

Hybrid Field Generator. The generator characteristics are a combination

of those of the permanent magnet generator and the wound rotor

generator. The Hybrid Field Generator can therefore operate as a

permanent magnet generator or a field controlled generator.

The Hybrid Field Generator operates in three modes: permanent magnet

mode, buck mode and the boost mode. The operation mode is determined

automatically by the captured energy level. Varying the excitation of the

field coils during buck mode and boost mode, controls the generated

voltage and this is achieved by making use of an electronic power

controller.

The power controller monitors the generated voltage and the charging

current to the battery bank. It ensures that sufficient current is supplied to

the load. It also maintains the energy dissipated by the battery bank and

the DC load within the rated values. During strong winds, when the

2

generator speed is over 598rpm, the captured wind energy is more than it

is needed. At this point, the excess energy is transferred to excess energy

load (over voltage load), normally referred to as a dump load. As a

result the generator is always kept running under load instead of shutting

the hybrid system down and leaving the generator running under no load.

This is to prevent uncontrolled acceleration of the generator, mainly when

strong wind is blowing, which will result in over-speeding.

Although the battery bank is one of the loads, it also ensures that the load

voltage remains stable regardless of the generated voltage fluctuations

within limits. The battery bank acts as a power source for the controller

electronic circuit. It also serves as an excitation field coils power source

during buck and boost modes.

The power controller applied in this context uses a micro-controller to

continuously monitor the generated voltage. It compares the voltage with

the reference voltage set within the software and thereafter generates

control signals that select operation mode. The controller also has an

electronic voltage monitor that compares the generated voltage to the set

PWM reference voltage to determine excitation pwm duty cycle during

boost mode and buck mode.

1.2 Problem Statement

The electronic power controller for the Hybrid Field Generator discussed

in the introduction does not operate at optimum efficiency.

1.3 Sub-problem Statement

The Hybrid Field Generator required an electronic controller that would

carry out the control functions efficiently and effectively. For effective

3

power transfer the circuit has to dissipate as little power as possible to

ensure optimum performance over a wide wind speed range.

Although the charge circuit seemed to be the circuit solely responsible for

the controller performance, other circuits such as monitoring circuit and

excitation circuit, also contributes positively towards the performance. To

ensure optimum performance, the circuits power dissipation has to be as

low as possible.

To increase control abilities of the power controller, the monitoring

circuit (micro-controller) has to have analog inputs and outputs as well as

digital inputs and outputs.

The excitation circuit DC/DC converter MOSFETs driver had to be self-

oscillating to minimize the driver components count and optimize DC/DC

efficiency. Therefore, the driver ensures converter stability and reliability

hence an improved converter performance.

1.4 Hypothesis

The need for an electronic power controller has been established for the

effective control of the hybrid field generator. Based on the results

determined by an analytical investigation of the technical design

parameters of the electronic power controller, the controller performance

will be optimised producing maximum efficiency.

1.5 Project Delimitations

The project’s emphasis is on the Hybrid Field Generator and its power

controller. The power controller protects the generator from overloads and

4

short circuits. It would also provide deep discharge and overcharge

protection to the battery bank.

1.6 Project Outline

The objective of the project was to set the standards with regards to where

and how the power controller is to be operated to optimize its

performance. The controller design considerations were also taken in to

account and will be discussed in the following sections. The following

topics were taken into consideration:

Charge control circuit to stabilize the charging voltage at

28VDC for different levels of the charging current.

Control and monitoring circuit to continuously monitor the

controller inputs and outputs, and thereafter generates control

signals.

Controlled exciting current injector circuit and controlled

DC/DC converter circuit which generates the exciting current.

5

Generator

Chargecontrol

Control &Monitoring Unit

ControlledExciting Current

Injector

Battery Bank

ControlledDC/DC

Converter

DC loadcontrol

Load

Figure 1.1 Project Block Diagram

1.7 Project Significance

The Hybrid Field Generator is a Variable-speed generator and as a result

the generated voltage fluctuates in direct proportion to the wind speed.

Although the generated voltage would tend to fluctuate, a charge control

circuit is used to keep it constant over a wide range of wind speeds. This

is achieved by continuous monitoring of the generated voltage and by

varying the generator air gap flux density.

The power controller ensures the improved generator reliability, lower

operation power losses and lower maintenance costs. The objective is not

only to maintain the generated voltage and to keep the generator operating

safely, but also to maximize the generator service period.

The generated voltage is dependent on the wind speed, therefore the

power controller controls forward exciting current when there is

6

insufficient wind speed and vice versa (Van der Linde, 2001, pp.63-100).

The controller does not only stabilize the generated voltage but also

protects the generator against overloads and short circuit. It also provides

protection against overcharging and deep discharging of the battery bank

as was mentioned earlier. This is necessary as it prevents the battery bank

from being damaged.

1.8 Methodology

The following procedure was followed in the project research:

A literature survey was done and related information was

accumulated from relevant information sources.

The generator was tested for performance although there

were some limitations imposed by the test equipment during

air gap flux density tests; limitations like temperature change

with respect to time (dT/dt) at maximum speed and maximum

load. The tests results were analyzed and documented.

Study characteristics of switching components like

MOSFETs and BJTs. Sample circuits will be designed, built

and tested on the project board as per section and the results

will be analyzed accordingly.

Some circuits will also be simulated with circuit-maker. This

will be done as a further study on the circuit performance.

The project schematics, as per controller part, will be drawn.

The project PCBs will manufactured and populated for

further testing of the power controller main parts. This will

7

be done to facilitate the controller performance data

acquisition for further analysis.

The PIC 16F877 development system board will be populated

and will be used as a sample micro-controller board for the

project. The project communication control command

software functions will also be tested and modified

accordingly.

The tests results will be complied and documented

8

Chapter 2

Project literature

2.1 Literature overview

The control of the voltage generated by the generator is achieved by

adjusting the generator’s air gap flux density during the buck, boost and

permanent magnet modes of operation. This is because the generated

voltage is directly proportional to the flux density. The generator

specifications and its magnetic response curves under boost mode, buck

mode and permanent magnet mode described in (Van der Linde, 2001,

pp.63-100) are similar to the ones discussed in this document. The

response curves were drawn after performing number of tests on the

generator. Refer to appendix B.

The power controller employed power and digital electronics as well as

software commands to continuously monitor the generator DC output and

the battery bank voltage. It also generates control signals during

generator’s three modes of operation:

Permanent magnet mode

Buck mode

Boost mode

Power electronics and digital electronics control techniques employed in

the project are discussed in the literature. The techniques are classified

according to the following subsystems:

Charge Control

Control and monitoring

Excitation control

9

2.2 Hybrid Field Generator characteristic background

The Hybrid field generator operates in the same manner as a permanent

magnet synchronous generator. The output of the generator is rectified

with either a controlled or an uncontrolled full-wave bridge rectifier. The

generator air gap flux density is controlled by a wound field coil.

Depending on the generator speed the field coil is either excited in buck

or boost mode to keep generated voltage constant.

2.2.1 Permanent magnet mode

When the generator is running at 149 rpm, the permanent magnet flux will

be sufficient to produce 24VAC . That is, ideally

ex = m 2.1

Where Φ ex is excitation flux

Φ m permanent magnet flux

This is true if the leakage flux is not considered. The speed could also

extend beyond the synchronous speed, 750 rpm (50 Hz), in this

application. Because the generator charges a 24V battery bank, the

permanent magnet mode is suitable for 149 rpm (Van der Linde, 2000,p

63).

Figure 2.1 shows the relationship between the generator speed in rpm and

the generated voltage, DC voltage. It is clear that when speed increases

above 149rpm, the generated voltage also increases to levels high than the

desired voltage level. Conversely, the voltage level decreases with the

decreasing speed.

10

Permanent mode V/Speed curve

0

100

200

300

0 500 1000 1500 2000

Speed (rpm)

DC

Vol

tage

(V)

Voltage

Figure 2.1 Permanent magnet mode generator speed vs generated voltage

Average permanent mode flux density Curve

0.00

2.00

4.00

6.00

8.00

10.00

12.00

0 1 2 3 4 5 6

Pole teeth position

Flux

den

sity

(mT)

0 A

Figure 2.2 Average permanent magnet air gap flux density distribution

over a five teeth pole.

2.2.2 Boost mode

During boost mode the flux density was increased by 65% over permanent

magnet flux density level at maximum exciting current, 5A.

An increase in forward exciting current resulted in a proportional increase

in the generated emf.

11

With reference to figure 2.3, it is clear that the generated voltage is

directly proportional to the exciting current and the generator speed. This

shows that although the generated voltage decreases with a decrease in

speed it can still be maintained by increasing the exciting current. An

increase in the exciting current causes a proportional increase in the air

gap flux density. See figure 2.4.

ex = (m + F ) - L 2.2

Where Φ F is field coil flux

Φ L is leakage flux

Boost V/Speed Curves (O.C)

0

100

200

300

400

500

600

0 500 1000 1500 2000

Speed (rpm)

Vol

tage

(V)

0 A1 A2 A3 A4 A5 A6 A7 A

Figure 2.3 Boost mode speed vs voltage at different exciting current

levels

Figure 2.3 shows that besides an increase in speed the generated voltage

also increases with the increasing exciting current. This is because the air

gap flux density is directly proportional to the exciting current.

12

Average Boost mode flux density

0.002.004.006.008.00

10.0012.0014.0016.0018.00

0 1 2 3 4 5 6

Pole teeth position

Flux

den

sity

(mT) 0A

1 A2 A3 A4 A5 A

Figure 2.4 Boost mode average air gap flux density at different exciting

current levels

During the boost mode, the field coils flux flows in the same direction as

that of the permanent magnet, therefore the active air gap flux density is

the sum of the fluxes.

2.2.3 Buck mode

During buck mode the generated voltage is kept constant at 24V

regardless of increased speed. This is achieved by reducing the exciting

current. The exciting current polarity is also reversed in the process.

Under these conditions the flux produced by the field coils flows in the

opposite direction to that of the permanent magnet, hence a reduced air

gap flux density. Therefore

ex = m – (F + L) 2.3

13

Buck V/Speed Curves (O.C)

0

100

200

300

0 500 1000 1500 2000

RPM (rpm)

Vol

tage

(V)

0 A-1 A-2 A-3 A-4 A-5 A-6 A-7 A

Figure 2.5 Buck mode speed vs voltage at different exciting current levels

Average Buck mode flux density

0.00

2.00

4.00

6.00

8.00

10.00

12.00

0 1 2 3 4 5 6

Pole teeth position

Flux

den

sity

(mT) 0A

-1 A-2 A-3 A-4 A-5 A

Figure 2.6 Buck mode average air gap flux density at different exciting

current levels

It was found that in buck mode flux density could be reduced by 60%

below the permanent magnet flux density.

2.3 Charge control

As the power controller is expected to be more efficient, an uncontrolled

rectifier is used for AC/DC conversion. The uncontrolled rectifier is

14

simple and it does not need a control circuit, as it is a case with a

controlled rectifier.

With reference to figure 4.9, Q1 occasionally transfers excess voltage to

the over voltage load to ensures that the bypass transistor is operated

below its VC E (SAT) .

The charge regulator maintains the output voltage at the rated voltage,

and provides the generator with short circuit and overload protection. It

also prevents the battery bank from being over charged or deep

discharged. It has the ability to reduce the AC component that is usually

superimposed on the charging current and voltage. The AC component

normally causes the battery bank to heat up and eventually starts gassing

(Lander, 1993, pp. 207-210).

The choice of power components was based on the merits of one

component over the other for a specific application such as MOSFETs as

opposed to BJTs with emphasis on power dissipation and operating

frequency bandwidth. MOSFETs dissipate less power during conduction

because their forward resistance is smaller than that of BJTs. They can

also handle high frequencies better than BJTs.

LM723 was selected over other linear voltage regulators because of its

popularity in DC power supplies. This is further discussed in section 4.4.

The blocking diode and the freewheeling diode are introduced to protect

the charge control circuit from the back EMF (Ahmed, 1999, p.157). The

blocking diode also protects the charge control circuit form the battery

bank discharge. This takes place when the charge voltage level is below

that of the battery bank.

15

Figure 2.7 Charge control block diagram

Figure 2.7 shows how the generated voltage is processed to charge the

battery bank. The first stage is the generator stator. The stator is a star

configured source that delivers the generated voltage to the uncontrolled

three-phase bridge rectifier. The rectified voltage is then fed to an over

voltage load control circuit. After this stage follows the 28V voltage

regulation circuit and the back emf protection diodes which are blocking

and freewheeling diodes.

2.3.1 Rectification

The rectification forms the first part the controller. It converts the AC

voltage from the generator into a DC voltage. The AC voltage waveform’s

magnitude and frequency are not stable. They fluctuate in sympathy with

the wind speed.

16

According to Faradays law, the voltage induced in each of the stator coils

is be expressed as:

ea = ωNΦ p sin ωt 2.3

= Ea max sin ωt

Because the generator is a balanced star configured generator, the induced

voltages are displaced by 120° from each other (Sen, 1997, p.215). That

is, if ea is at 0,° eb is at +120° and ec is at -120° if positive phase

sequence is followed. They are also expressed as:

ea = Ea max sin (ωt) 2.4

eb = Eb max sin (ωt +120°) 2.5

ec = Ec max sin (ωt - 120°) 2.6

Owing to the fact that the generator configuration is balanced, all the

induced voltages are equal in magnitudes and so are the maximum

voltages. The latter is expressed as:

Emax = 4.44fNΦ pKW 2.7

The line voltages: VAB , VB C and VC A are also equal and are expressed as

VAB = √3 E sin ωt . 2.8

Figure 2.8 is the graphical representation of the line voltages and the DC

voltage after line voltage rectification. The ripple voltage is smaller

because the line voltages are rectified with a six-pulse diode rectifier and

therefore a reduced AC component exists in the charging DC voltage.

17

The ripple fluctuates between 1.414 V s and 1.225 V s where V s is the RMS

value of the line Voltage. An average output voltage is expressed by

Vo (avg) = 1.654 Vm 2.9

Where Vm is maximum phase voltage, therefore in terms of line voltage it

is given as

Vo (avg) = 0.955 V lm 2.10

Where V lm is the maximum line voltage. According to Ohms law an

average output current is given by:

Io (avg) = Vo (avg) / R (Ahmed, 1999, pp. 200-202). 2.11

Figure 2.8 Three-phase AC/DC converter

18

Uncontrolled rectification

One of the wind turbine system’s objectives is to capture and store most

of the energy in a cost effective manner. An uncontrolled rectifier

presents the system with a simple and cost effective AC/DC power

conversion. This is because it does not need a drive circuit and has

minimum power loss, because there are only six converting components.

See figure 2.9.

3phgenerator

Voltageregulator

ABC

Figure 2.9 Six-pulse bridge rectifier

Controlled rectification

A controlled rectifier’s main drawback is the production of radio

frequency interference (RFI) and generation of harmonics (Ahmed, 1999,

p. 151). This rectifier needs a drive circuit that introduces complexity in

rectification mainly in rectifying the generator AC voltage. The other

problem is that the generator is a wind driven generator. The result is that

the frequency of the generated AC voltage is not periodic. Therefore, an

on-delay timing circuit becomes very complex.

2.3.2 DC voltage and current regulator

The voltage regulator is the key component of the charging circuit. The

regulator ensures that the charging voltage does not follow the input

voltage fluctuations. It maintains the voltage stable at 28V, which is used

19

to ensure that the charging current flows from the regulator to battery

bank as well as to other DC loads connected to it . The voltage level

accommodates the battery bank full charge voltage level. The voltage

level is 27.24V for a 24V battery bank at 25oc (Clive, 1991, p. 208) . Table

3.1shows lead acid cell voltage.

The regulator circuit incorporates current limiting to provide over load

and short circuit protection to the charging circuit. This is also protecting

the generator. In most cases current limiting with fold back characteristics

is preferred since it restricts the short circuit and overload current to a

low value, less than the short circuit current and the overload current.

Since the input voltage can be as high as 125VDC , an over voltage load is

used to reduce stress on the regulator circuit bypass transistor. This way

the circuit is protected from high voltages that lead to high power

dissipation on the bypass transistor, which could result in the transistor

failing.

Foldback current limiting

The charge current protection employed is foldback current limiting. This

technique provides both over current and short-circuit protection to the

charge circuit. It is very dominant in linear power supplies (Billings,

1999, p. 113). It is capable of limiting current to less than overload

conditions when there is a short circuit or an overload, hence power loss

reduction on the linear series components of the circuit.

20

2.3.3 Charge load protection

Although the charge circuit maintains the charging voltage, it also

provides the battery bank with an overcharge and deep discharge

protection. It also eradicates under voltage and over voltage conditions

from the 24V sensitive DC loads by isolating them from the battery bank

when the condition exists.

2.4 Excitation control

This sub-system deals with the generator air gap flux density control,

which is performed by adjusting the exciting current during buck and

boost mode (Van der Linde, 2001, pp. 63-100). The DC/DC converter is a

push-pull converter. It is used because of its advantages over other

converters; one of them being the fact that it uses transformer coil

bidirectionally. The converter is driven from a self-oscillating FET driver,

IR2153D, to switch current in an alternating manner through the

transformer primary coil.

The exciting current is adjusted with a step down DC/DC chopper during

the forward and reverse excitation. Refer to section 5.2 for further

discussion on step down DC/DC chopper operation.

The exciting current is generated from the DC/DC converter that is driven

from the battery bank. Its magnitude is determined by the duty cycle of

the PWM signal that drives the step down DC/DC chopper. It is then fed

to the polarity changer that changes the current direction through the

generator rotor coil. See figure 2.10.

21

Figure 2.10 Excitation control block diagram

2.4.1 DC/DC converter

A power electronic converter employing switching devices such as

MOSFETs is used to control rotor excitation of the generator. This

includes the associated control and interfacing circuits (Rahman, 2002, p.

4). It allows fast control of exciting current and voltage. The control of

the converter takes the efficiency and reliability of the converter into

account.

The power converter’s power supply is a fixed DC/DC converter. In order

to maintain the generated voltage, the air gap flux density of the generator

is either increased or decreased by exciting the rotor. Therefore, the DC

current has to be controllable. The excitation is inversely proportional to

the rotor speed and the generated voltage.

The power converter configuration used in this application is a class A

type buck chopper. The chopper is driven from a switched PWM

generator. The configuration uses gate turn-off devices such as MOSFETs.

22

The DC/DC converter is a push-pull type. The DC converter in this

application uses power semiconductor switches that are operated in the

switched mode. When they are turned off, they block the supply voltage

across them with no current flow, hence dissipating zero watts. This is

due to

P = I * V (w) 2.12

where P is power, I is current and V is voltage. Therefore, when I is equal

to zero amperes P will equal zero watts. When they are turned on, the

voltage drop across them is very low for IRF9640 when VG is equal to

12V. Refer to figure 4.8. When VG is 12V Vd s is approximately 0.011V.

At full load, at 10A, the power dissipation is equal 0.11watts. Therefore

they do not consume significant power in both the on and off state.

An overlap of the switches turn-on and turn-off transients is carefully

considered in selecting the switching frequency devices. This is so to

ensure that power losses due to switching are lower compared to the

output power.

Converter transformer

Ferrite is an ideal material for inverters transformers and inductors

operating in the range of 20 kHz to 3 MHz. This feature makes it an

excellent material for the named devices. The material has high

permeability and very high resistance to eddy currents. Therefore, eddy

current losses are negligible. However, it is not recommended for high

current applications because it has lower saturation flux compared to that

of laminated and powdered iron material (Pressman, 1989, pp. 240-241).

23

A push-pull inverter is efficient because it makes bi-directional use of a

transformer core, thus providing an output with low ripple and low noise.

Although the inverter is efficient, it suffers from flux imbalance that in

most cases leads to failure. Therefore, the choice of switching devices is

very important. The flux imbalance in the inverter becomes less serious

when MOSFETs are used as switching devices.

The inverter transformer core is gapped to avoid saturation under a DC

bias condition. This is because the saturation causes transistor failure in

push-pull topology when transistors have unequal switching

characteristics (Magnetics). The core can then handle larger volt-second

inequality (Pressman, 1998, p 49).

When the gap is placed in series with the magnetic flux lines, it t ilts the

slope of the hysteresis loop, keeping the point where the loop crosses the

zero-gauss (0-G) level (called coercive force Hc) fixed (Pressman, 1998, p

49). See figure 2.11.

Figure 2.11 Gapped and un-gapped core magnetic characteristic curves

24

EMI filters

The switch mode power supply generates excessively high frequency

noise, RFI and EMI, which hampers the performance of the micro-

controller. The noise can be reduced with an EMI filter. The filter is

inserted between the switch mode power supply and the control circuits of

the controller. The commonly used EMI filters are:

• Common mode noise filter

• Differential noise filter

Common mode noise filter Differential noise filter Figure 2.12 Line noise filtering

Windings are connected in series with input power lines in common mode

noise filters. The windings are such that the flux set by one winding

cancels the one set by the other winding, keeping the power lines free of

noise.

Screening

Although gapping reduces residual magnetic effects, it radiates EMI and

RFI that causes interference in the other parts of the circuitry. The

radiation effects are reduced by screening the transformer with a thin

copper screen that is 1% percent of the rated output power depending on

the air gap (Billings, 1999, pp. 1.46-1.48)

The air gap for ETD core is usually filled with a plastic shims in the

center and outer legs.

25

The shims reduce RFI generation in the transformer as the gaps are

constant but RFI is reduced further when the center leg is ground down to

twice the shim thickness (Pressman, 1998, pp. 49-50).

Figure 2.13 Screened transformer

2.4.2 Exciting current control

This section describes figure 5.3 . A DC/DC converter is the first stage of

the exciting current control. It converts 24VDC to 48VDC . The converter is

driven with self-oscillating PWM IC, IR2153 , for the DC conversion. The

converter output is fed to the PWM chopper whose main task is to vary

the exciting current in proportion to the generated voltage deviation from

the control reference voltage. The deviation determines the driving signal

duty cycle. See figure 5.21.

The polarity of the exciting current is also determined by the nature of the

voltage deviation. The positive deviation occurs when the generated

voltage is greater than the control reference voltage. During this time

26

buck mode is selected and the exciting current polarity is reversed so that

the air gap flux density is reduced. Conversely, the boost mode is selected

and the flux density is increased.

DC buck chopper

A DC/DC converter is used to convert a fixed battery bank voltage to a

DC voltage that is adjusted with a DC buck chopper for the rotor

excitation. The chopper can either be driven from a PWM or PFM circuit.

A DC chopper varies generator rotor exciting current to levels that best

suit the excitation requirements. This happens at every instant of change

of generated voltage above or below the generated voltage reference

voltage. As a result, air gap flux density is varied accordingly.

Vex = tON * VS 2.13 (tON + tOFF)

= tON * Vs

T

Therefore

Iex = Vex Ra

PWM is used to vary the excitation voltage at switching frequency of

2KHz.

The chopper is operated in two modes, continuous and discontinuous

mode under both buck and boost mode of the generator.

27

During continuous mode the period T is very small compared to the

chopper time constant τ, that is T << τ. The time constant is a ratio of the

armature (rotor) inductance to the armature resistance.

τ = La 2.14 Ra

In continuous mode, the current Ia continuously flows into the rotor. This

feature makes continuous mode preferable in armature excitation.

Discontinuous mode is undesirable because of the breaks in the Ia .

Normally this mode comes into play when TON is approximately equal or

greater than τ.

TON

TTOFF

Imax

Imin

Ia

iin

iD

Continuous mode

Va

Figure 2.14 Continuous waveform

Discontinuous mode occurs as a result of a small armature inductance.

This can be avoided by increasing the armature inductance. The

inductance is increased by adding more armature coil turns such that

2πfLa >> Ra (Rahman, 2002, pp. 4-17).

28

The minimum required inductance is determined as

La = TOFF * Ra 2.15 2

TON

T TOFF

Imax

Ia

iin

iD

Discontinuous

Discontinuous mode

Figure 2.15 Discontinuous waveform

When the rotor coil inductance is increased, its current peak-to-peak

ripple reduces. The other way is by increasing the switching frequency so

that the period T is much smaller than the time constant τ.

If the rotor (armature) inductance is high, the excitation current Iex ripple

is approximately Zero. Then the current can be given by

Ia = Va 2.16 Ra

where Iex = Ia

In addition, the armature inductor voltage is expressed as

Ea = L dia 2.17 dt

29

2.4.3 Current protection

The excitation current protection is achieved by continuous monitoring of

the exciting current, so that the converter can be shutdown when the

current exceeds the rated current. In addition to this, a rated current fuse

is also included in the control circuit.

2.5 Charge and excitation control system

This sub-system forms the core of the hybrid field wind energy system. It

monitors the inputs and outputs of the power control system and it

generates control signals accordingly. The micro-controller interfacing

circuits are discussed in chapter 6. The circuits isolate the micro-

controller from the noisy analog circuit and also protect the micro-

controller input and output pins from voltages higher than 5V. The

circuits also provide the micro-controller with noise protection and

interference protection to ensure an effective and efficient operation. See

section block diagram, figure 2.17.

Active filter design

In analog signal processing, filtering plays an important role. It is used to

get rid of unwanted frequencies, commonly known as noise. Practical

filters are normally of the second or higher order. There are three

configurations of active filters, namely

• Butterworth filter which presents flattest response characteristics

• Chebyshev filter which presents steeper roll-off with less flat

response characteristics

• Bessell filter which presents rapid roll-off with a number of

ripples

30

Figure 2.16 Frequency response of three common types of filters

The filters present the following advantages to the signal translating

circuit:

• Provide gain to overcome signal attenuation in the filter.

• Present high input impedance to prevent excessive loading of the

driving circuit and present low output impedance to prevent the

filter from being affected by the load that it is driving.

• The gain can be easily adjusted over a wide range of frequencies

without changing the desired response (Boylestad, pp. 683-687).

• Sharp cutoff characteristics and high-level attenuation of unwanted

signals.

They are compact and cheaper than those that use passive components.

31

Figure 2.17 Charge and excitation control system

2.5.1 Micro-controller selection

With reference to figure 2.17, nine inputs are monitored with a micro-

controller. The inputs are read from three main parts of the system, which

are:

• Stator line current and voltage circuit,

• Rotor excitation circuit and

• Charge and monitoring circuit.

All the line currents and speed are measured at the stator side of the

generator. They are depicted as Ir , Iy, Ib and Sp respectively. The four

inputs are linked optically to the micro controller.

32

The exciting current Ir t , l ike the rotor voltage V r t are on the rotor side of

the generator. Ir t is magnetically linked to the microcontroller with a hall-

effect sensor and V r t is optically linked with a linear opto-coupler. This

way, the microcontroller is isolated from the rotor

The generated voltage, charge current and the battery voltage are read

directly from the charge circuit. This is due to the circuit being supplied

from the same source with the micro-controller, unlike the inputs from the

stator and the rotor.

When the inputs are read and executed, the control command signals are

sent to: DC/DC converter, excitation polarity changer, fault load connect

and disconnect (Con and Dis) and DC load connect and disconnect

respectively.

The description in the preceding paragraph and figure 2.17 shows that a

micro-controller with at least 8-analog input channels and 5-digital output

may be used. A single channel can still be used at the cost of execution

time. This is because the analog inputs are multiplexed before being read.

Digital inputs requirement is determined by the method used to read

speed. That is if an 8-bit counter is used to count the generator speed

pulses an 8-bit inputs port would be required for speed. Alternatively, the

micro-controller could be used to read speed pulses directly from the

speed sensor.

2.5.2 Signal conditioning

In section 2.5.1, 8-analog inputs are read with the micro-controller but

prior to the actual reading, the signals are conditioned so that their

magnitudes fall within the 0 to 5V voltage range. The conditioning

process is performed with linear operational amplifiers.

33

The linearity of the signals eliminates a need for a complex mathematical

function within software commands. Therefore, the analog inputs can be

expressed as:

y = mx + c 2.18

Where y is the tailored analog output that corresponds with the actual

parameter, m defines the gradient of the y, dy/dx. x is the analog input

and c is the constant value that exists in the input signal when the

measured quantity is zero. This value signifies the value of y when x is

zero. It is set to zero with a zero adjust.

Figure 2.18 Line current sensor block diagram

Figure 2.18 shows an example of a signal conditioning circuit block

diagram. From the current sensor, the current signal is fed to a low pass

filter and then an opto coupler. The signal is then sent to a differential

amplifier. After the amplifier, the signal is converted from analog to

digital value that can be understood by the microcontroller.

2.5.3 Interfacing

As the generator speed is the only input that involves timing, two methods

of speed sensing are discussed in the following paragraphs.

34

Opto-electronic speed sensor

This type of sensor operation principle is based on counting the number of

either positive half cycle, negative half cycle or both half cycles of an AC

voltage waveform. This takes place after the AC signal has been half

rectified or fully rectified. Under a fully rectified signal condition, the

counter gives the count that is twice the actual number of cycles

completed. Therefore, two to get the actual number of cycles completed

divides the count.

The generator speed measured in hertz with the opto-electronic speed

sensor. The generator speed can be expressed in RPM as:

N = f * 60 2.19 p

Where: N is revolutions per minute (rpm)

f is generator frequency in hertz (Hz)

p is number of pole pairs

Each negative half cycle generates a pulse across an opto isolator. These

pulses are read with a digital counter that converts them into binary data.

This can be used to generate control signals within the micro-controller.

The sensor has a pulse count limit of 60 Hz set within hardware as

indicated in figure 6.4. Further details about this sensor are discussed in

the following sections.

Figure 2.19 Speed sensor block diagram

35

Opto-mechanical speed sensor

With reference to figure 2.20, the speed disk is an optical encoding disk.

The disk has holes around its outer edge. On either side of the disk, there

is an infrared LED and an infrared sensor. The spaces between the holes

in the disk break the light beam from the LED, so that the infrared sensor

picks up the light pulses as the rotor shaft rotates. The rate of pulsing is

directly proportional to the speed of the rotor. The pulses are read with

the micro-controller that turns them into binary data. An added advantage

of the opto-mechanical isolator is that it does not need secondary power

supply compared the opto-electronic sensor.

In this case, the generator speed is determined by dividing the pulses

count per second with the number of disk holes. That is

f = Cp /CH (Hz) 2.20

N = f * 60/p (rpm) 2.21

Where Cp is the pulse count per second

CH is the number of disk holes

36

+5V +5V

Pulses

Rotor Shaft

Speed disk

Optical Speed detector

Shaft

Disk

R1 R2

Figure 2.20 Opto mechanical speed sensor

Opto-mechanical (stray light)

It was found that almost every opto-mechanical sensor suffers from

optical noise (Gauvin, Freniere). The noise is normally caused by bright

sources of light, light that manifests itself in two ways: ghost images and

scattered light that reaches the infrared receiver and introduces false

pulses. In this application, a stray light source is an arcing between the

generator brushes and the slip rings. This is due to the generator

vibration.

Stray light effects can be reduced by shaping a speed disk as indicated in

figure 2.21 and by using non-reflective material.

37

Figure 2.21 Speed disk mounting

2.6 General aspects of the project literature

PIC micro-controller and the 8051 micro-controller families development

systems, like MPLAB, are found online from Microchip and from Intel

respectively. The micro-controllers are easily affected by noise and

interferences, like EMI, and these hamper them from performing at

optimum efficiency. The microcontrollers have to be protected against

noise and interferences for a better performance. The high efficiency and

accuracy in micro-controllers can be achieved by using linear amplifiers

and linear signal translating circuits such as an instrumentation amplifier.

Most of the switching, in power controllers, is done by means of

MOSFETs. VGS of the MOSFETs is limited to voltages up to 20V

therefore MOSFETs should have gate to source voltage protection to

counter act negative effects of gate to source over voltage. The effects

such as gate to source over voltage lead to MOSFET failure.

38

According to the generator tests and specifications, the exciting current is

limited to 5A maximum (Van der Linde, 2001, pp. 69-71). It is therefore

important to consider current protection for the project. Foldback current

limiting presented a continuous load connection to the generator as its

short circuit and overload currents are lower than the load current

(Loveday, 1992, pp.83-87).

Figure 2.22 Line currents and speed sensors power supply and rotor

excitation current control power supply

The controller has three independent circuits that are electrically isolated

from each other. See figure 2.17. The circuits are linked optically and

magnetically to bring about communication between them and the micro

controller.

Although the line currents and the generator speed sensing as well as

rotor excitation circuit are powered from a DC/DC converter, batteries

can be used as an alternative.

39

Chapter 3

Controller specifications and considerations

3.1 Overview

The controller specifications and considerations are determined by the

generator specifications and its optimum operation requirements. Due to

these requirements, the generator was tested for more information about

its hardware as well as its performance.

3.2 Generator specifications

Generator Specifications:

Power 1.25 kW

Number of phase 3 Ph star configuration

Generator voltage 24 VAC => 32.414 VDC

Load current 7.404AAC => 10ADC

Design speed 264 rpm

Cut-in speed 149 rpm

Cut-out speed 548 rpm

Field: Coil resistance RC 3.367 Ω

Current 5 ADC

Coil power loss 84.175 W

(Van der Linde, 2000, pp.46-47)

40

Measured rotor and stator quantities:

Rotor: Coil inductance La 40.9 mH

Coil resistance Rc 3.5 Ω

Field circuit resistance RFC (Ra) 4.18 Ω

Brushes resistance, slip rings resistance and contacts resistance are a

difference between RC and RFC .

Therefore, their resistance is 0.680 Ω .

Stator: Coil inductance per phase Ls 4.75 mH

Coil resistance per phase R s 0.611 Ω

3.2.1 Generator tests

The generator tests discussed in this section were performed to determine

some of the hardware quantities, such as rotor inductance, that were not

specified in (Van der Linde, 2001, p 46). The following tests describe the

tests performed on generator:

• Rotor DC test and AC test to determine rotor actual resistance and

inductance respectively.

• Stator DC test and AC test to determine stator actual resistance and

inductance respectively.

The following test were performed under permanent magnet mode, buck

mode and boost mode of the generator excitation at various speeds except

for static test:

• Open circuit test and short circuit test

• Static air gap flux density distribution as per pole

• Dynamic air gap flux density

• Load tests at mid range speed, 374rpm, and at 600rpm

• Synchronous load test

41

• Open and closed rotor terminals back emf tests with blocking diode,

freewheeling diode and zener diode for voltage clamping. The test

was also performed without the zener diode.

Refer to appendix B and C for the test results.

DC/ACcurrentsource

A B

Primmover

VSD3 ph

mains

vA

vv

A A

Wind generator

Stator

Rotor

R

Y

B

RL

A DC/ACcurrent sourceB

Y

C

Figure 3.1 Generator tests setup

Rotor and stator hardware tests

The fist rotor test was the DC test. Switch A was closed for this test. A

small DC voltage was applied across rotor terminals so that one-ampere

flowed into the rotor. So according to Ohm’s law

R = V 3.1 I

Where V is applied voltage

I is the current flowing into the rotor

R is the rotor resistance

Therefore R ro t or = 4.18Ω (V s = 4.2V)

42

The similar DC test was performed for the stator test although R, in this

case, was the sum of two stator-coils resistance. For example, when DC

was applied between line Y and B, as shown in figure 3.1, R was the sum

of Rco i l Y and Rco i l B . Therefore, the stator R per coil was

R = R = 0.611Ω (V s = 9.61V) 2

The rotor AC test was performed in a similar manner to the DC test except

that an AC source was used. The current flow was once again set to one

ampere. So, because the rotor impedance Z expression was:

Z = V = √(R ro t or2 + XL ro t or

2) 3.2 I XL ro t or = √(Z2 – R ro t or

2) 3.3

Therefore Lro t or = XL ro t or*ω-1 = 41.195mH (V s = 13.6V)

The stator AC test was performed in a similar manner to that of the rotor.

Because the stator coils were in series, they were treated in the same way

as the resistors in series. Each coil inductance was equal to half the two

stator coils inductance. The stator inductance per coil Ls t a t o r was

Ls t a t o r = L = 4.75mH (V s = 9.6V) 3.4 2

Open circuit test was performed with switch A closed at all levels of

exciting current at different speeds. The generated voltage was measured

after a six-pulse bridge rectifier with switch B open. Similarly, the short

circuit was performed with switch A, switch B and switch C closed. The

short circuit current was also measured after the bridge rectifier. The test

results indicated that the generated voltage and the load current were

directly proportional to the generator speed and the exciting current.

43

The load tests were also performed in the similar manner to those of the

open and short circuit except that switch C was left open. It was found

that the generated voltage and the load current were indirectly

proportional to each other.

3.3 Battery bank specifications

Battery bank specifications were determined from table 3.1. Because a

24V battery had 12 cells, the maximum voltage range was 27V – 27.24V

over temperature range of 25°c - 29°c respectively.

Table 3.1 Recommended battery cell voltages

3.4 Components selection

The component selection was based on the merits of individual

components towards the controller hardware for optimized performance.

Components with minimal power losses were preferred, mostly the

temperature compensated components. These types of components assured

a stable and reliable hardware and linear response for accurate hardware

monitoring.

44

3.4.1 Discrete components

Switching components such as MOSFETs are susceptible to failure in

switching converters. This is because they are subjected to severe

switching stress due to rapid change in voltage and current. Although

MOSFETs have increasingly replaced BJTs in switching converters, they

still need to be operated within their safe operating areas to ensure

reliability and optimum converter performance (Ang, 1995, p. 320).

MOFETs are preferred in switch mode power supplies and push pull

inverters, because they do not have storage time, as is the case with BJTs.

More importantly, the on-state voltage in MOSFETs increases with the

increasing temperature. Thus the runaway condition would not occur. The

fact that the MOSFET on-state VGS voltage is proportional to temperature,

it provides feedback, which tends to correct the flux imbalance (Pressman,

1998, pp. 44-45).

BJT drive circuits are complex as BJTs are current and voltage controlled

devices unlike MOSFETs. MOSFETs are voltage-controlled devices and

they need simple drive circuits (Ahmed, 1999 p. 39) although their VGS

has to be maintained at 12VDC by means of a zener diode.

Schottky diodes were preferred over junction diode for their fast

switching speed. The diodes turn-off time made them suitable for high

frequency applications such as DC/DC converters. The converter

switching frequency is 100 kHz and PWM frequency is 2 kHz.

Switching loses

Switching losses at high frequencies in a power transistor, BJT, could

contribute to more than half of the total power loss. This could be because

45

of overlapping of the collector current and collector-to-emitter voltage

during turn on and turn off process of the BJT. The power loss during this

process would be the sum of power during delay time and the rise time

(Pressman, 2002, pp 4-17, 60-62)

IGBTs have an advantage over both MOSFETs and BJTs because they are

a combination of the two named transistors. Thus, they can serve as either

MOSFETs or BJTs. However, they have a very limited frequency range,

10 – 50 kHz, (Ang, 1995, p. 351). Owing to this fact, MOSFETs were

chosen over the IGBTs because their range extends into Megahertz. This

was because the converter frequency was chosen to be 100 kHz.

3.4.2 Passive components

Fuses

Most applications call for protection against fault conditions that in most

cases destroy semiconductor switches like those of the IR2153. Fusing is

the simplest protective option although it is often ineffective at

preventing damage to the switches, and is not resettable. This is a

problem with applications where very fast response is required. However,

protection is sometime achieved by a rapid turn off of the FETs.

Resistors

Trimpots, cermet type, were found to be reasonably high quality

components with good stability. They are very dominant in applications

where small signals are amplified like in measuring instruments that are

microprocessor or controller based.

46

Metal film resistors also play an important role in bringing quality,

stability and accuracy in a signal translating circuit. It is because they

have very low tolerance, equal or less than +1%, and very low

temperature coefficient, + 50 ppm, as well as extremely low current noise

level (RS Components, 2003, p. 801).

3.4.3 Integrated circuits

Figure 3.2 IR2153 functional blocks

IR2153D is an upgrade of self-oscillating control IR ICs and it has

enhanced electrical performance and functionality. The IC has a built in

feature, under voltage detect UVLO, that ensures that the gate drive

outputs, HO and LO are both low when bias becomes too marginal for

comfortable gate drive to the output transistors. UVLO also ensures a

repeated start-up sequence and control bias current required for various

IC elements.

Most of the control ICs like IR2153D, are susceptible to electrical noise

that impairs their performance. However, the noise is reduced with a

47

decoupling capacitor and a reservoir capacitance from Vcc to com (IR

design tips).

IR2153 also provides a shutdown mode that is used to shutdown its

operation when there is a fault. This mode is also used to stop the

conversion process. In addition to this, the mode acts as excitation master

switch. Refer to section 5.2 for application description.

Operational amplifier

Operational amplifier, op-amp, is one of the key components in analog

design circuits. It is used dominantly in signal translating circuits. A

required op-amp configuration is determined by the application where it is

to be used.

The following op-amp configuration are used in the signal translating

circuits:

Inverting amplifier

Non inverting amplifier

Differential amplifier

Voltage follower

Instrumentation amplifier

R1

R2

R3

ViVo

Com

(a)

R1

R2

Vo

Com

Vi

(b) Figure 3.3 (a) Inverting op-amp (b) non-inverting op-amp

48

Inverting amplifier’s gain is determined by

Av = -R2 3.5 R1

where R2 is a feedback resistor. This amplifier has no common mode error

and temperature drift is reduced by making

R3 = R2 3.6 R1

See figure 3.3 (a).

Non-inverting amplifier’s gain Av is equal to

Av = 1 + R2 3.7 R1

In addition, it has a small common mode error.

See figure 3.3 (b)

R1

R2

R3

R4

dVi

Vo

Com

(a)

Vi

Vo

(b) Figure 3.4 (a) Differential op-amp (b) Voltage follower

A differential amplifier has a common mode problem that can be reduced

by matching the resistance ratios.

49

That is:

R1R4__ = __R2R3__ 3.8 (R1 + R4) (R2 + R3)

and its gain is determined by

Av = R4 3.9 R1

See figure 3.4 (a)

Differential amplifier’s output voltage is proportional to the product of its

gain and the difference between two voltages at its inverting and non-

inverting input terminals (Carr, 1991, p. 249). That is

Vo = Av (V1 – V2) 3.10

Where Av is the amplifier gain.

Differential amplifier circuit is very economical. It needs one IC like LM

741. Although it has the ability to reject common-mode voltages when

resistor ratios are matched, it is not suitable for high gain applications.

This is due to its low input impedance (Carr, 1991, p 250).

An instrumentation amplifier is used to alleviate differential amplifier

problems.

In the case of instrumentation amplifier, R2 and R3 have to be equal to

avoid voltage gain error.

50

Its gain is represented as follows

Av = 1 + 2R3 3.11 R1

provided R2 = R3 = R4 = R5 = R6 = R7 . See figure 3.5.

Besides solving a differential amplifier technical problem, an

instrumentation amplifier presents very high input impedance and high

common mode rejection.

R5

R6

R7

R1

R2

R3

R4

VodVi

Com

Figure 3.5 Instrumentation op-amp

Figure 3.4 (b) shows how an op-amp is configured to operate as unity gain

amplifier, commonly known as voltage follower. An op-amp in this

configuration has an improved output current.

Voltage regulator ICs

Voltage regulators play an important role in keeping voltage constant as

per controller circuit hardware section. Three terminal voltage regulator

ICs such as LM78xx and LM79xx are an on-board mount type and

therefore design with these ICs is simple.

51

Three terminal voltage regulator ICs have all the important features of

series regulators as well as overload protection built into a package. A

fixed voltage regulator circuit is simpler than that of an adjustable voltage

regulator circuit. However, the former circuit is limited to the

manufacturer’s preset voltage level whereas the latter allows the user to

set the voltage level to suit the application.

Fixed voltage regulators have the following advantages that ensure a cost

effective and reliable regulation:

• The ease of use due to few external components requirement

• Reliable operation

• Built in overload protection

• Internal thermal drip

Refer to section 6.2 for further discussion on fixed voltage regulators and

section 4.4 for discussions on high power voltage regulation. The high

power voltage regulation is built around LM723 because it was found to

be common in high power voltage regulation systems.

3.5 Transformer and choke

Ferrite is an ideal material for transformer core operating in the frequency

range 20kHz to 3MHz. The core is gapped to avoid saturation under DC

bias conditions.

ETD core is used in high frequency applications due to their low cost,

ease of assembly and winding. The core is readily available for a variety

of hardware wiring. It gives adequate space for the large size wires. This

allows air to flow through the transformer window. This feature keeps the

assembly cooler.

52

The ETD core center post is round and thus windings have shorter path

length around it , 11% shorter than that of the square post. Therefore, this

reduces winding losses by 11% and enables the core to handle higher

output power (Magnetics, Page 4.6)

Transformer power handling is determined by its WaAc product where

Wa is a core window area

Ac is an effective core cross-section area

The WaAc defined by the power output relationship is obtained by starting

with Faraday’s law:

E = 4BN f Ac * 10 8 (for square wave) 3.12

WaAc = EAW * 10 8 = Po C * 10 8 3.13 4Bfk 4Bfk

k = NAw 3.14 Wa

Where:

E = applied voltage (rms)

B = flux density in gauss

Ac = core area in cm2

N = number of turns

f = frequency

Aw = wire cross sectional area in

cm2

Wa = window area in cm2

C = current capacity in cm2 /amp

k = winding factor

I = current (rms)

Pi = input power

Po = output power

e = transformer efficiency

53

Switching power transformer

Switching power transformers are called either: buck, boost, converter or

inverter depending on the application. These are specified for high

efficiency, small size, and low weight applications. In this application, it

is a converter and it operates from a 24V DC power source that is

switched at 100 kHz. The switched DC power is seen as a square wave AC

at the transformer as used in a push pull configuration. See figure 5.6.

The configuration is efficient as it makes bi-directional use of the

transformer core windings. Therefore provides an output with low ripple.

3.6 Sensors

3.6.1 Hall-effect sensors

Hall-effect sensors offer a non-contact sensing, a high degree of accuracy

and the ability to measure DC and AC currents. Owing to the fact that a

hall sensor presents non-contact sensing, there are no electrical power

losses.

The hall sensor sensitivity is proportional to the number of current

carrying conductor turns around the sensor core.

Hall effect open-loop current sensors have the following advantages over

resistive current sensing techniques:

The sensor has temperature compensation circuitry for stability and

accuracy.

It presents non-contact current sensing hence, there are no electrical

power losses.

The open loop sensors are also preferred in battery-powered circuits

due to their low operation power requirements (Bell, 2001. p 2).

54

3.6.2 Optical and electromagnetic isolators

Opto-couplers

Since mechanical switches introduce a series of narrow pulses in

electronic circuits every time when pressed, this effect contributes

positively towards noise problems in the circuits. Besides isolating the

controller from the electronic circuit, opto-couplers provide the circuit

with a bounce free switching and can only allow signal flow in one

direction.

Opto-couplers use light flux between an LED and a phototransistor to

couple digital signals and analog signals from one circuit to the other.

The coupler current transfer ratio is improved by adding a second

transistor in a darlington pair configuration.

Therefore

Ic = BIb (before adding an extra transistor) 3.15

Ic = (B1B2) * Ib (darlington pair) 3.16

Where B1 is the opto-coupler current gain

B2 is the extra transistor current gain

Although opto-couplers are popular in electronic circuits not all of them

maintain a linear response when the internal infrared diode junction

temperature increases. Nevertheless, the linear opto-coupler (LOC110)

family solves the in-linearity sad-back.

55

LOC110 assures accuracy and linear response from the input to the

output. Unlike 4N25, it has extra phototransistor that operates as a

feedback mechanism to control the diode drive current effect of

compensating for the diode’s non-linear time and temperature

characteristics (Clare, p.2). See figure 3.6.

The common methods in which LOC110 is utilized as an isolating

amplifier are photoconductive and photovoltaic configuration.

Photoconductive configuration is best suited for high frequency

applications. This is because its bandwidth is up to 200kHz. The linearity

and drift characteristics of this configuration are comparable to those of

an 8-bit D/A with ±1 bit error (Clare, p 4). This error is mitigated when

photovoltaic configuration is utilized.

Photovoltaic configuration presents the best linearity, lowest noise and

drift performance. Linearity of up to a 14-bit D/A is achieved in this

configuration although its bandwidth is limited to 40kHz. In spite of the

limitation, the configuration is still favored in the controller application.

This is because only DC analogue signals are processed. LOC110

phototransistors in this mode are 0V biased to eliminate voltage

dependence of a photogenerator and its non-linearity. This way the

linearity improves further (Clare, p 4).

56

Figure 3.6 LOC11X

Figure 3.7 LOC11X V i n vs Vou t

Relay

Relays play an important role where a certain degree of isolation is

required between the controller and the controlled circuit like the fault

load. This becomes a critical factor, especially when the two circuits are

operated from different power sources. Refer to section 6.3.

57

Chapter 4

Charge voltage regulation design

4.1 Overview

The charge voltage regulation played an important role in the controller

hardware. It ensured that the charging voltage and current were

maintained within limits, 28V and 10A respectively. It also protected the

generator against overloads.

Other important aspects were an over voltage load, battery bank

overcharge protection and deep discharge protection. It also protected DC

loads from over-voltage and under-voltage supply. All the protection

techniques ensured robustness of the controller charge unit.

4.2 Uncontrolled rectification

The literature indicated that an uncontrolled rectifier is more cost

effective compared to the controlled rectifier in this application.

Therefore, uncontrolled three-phase bridge rectifier was used to rectify

the generated voltage.

The generated line voltage and the line current waveforms were not pure

sinusoidal waveforms. See figure 4.1a. The waveforms indicated that the

generated line voltage and the line current had some harmonics. The

harmonics were considered as noise to the controller. Therefore, their

effects on the fundamental waveform are discussed in the following

paragraphs.

58

With reference to the line voltage harmonics chart in appendix C, odd

harmonics were dominant in the line voltage. The eleventh harmonic was

4.2% of a fundamental harmonic. Its period was 2.283ms.

Although even harmonics were dominant in the line current, the 19 t h

harmonic was 0.78% of the fundamental current harmonic. Refer to line

current harmonics chart in appendix C. All the information and the

waveforms were captured with Tektronix wave star software for

oscilloscopes.

TT

TT

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 50 Volt 5 ms 2) Ch 2: 10 mVolt 5 ms

Figure 4.1a Line voltage (Ch 1) and line current (Ch 2) waveforms

The waveform displayed in Figure 4.1b indicates that each diode of the

six-pulse bridge rectifier conducted for 120° . That was because each

diode conducted for 60° in each of the two line voltages cycles, like in

VAB and VAC .

59

The waveform showed that average diode current was one-third of the

average load current. See figure 2.8 for the VAB and VAC .

The current could be expressed as:

ID av = IL av/3 4.1

In addition, the RMS value of the bridge diode was

ID R M S = IL av/√3 4.2

T

2 >2 >2 >2 >

2) Ch 2: 1 Volt 5 ms

Figure 4.1b Line current 5A waveform

Figure 4.1b shows line current waveform when the six-pulse bridge

rectifier was connected.

Because the generator is a 1.25kW generator, at 10A load the generated

voltage would be 125V. Although the generated voltage could be much

higher than the charge voltage, a desired charging voltage would still be

maintained at 28V.

60

This was because the excess voltage was diverted to an over voltage load.

Refer to section 4.2.

Figure 4.2 shows that 24V was maintained over a wide range of speed,

from 149rpm to 600rpm. During the time when the line voltage was

maintained the dump load was off. At the time when speed was above

600rpm, the line voltage increased to magnitudes greater than 24V.

Therefore, the dump-load voltage was a difference between the generated

DC voltage and the steady state voltage of 32V.

At speed higher than 600rpm, the generate voltage started to increase.

That was because the exciting current would not be increased any further.

The generator operated in the boost mode for 22.27% of the speed range.

Therefore, the generator reverse exciting current flowed through the rotor

for 77.73% of the speed range while maintaining 24V line voltage. During

boost mode the forward exciting current increased exponentially at a

higher rate than in the buck mode, and so did the power losses.

At 24V the ratio of power losses to the controller input power was higher

than when the voltage was increased to 48V. When the battery voltage

was doubled to 48V, the efficiency of the charge control was increased by

9.31% from 85.89% at full load current (10A). The operation speed range

was increased by 66% although the cut-in speed was changed to 250rpm.

See table 4.1.

The operation speed range was determined by the exciting current.

Therefore, because of an increase in full exciting current a wider speed

range was achieved. A disadvantage of the current increase was the

exponential increase of the armature loses.

61

Open circuit Excitation IV curves at 24V V-line

0

10

20

30

40

50

60

100 200 300 400 500 600 700 800

Speed (rpm)

Vol

tage

s (V

)

-8-6-4-20246

Exc

iting

cur

rent

(A)

Vline (V) Vdc (V) Vrotor (V) Iexc (A)

I => boost mode region II => buck mode region III => Excess voltage region Figure 4.2 Open circuit excitation characteristics curves at 24V line

TT1 >1 >1 >1 >

Figure 4.3 Ripple voltage waveform at 0A load current

T1 >1 >1 >1 >

Figure 4.4 Ripple voltage at 2A load current

T1 >1 >1 >1 >

Figure 4.5 Ripple voltage waveform at 2A load current and 2A excitation

current

CH 1 10V 1mS

CH 1 10V 1mS

CH 1 10V 1mS

62

TT1 >1 >1 >1 >

Figure 4.6 Ripple voltage waveform at 2A load current and 4A excitation

current.

After rectification, some voltage ripple waveforms were also captured and

analyzed. Refer to figure 4.3 up to 4.6. The analyses proved that an

increase in load current and in an exciting current caused a reduction in

ripple voltage magnitude. Although high currents increased the power

losses on board, they reduced an AC component in the charging voltage.

See ripple data in appendix C. The AC component would cause heating of

the battery bank and consequently gassing would take place.

4.2.1 Controller protection

The controller protection was carried out in two ways: by monitoring the

generated DC voltage within hardware and within software. Within

hardware, the protection was achieved by comparing the input charge

voltage tracking reference (V t r c) and the protection reference voltage

(V re f).

With reference to figure 4.7, the comparator generated a positive error

signal when V t r c was greater than V re f . This happened when V i n turned out

to be greater than a desired voltage level, 35V. The error forward biased

D2 and switched Q1 on. Alternatively, a “Low” (approximately 0V) at S4

turned Q1 on, and relay 1 (Rly1) became energized. Refer to section 6.5.

When the relay was energized, the fault load became connected to the

generator and the charge circuit was isolated form the generator. The

process was reversed when V i n was less or equal to 35V.

CH 1 10V 1mS

63

Voltageregulator

Over voltageload control

+15V

+5V

S4Vtrc

Vref

D1

Q1

R1

R2

R3R4

Pot1

Rly1

VinVg

+_

Figure 4.7 Voltage protection circuit

4.3 Over voltage load

An over voltage load (Ov) was an auxiliary DC load to the generator. See

figure 4.9. It only became part of the circuit when VDC exceeded 32V.

That was because Q1 turned off and forced charging current to flow

through Ov. As the current flowed through Ov, voltage VOv developed

across Ov and as a result, power dissipation occurred. The dissipated

power was not seen as a loss to the system. It could be used to reduce

costs in heating systems e.g. water heating system. The voltage across Ov

appeared as an excess voltage hence why the charging current was

diverted to Ov. Therefore

Vg = VOv + VDC 4.3

Where VDC was a sum of all voltage drops across series components and

thus

VDC = VQ4 + VR 2 2 + VD7 + VC g 4.4

In addition, VQ4 was forward VC E of the bypass transistor Q4

VR 2 2 was charge current sensing resistor voltage

64

VD7 was blocking diode D7 forward voltage.

When Vg = VDC

V re f > VDC t rc

the comparator generated a positive error signal whose potential was

divided between R3 and R4 and as a result, Q2 was turned on. That was

because VR 4 was greater or equal to Q2 VB E and would be expressed as:

VR 4 = VQ2 VB E = Vo * R4 = 0.6V (electrical characteristics) 4.5 (R4 + R3)

When Q2 was on, DZ clamped Q1 VGS to 12V such that VR 2 is:

VR 2 = Vg - VDz 4.6

When VDC > 32V

V re f < VDC t rc

and the comparator generated a negative signal that turned off Q2 . When

Q2 was off, there was no current flow through both R1 and R2 hence VR 1

was zero and Q1 was turned off. During that time, VOv was greater than

zero. That holds truth because according to Ohms law VOv was a product

of ROv and the charging current.

Figure 4.8 showed switching characteristics of international rectifier P-

channel MOSFET, IR9640. When the MOSFET gate to source voltage was

12V, its drain to source voltage was approximately zero. During that time,

the MOSFET was fully on. The power MOSFET could effectively be

turned on at voltages higher than it threshold voltage, 4V.

65

IRF9640 Vgs vs Vds

0.0001

0.001

0.01

0.1

1

10

100

0 1 2 3 4 5 6 7 8 9 10 11 12

Vgs (v)

Vds

(v)

Figure 4.8 MOSFET switching voltage levels

Because Ov was in series with Q4 , R2 2 and D7 , i t also dissipated power

that was proportional to a square of a charge current. The power was also

a product of the current and VOv . It was not regarded as a loss because it

could be stored, most commonly in water in a form of heat energy for

future use. Therefore, the only main power losses on board were those of

other series components.

The fact that power in DC circuits was a product of current and voltage

product, the losses were expressed as:

P los s = IC g (VQ4 + VR 2 2 + VD7) 4.7

= 46W

when IC g = 10A (maximum charge current)

VQ4 = 3V (forward voltage (Loveday, 1992. p 63))

VR 2 2 = 1V (R2 2 = 0.1)

VD7 = 0.65V (diode forward voltage)

66

P i n = Pou t + P los s 4.8

= (IC gVC g) + P los s

= 326W

Therefore the minimum charging efficiency was

η = Pou t * 100 = 85.89% (without POv) 4.9 P i n

When Pou t was 280W.

+Vref

Dz R1

R2

R3

R4

R5

R6

Q1

Ov

Vg Vin

VDC

VDC trc

Q2

Vo

Vg

Figure 4.9 Over voltage load drive circuit

For higher voltage battery banks, like 48V, the charging voltage could be

maintained at 54V and the line voltage at 42V. In this case, the input

charging power would be 567.24W and the output power would be 540W

at full load. Therefore, the efficiency would be increased to 95.2%.

At maximum exciting current 104.5W was dissipated into the rotor and

that brought the system efficiency down to 53.8% and 76.86% for 24V and

48V battery banks respectively.

67

The hybrid system efficiency included the stator power losses, 3I2R,

which was 100.49W at full load. That brought down the efficiencies to

22.6%, 55.8% and 79.93% for 24V, 42V and 92.55V respectively.

Although the controller efficiency was optimized, the system efficiency

still had to be improved. It could be improved by changing the generator

to a brush-less generator. That would reduce the generator excitation

power losses by 16.27% and improve the controller efficiency by 19.21%

at 24V.

The fact that the generated voltage dropped linearly with increasing load

current, the current could be used to reduce the generated voltage to a

desired level before excitation process. In that way the speed range would

be increased further. That would be achieved by including a controlled

load that would be used to increase and maintain load current at full load

current. The load current would be a sum of the controlled load current

and the charge current. It could be expressed as:

IL = IC L + IC g 4.10

Power dissipation on the controlled load could still be treated in the same

way as that of an over voltage load.

Beside the improvement in the controller efficiency, at 42V line voltage

the speed range could be increased by 66%. Refer to table 4.1.

68

24V Battery bank 48V Battery Bank

Speed in RPM 149rpm – 598rpm 250rpm – 1000rpm

Charging efficiency

without excitation 85.89% 95.2%

Charging efficiency

with excitation 53.8% 76.86%

Table 4.1 24V and 48V charging systems comparison table

The fact that VOv was outside the voltage regulation window, it increased

with increasing excess voltage ∆VOv above the desired input charging

voltage and vice versa. Ideally, IC g was equal IC t ot a l. A difference between

the two currents was very small due to very high shunt resistors like: R5

and R6 .

Therefore, the total charging efficiency could be expressed as

η t o t a l = Pou t * 100 = (IC g (VC g + ∆VOv)) – P los s) * 100 P i n (IC g Vg)

Determine Ov, Ov = VOv max IC g

4.4 Charging voltage regulation

Linear voltage regulator LM723 was used to maintain charging voltage

VC g at 28V irrespective of fluctuations in an unregulated input voltage,

V i n C g . A 12V zener diode was used to fix regulation reference voltage at

11.2V.

69

With reference to figure 4.10, R4 and R5 were used to determine the

reference where by it was expressed as:

V re f = VR 5 = VDz * R5 = 11.2V 4.11 (R5 + R4)

where VDz was 12V

It would have been simpler to just short circuit LM723 pin 5 and pin 6

and thus V re f would ideally be equal to VDz. That would be when the

internal reference voltage, 1.2V is discarded. Therefore, that implied that

then the reference would be 13.2V.

At switch on, VC g increases from zero volts level to the charging voltage

level, 28V. To ensure that 28V level was not exceeded with further

increase in the input voltage, the regulator internal comparator (Loveday,

1992, p. 99) used a potential divider network to monitor VC g to generate a

regulation signal. The regulation signal was positive to allow charging at

the correct voltage level. The signal only changed state to negative when

VC g exceeded 28V.

Figure 4.10 shows how R3 and VR were connected to a regulator. They

were connected such that VVR kept track of VC g . When VC g was equal to

28V, VVR was also equal to V re f . Every time when VVR turned to be either

greater or less than V re f , the comparator would generate the regulation

error signal that was either negative or positive respectively. The signals

determined the state of the bypass darlington-pair transistors. The

negative signal turned off the transistor pair and the positive signal turned

the transistors on.

70

R3 was assumed to be 20k and VR was determine as:

VR = VVR * R3 (VVR = V re f) 4.12 (VC g – VVR)

VR = 13.31k (when VC g = 28V)

= 6.09k (when VC g = 48V)

Refer to LM723 data sheets in appendix D.

4.4.1 Overload protection

Overload protection employed foldback current limiting technique. The

foldback current limiting was found to be a suitable protection for the

application. It prevented excess heat dissipation on the series components

when the output was short-circuited and when there was an overload.

During the short circuit and overloading of a charge circuit, the knee

point current was exceeded. The charging voltage dropped to zero when

the output was short-circuited. At short circuit, Isc was limited to a value

lower than the rated charging current.

1 2 3 4 5 6 7

891011121314LM723

Rsen

R1

R2

R3R4

R5VR

Dz

Vin VCgQ1Q2

Figure 4.10 Battery bank voltage regulator

71

V sen was given as

V sen = IC g R sen – (VC g + IC g R sen). R1 /(R1 + R2) 4.13

Therefore IC g = V sen(R1 + R2 + VC g R1) 4.14 R sen R2

And thus

Ik = 0.6(R1 + R2) + VC g R1 4.15 R sen R2

Where Ik was the knee point current and V sen was 0.6V.

When the output was short circuited charge voltage VC g was zero and

Ik = Isc .

Therefore

Isc = 0.6 (R1 + R2) 4.16 R sen R2

(Loveday, 1992, p. 84)

When VC g = 28V, IC g = Ik = 10A, R sen = 0.1Ω

R2 = 71.5 R1

R1 = 1k and R2 = 71.5k 75k (1/4W series metal film resistors).

The Ik = 9.813A and Isc = 6.08A.

During short circuit, Ik was reduced to 61.96 % of its maximum value.

The power dissipation on the series components also was reduced

proportionally.

72

That was because in DC circuit

P = IV = I2R. 4.17

4.5 Battery protection

The battery bank was protected from overcharging and deep-discharge by

means of a voltage window comparator. The comparator monitored the

bank voltage continuously to ensure that the required voltage level limits

were not exited (Loveday, 1992, p. 140).

Battery bank window comparator

With reference to figure 4.11 an output changed state, from (High) to

(Low), when Vb a t was outside a defined battery bank voltage limits (Vb mx)

and Vb mi n). That was true when S2 was Low. DC load could be

disconnected with a micro-controller via S2. The limits were set within

software hence a similar operation like the one described in the next

paragraphs, could be achieved with a micro-controller. However, with the

micro-controller the load could be disconnected irrespective of whether

Vb a t was within limits or not. That was achieved by setting S2 High within

software. The output (L-disc) stayed Low while S2 was High. DC load

was disconnected from the battery bank when L-disc was Low and was

connected back when L-disc was High.

When Vb a t was greater than Vb mx , (A) was High and (B) was Low. When A

was High diode A was forward biased and diode B was reverse biased.

Diode B was reverse biased because op-amp B was current sinking and

there was no current flow because the diode B state. That way (C) would

be High.

73

Vb mi n < Vb a t > Vb mx C = High L-disc = Low (S2 = Low)

When Vb a t was less than Vb mx and was greater than Vb mi n both A and B

were Low. Therefore both A and B diodes were reverse biased. Then

leakage currents flow through the diodes. During that time, C was Low

and L-disc was High.

Vb mi n < Vb a t < Vb mx C = Low L-disc = High (S2 = Low)

+

+

Vbmx

Vbmin

Vbat L-disc

A

B

S2

C

Vbmin Vbmx

Ouput (L-disc)

Battery bank voltage (V)

OFF OFFON

Figure 4.11 Battery bank voltage monitor

During the time when Vb a t was less than both Vb mi n and Vb mx , A was Low

and B was High and thus B diode was forward biased and A diode was

reverse biased. As C was the sum (A+B=C) of A and B, it was High when

either A or B was High. Therefore, L-disc was Low.

74

Chapter 5

Excitation circuit design

5.1 Overview

This chapter is focused on excitation of the rotor with the main accent on

the exciting current and the induced emfs. The emfs were induced because

of a rotating flux in the generator. Therefore, the excitation power supply

had to overcome back emf before controlled excitation could take place. A

freewheeling diode allowed continuous current flow into the rotor due to

the back emf.

The induced line emf was directly proportional to the generator speed and

the air gap flux density. So, the fact that the flux density was proportional

to an exciting current and the speed the emf could be expressed as

ea = Emax sin ωt

and the rms value of the Emax was

E rms = 4.44 f NΦ p KW

Where f was the frequency in hertz

Φ p was the flux per pole

N number of rotor series turns

KW was the winding factor

(Sen,1997, pp. 215-216)

75

Static operation

This section entails studying the effects of exciting current on the rotor.

Static exciting flux test per-pole was performed. The test results proved

that an increase in exciting current caused a proportional increase in the

exciting flux density in the generator air gap. Conversely, a decrease in

the exciting current caused a proportional decrease in the generator air

gap flux density. See figure 5.1.

Static air gap flux density

0.002.004.006.008.00

10.0012.0014.0016.0018.00

-6 -4 -2 0 2 4 6

Exciting current (A)

Air

gap

flux

dens

ity (m

T)

Pt1 AvPt2 AvPt3 AvPt4 AvPt5 Av

Figure 5.1 Static air gap flux density

The flux density responses indicated in figure 5.1 showed clearly that the

air gap flux density could be varied to either boost or buck the generator

operation, and as a result, vary the line voltages accordingly.

Dynamic operation

In the preceding section, it was proven that the air gap flux density

changed proportionally with the exciting current. The similar flux density

response also existed in the dynamic operation. It was confirmed that a

76

change in exciting current had a direct influence in the generated voltage.

Therefore, the exciting current not only had a proportional influence on

air gap flux density but also had a similar influence on the generated

voltage. See figure 5.2.

Dynamic air gap flux density

0

2

4

6

8

10

12

-6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6

Exciting current (A)

Flux

den

sity

(mT)

149 rpm

264 rpm

374 rpm

598 rpm

750 rpm

Figure 5.2 Dynamic air gap flux density

During dynamic operation, air gap flux density was increased by 69.98%

above permanent air gap flux density. Conversely, the flux density was

reduced by 57.63%. That was the exciting flux density was 42.97% of the

permanent exciting flux density.

77

5.2 DC/DC converter

Exciting current was supplied from a 24V battery bank with a DC/DC

converter. The converter did not only convert DC but also isolated the

exciting current circuit form the battery bank.

DC/DC converter converted the 24V of the battery bank in to a 48 VDC

source that supplied a PWM buck chopper. The chopper controlled the

armature exciting current. The converter was operated at 100 KHz to

reduce its transformer size and filter size. It gave lower ripple voltage on

the output. The basic switch mode power conversion had evolved from

basic PWM converter to soft-switching converter (Bordry, Dupaquier)

because of reduced switching losses.

Switching converters, in particular buck and boost converters were known

to be the worst EMI generators due to their pulsating input current. That

was due to a switching action of their semiconductor switches. However,

the EMI was reduced with an EMI filter and by making use of soft

switching converter (Bordry, etal).

Soft switching advantages were:

• Reduced switching losses.

• The improved reliability due to reduced high voltage and current

stresses.

• A limited frequency spectrum, which was an advantage with respect

to EMI and losses in passive components.

• A reduction in size of the components resulting from higher

switching frequency.

Figure 5.3 was a DC/DC converter schematic diagram. On the input stage,

it showed a 12V driven PWM self-oscillating MOSFET driver IC, IR2153,

with external components.

78

An opto isolator phototransistor ensured that transistor Q sd shutdown

when opto-diode was on. RT and CT were the timing components.

HO

LO

VB

VS

VCC

RT

CT

Com

12 V

PWMGen

Excitationpolaritychanger

Ra

La

Ea

Dz

Dbd

Dfd

Q1

Q2

Q3

Qsd

opt 24V

Inductance (48uH / coil) Figure 5.3 Rotor (armature) excitation schematic diagram

IR2153 shutdown when VC T was less than the threshold voltage, Vcc / 6.

During that time both outputs, HO and LO , were set low within minimal

delay.

IR2153D presented zero switching losses. It eliminated cross conduction

by providing enough dead time during alternate switching of MOSFETs,

Q1 and Q2 . See figure 5.5. During that time both HO and LO outputs were

low. Dead time was fixed inside the IC. In addition to the mentioned

advantage, the dead time helped to maintain zero voltage switching and

soft switching (IR design tips).

The switching frequency was determined as:

fs = 1 5.1 (1.38 CT (RT + 75Ω)

79

CT and RT could be determined form figure 5.4 for the operation

frequency of 100kHz as used by Magnetics to test P type ferrite

transformer cores.

Figure 5.4 IR2153 RT vs frequency curves

T

T

1 >1 >1 >1 >

2 >2 >2 >2 > 1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us

dX: 1.17 us X: 5.17 us

Figure 5.5 DC/DC converter drive signals, signal 1 for Q1 signal 2 for Q2

80

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us

Figure 5.6 DC/DC converter transformer primary signals with respect to

battery bank positive terminals.

T1 >1 >1 >1 >

1) Ch 1: 20 Volt 2.5 us Figure 5.7 DC/DC converter transformer secondary signal

The transformer efficiency at full excitation was 80%. To achieve the full

excitation (104.5W) the battery bank delivered 130.62W.

81

The fact that general-purpose diodes have poor response at high

frequencies, like 100kHz, schottky diodes were used to rectify DC/DC

transformer secondary voltage.

T

1 >1 >1 >1 >

1) Ch 1: 10 Volt 500 ns

dX: 800 ns X: -50.0 ns

Figure 5.8 DC/DC signal overshooting and ringing

Output pulse shape

At switch ON the voltage rose rapidly from zero volts, overshoots to a

peak voltage and then settled back to normal pulse height, which was the

pulse voltage. The pulse voltage was equal to a DC power supply voltage.

A rapid pulse voltage rise time resulted in some oscillations and ringing

at the pulse maximum voltage. Those appeared before the voltage settled

back to normal pulse voltage. See figure 5.8. The oscillations became a

serious problem when switching MOSFET pair was not matched. Refer to

DC/DC converter waveforms in appendix C.

The pulse rise time was defined as the time required for the voltage to rise

from 10% to 90% of the peak voltage. Instantaneous change in voltage,

82

dv/dt, was a slope of the voltage rise in volts per microseconds. It was

approximately 80% of the peak voltage divided by the rise time

(ABB,1998, p 11). Therefore, it could be expressed as:

dv = Peak voltage * 0.8 5.2 dt t r

T

1 >1 >1 >1 >

Figure 5.9 Open circuited rotor back emf waveform

The waveform in figure 5.9 was captured across an open circuited rotor

terminals. The waveform showed that the exciting flux also induced emf

into the rotor coils. Likewise the generated voltage, back emf increased

with increasing generator speed. It also increased with increasing load

current. It was therefore evident that a blocking diode and a freewheeling

voltage clamping diode were required to protect the exciting circuit

against back emf. The diodes assured reliability of the circuit. Refer to

appendix C for back emf analyses data.

T1 >1 >1 >1 >

Figure 5.10 Freewheeled back emf waveform

83

With a freewheeling diode included in the exciting current circuit, the

negative half of the back emf was clipped off. The negative overlap was

due to the freewheeling diode forward voltage. See figure 5.10.

TT1 >1 >1 >1 >

Figure 5.11 Zener clamped freewheeled back emf waveform

When a zener diode was connected in parallel with a freewheeling diode,

as indicated in figure 5.3, the waveform in figure 5.11 appeared across the

rotor terminals. The waveform showed clipped positive pulses. The

maximum magnitude of the pulses was determined by the zener diode. It

was high enough to drive 5A into the rotor. The zener diode not only

limited the exciting voltage but also protected both the blocking and the

freewheeling diode from being driven into avalanche by the positive

pulses of the back emf.

DC buck chopper

A DC buck chopper was used to adjust the exciting current to the required

levels in order to adjust air gap flux density of the generator. The chopper

was found to be more energy efficient compared to a DC boost chopper

(Beak et al. , 2001). The project served as one of many applications of the

power electronics choppers. (Ang, 1995, p. 357)

A buck chopper DC input was pulsed to an armature in a manner that it

increased and decreased the exciting flux density in direct proportion to

the armature current. Duty-cycle D determined average voltage across the

84

armature terminals. During the chopper off state, the exciting current

continued to flow into the armature because of the induced back emf. The

emf forward biased the freewheeling diode. Therefore, the diode

conducted back emf current back into the armature. Therefore during that

time the armature current (exciting current) Ia was equal to the diode

current Id .

EaLaRcRloss

Ra

Figure 5.12 rotor circuit

PWM

Ra

La

Ea

Dfd

DbdQ

Vin

Figure 5.13 Exciting current control circuit

The field circuit total resistance, as shown in figure 5.12, was a sum of

the brushes resistance, slip-rings resistance, contacts resistance and the

field coil resistance. The first three resistances caused excitation power

loss. They were defined as R los s , therefore the excitation power loss was

expressed as:

Ploss = Ia2*Rloss = 17W

85

The field coil resistance (RC) determined the actual excitation power. The

excitation power was expressed as:

Pex = Ia2*RC = 87.5W

With reference from figure 5.13, the required input power was determined

by Ra as:

P i n = Ia2*Ra = P los s + Pex = 104.5W

Therefore the field circuit efficiency was

η = Pex* (P i n) -1*100 = 83.73%

It was assumed that DC/DC buck chopper would operates in two modes,

which were continuous current conduction and discontinuous current

conduction. The two modes would come in to perspective respectively

when

Tof f < Td s Td s = (-dia / dt) * Tcg 5.3

and

Tof f > Td s .

where Td s was discharge time and Tcg was charge time

Voltage and current continuous mode

With reference to figure 5.3, when Q3 was switched on, the armature

current would be increased exponentially to Ia max due to the armature

86

inductance. Similarly, when Q3 was switched off, the current would

decayed exponentially to Ia mi n . Ia could not decay to 0A because T s was

much smaller than the armature time constant La/Ra . The turn on and turn

off, Ton and Tof f , respectively would be given by:

Ton = DT s 5.4

and

Tof f = (1 – D)T s 5.5

where

T s = 1/fs , ω = 2πf 5.6

During Ton , Db d would be forward biased and D fd reverse biased therefore

Ia would flow into the armature and VLa would developed because of La

dia/dt. During Tof f similar characteristics would exist because of decaying

Ia . During the decay time, VLa polarity would change and thus had D fd

forward biased. When D fd was forward biased, IDfd would be equal to Ia .

VLa = Ea where Ea is an armature emf

Figure 5.14 is a graphical representation of the preceding description of

how an armature current would respond in continuous mode.

87

TON

TS

TOFF

Imax

Imin

Ia

iin

iD

Vi - Vo

Vo

VL (avg)

Continuous mode

TON = DTSTOFF = (1 - D)TS

Vo

Figure 5.14 Continuous mode waveforms

The average voltage across the generator armature terminals would be

Va = V s DT s = DV s (V) 5.7 T s

During interval 0 ≤ t ≤ Ton

V s = Ra ia + L (dia/dt) + Ea 5.8

An average current through a generator armature coil would be

ia = (V s – Ea) (1 – e – t / τa) + Ia mi ne – t / τa 5.9 Ra

88

τa = La / Ra

During interval Tof f ≤ t ≤ T s

Va = 0 = Raia + La (dia/dt ′) + Ea 5.10

where

t ′ = t – Ton = t - DT s 5.11

Ia = -Ea/Ra (1 – e – t ′ / τa) + Ia max e - t ′ / τa 5.12

In steady state, voltage across La would be zero because VLa = Ldia/dt and

di/dt = 0v

Therefore

Va = DV s = RaIa + Ea 5.13

Ea = K ′E φƒ ω = DV s – IaRa 5.14

Ia = (DV s - Ea) / Ra 5.15

(Rahman, 2002. pp. 9-10)

89

TON

TS

TOFF

Imax

Ia

ia

iD

Vi - Vo

Vo

VL (avg)

Discontinuous mode

tr

TON = DTSTOFF = (1 - D)TS

t

Ea

Vo

Figure 5.15 Discontinuous mode waveforms

Discontinuous mode would take place when armature current becomes

zero during switching period T s . Then Q3 would be off and a freewheeling

diode would be on. Va would then be equal to back emf across the

armature terminals. Refer to figure 5.15.

An armature current would drop to almost zero amperes at time tϒ

(corresponding to the conduction angle ϒ). That would be due to the back

emf induced into the armature by the rotating flux in the generator. The

armature voltage would therefore be:

Va = V s for 0 ≤ t ≤ T s

≈ 0V for DT s ≤ t ≤ tϒ

= Ea for tϒ ≤ t ≤ T s

90

The average armature voltage would be

Va = DV s + (1- (tϒ / T s)) 5.16

During 0 ≤ t ≤ DT s

With discontinuous mode armature current would start form zero because

Ia mi n would be zero. Therefore

ia = V s – Ea (1- e – t / τa) 5.17 Ra

During freewheeling Ia would be

ia = -Ea (1 – e - t ′ / τa) + Ia max e - t ′ / τa 5.18 Ra

And at tϒ ia i t would be approximately equal to zero.

An armature inductance La and resistance Ra determine the conduction

time constant τa for a period during which ia flowed through the armature.

τa = La/Ra

The time constant formula shows that the higher the inductance the lesser

the chances of discontinuous armature current (Rahman, 2002, p. 10). The

inductance also provided current smoothing. It limited Ia peak-to-peak

ripple value to lower losses that lead to the generator derating (Beak,

Buddingh, 2001, p.1162). Therefore the minimum inductance required to

ensure continuous mode when Ton < Tof f was determined by

La mi n = Tof f * Ra/2 5.19

91

because the chances of discontinuous mode would be high when

Ton ≈ τ = 250us then La minimum would be:

La mi n = 522.5uH

Cosequently, it was obvious that the generator would only operate in the

continuous mode. The armature inductance was much higher than the

minimum required inductance. It was 40.9mH.

Armature power losses were expressed as:

P = Ia2 Ra + Ea Ia 5.20

EaIa represents power that includes mechanical power, friction and

windage (Rahman, 2002, p. 11)

T

TT1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 100 mVolt 250 us 2) Ch 2: 5 Volt 250 us

Figure 5.16 Continuous mode waveform at 30% duty cycle

Figure 5.16 showed the armature current ripple waveform, top waveform,

and the transistor voltage waveform, bottom waveform. The current

waveform indicated that although the duty cycle was 31.39% at 1.23A, the

armature response was still that of continuous conduction. This was

92

because the armature inductive reactance, 2πfLa was much greater than

the armature resistance. TS was also much smaller than the time constant

τa .

5.2.1 Transformer Selection

The choice of a DC/DC converter transformer was such that the losses

would be kept as low as possible. It was done to ensure high efficiency

and reliability of the converter. Although the transformer efficiency was

assumed to be 90%, (Magnetics, p 4.3), it was limited by the transformer

temperature rise and the core saturation.

Ferrite core transformer offered low core losses and high saturation flux

density when it was operated below ±2000 gauss (Magnetics, p 4.4). This

ability made the ferrite transformer suitable for high power and high

temperature operations. From core material point of view, materials with

negative temperature coefficient were more suitable because their

maximum saturation flux density decreased with temperature rise. For

example, P material core losses decreased with increasing temperature up

to 70°c and an R material up to 100°c (Magnetics. p 4.2).

Although core losses increased at frequencies above 20kHz, they could be

reduced by operating with the core flux level lower than ±2000g. Refer to

flux density vs frequency curve in appendix D for reduced gauss as per

frequency level.

The ETD59 transformer core was used for DC/DC converter as it was

decided. That was because the core and wound bobbin were readily

available. The physical dimensions and magnetic data tables in appendix

D were used to determine the core WaAc .

93

Although the transformer was readily available, its windings had to be

tested to see if they meet minimum DC/DC transformer test requirements.

The core typical power handling capacity was 2500watts and operation

flux density was 900 gauss at 100kHz. Refer to ferrite core selection table

in appendix D. The primary turns Np and secondary turns N s were

determined as follows:

Np = Vp * 108 = 6.7 turns 5.21 4BAwpf

Where Vp was 24V and V s was 20.2V at 5A Is

N s = V s * Np = 5.6 turns 5.22 Vp

Ip = Po = 4.838A 5.23 eVp

Where Po was the excitation power, 104.5W, and the efficiency η (e) is

0.9. N s is multiplied 1.1 to allow for losses (Magnetics. p 4.9).

Because of transformer windings test, it was found that the transformer

could still be used except that its ratio was 1:2. Therefore, the ratio

limited the PWM signal to 50% duty cycle. This was because the

maximum required Va was 20.2V to obtain ±5A armature excitation. Refer

to open circuit test at 24V line voltage table in appendix B.

Due to the transformer ratio test V s would ideally be 48V and Ip would be

10A.

94

5.3 Exciting current control

The exciting current control was based on PWM technique whose duty

cycle was proportional to dv/dt of the generated DC voltage. Figure 5.17,

was a block diagram of the exciting current control circuit. The PWM was

generated because of comparison between the ramp signal and the

generated voltage difference, dvDC to the pre-set voltage reference. The

difference was proportional to the generated voltage and the wind speed.

Therefore, the duty cycle would vary with a change in the voltage

difference.

The PWM ramp signal was generated with a 555 timer and the signal was

then fed to a non-inverting amplifier. The amplifier magnified the ramp

signal as shown in figure 5.17 . In that way, the ramp could be adjusted to

suit the PWM requirements of the current control.

With generated voltage difference, an instrumentation amplifier was used

as it presented easy gain adjust with reduced CMRR. Either positive or

negative voltage difference could be obtained from the amplifier. The

positive difference was fed to a non-inverting amplifier whilst the

negative was fed to the inverting amplifier. In that way, a comparator

only received positive voltage difference that compared to the ramp signal

easily to generate pulses.

RampGenerator

Noninvertingamplifier

Inst.Dif.

Amp.

Vg >

Vg <

Non inv.amp

Inv.amp

Signalblocking

diode

PWMpresetlimit

Excitationbuck and

boostchopper

+

SignalComparator

Vg+

+

+

Vref

Figure 5.17 PWM generator block diagram

95

Figure 5.18 shows a graphical representation of voltage difference

inversion as compared to a ramp signal. According to the foregoing

description, a negative voltage difference was inverted whilst dv/dt was

left negative. Therefore, it meant that the PWM translating circuit

operated in the first and second quadrant.

+ dVdc

- dVdc

Non inverting

Inverting

dv/dt-dv/dt

DC 3

-DC 3

100%100%

0%

Figure 5.18 Characteristic behavior of the PWM generator

Figure 5.19 showed how varying duty cycle is achieved at different

voltage levels. DC1 level sets the duty cycle D to 25% of PWM period.

That is only 25% of the input power is deliver to the output. DC2 level set

D to 50% and DC3 level set D to 75%.

The duty cycle also had a pre-set limit. The limit was dependent on the

requirements of the application. The PWM duty cycle limiter was

positioned just before the chopper to make limit setting easy and simple.

However, it would still be possible to place it before the comparator,

although the circuit would be complex.

96

25%50% 75%

0% 100%

DC 1DC 2DC 3

Ramp signal

PWM signal

+dVdc

Figure 5.19 PWM duty cycle levels at different DC levels.

CTL5

GN

D1

OUT 3

RST4

THR6

TRIG2 DISC 7

U1

R1

R2

Vin1

GN

D2

Vout 3RG1

Vin1 GN

D2

Vout 3

RG2

C1

C2 C3

C4

+B

-B -10

GND

VCC

-B

GND

R3

R4

R5

VR1

Q1

C5

C6

VCC

GND

GND

VCC

VR2

R6

3

21

411

U2A

5

67

U2B

10

98

U2C

12

1314

U2D

VCC

-10R7

R8

U5

Q2

GNDR9

VR3

VCC

GND

R22

VR4

VR5

R10

R11

R12

R13 R14

R15

R16

R17

R18

R19

R20

R21 3

21

411

U3A

5

67

U3B

10

98

U3C

12

1314

U4D

3

21

411

U4A

5

67

U4B

GND

GNDGND

-10

VCC

VCC

-10

LK

LK

-B

+15V

VgP

+15V

+PWM

-PWM

R8a

D1

D2

1

2

J1

DZ1

Figure 5.20 PWM generator schematic diagram

The PWM generator was divided into two sections: ramp generator and

DC level generator. The ramp signal and the DC level signal were

compared to generate the PWM signal.

The ramp generator was build around the LM555 timer. LM555 timer is a

highly stable device for generating accurate ramp oscillations. It was for

97

this reason that the LM555 timer was selected for generating a linear ramp

signal. The ramp signal was amplified with a non-inverting amplifier and

was then fed to a voltage follower. After the voltage follower, the signal

was compared with a DC level. The comparator output went high every

time when DC level was greater than the ramp signal. The ramp period T

was determined as

T = 2/3 Vcc VR1 (R3 + R4)*C5 = 0.5mS 5.24 R3 Vcc – VB E (R3 + R4)

Therefore its frequency was

F = T -1 = 2KHz 5.25

The DC level was generated at a voltage difference between the generated

voltage ratio and a preset voltage reference. The difference was amplified

with an instrumentation amplifier. It was also amplified with an inverting

and non-inverting amplifier afterwards. That depended on whether the

difference was negative or positive. The inverting amplifier inverted the

negative difference to positive difference as shown in figure 5.18.

When the ramp signal level was greater than DC signal level, the

comparator U2C generates negative error signal that reverse biased U5

LED. Therefore coupling light flux was zero and U5 transistor stayed off.

Because the transistor was off the output pulse level was low. Conversely,

a positive error signal was generated and then an LED was forward

biased, hence the transistor was turned on and the output pulse level was

high. Because of the comparison, square pulses were generated. See figure

5.21. The DC level determines the pulse on time. Therefore, the pulse on

time was proportional to the DC level.

98

A final stage of the PWM generator was the PWM limiter. The limiter

compared the ramp signal level with a preset voltage level. The level was

set such that it was less than Vcc. At every instant of time when a ramp

signal exceeded the reference comparator U2D generates a negative error

signal that turned off transistor Q2 and thus U5 LED stayed off

irrespective of U2C output. Jumper J1 could be used to overwrite Q2

effects on the LED.

Figure 5.21 PWM generator signal waveforms

5.3.1 Exciting current polarity changer

The polarity of an exciting current was changed with a double-pole-

change over relay. The relay home position was in reversed excitation

mode. That was indicated in figure 5.22. When a relay coil was energized,

the relay poles change their positions from a1 to a2 and b1 to b2 . When the

99

poles positions changed the exciting current polarity changed from

forward mode to reverse mode.

DC chopper

Armature(rotor)

Relay coil

a1

b1

a2

b2

+

Figure 5.22 Exciting current polarity changer

Positioning of a freewheeling diode and the clamping diode was such that

the change in polarity did not affect their purpose in the exciting current

circuit. For instance, when the diodes were placed after the relay, they

would short circuit the exciting current when its polarity changed from

reverse to forward mode. Therefore, the diodes were placed before the

relay. See figure 5.3.

100

Chapter 6

Control Circuit design

6.1 Overview

The control circuit is a very integral part of the controller hardware. This

is because it continuously monitors an operation of each section of the

controller hardware. It reads analog and digital signals generated from

other hardware parts and compares them, within software, to the preset

control references. In addition, there after generates control command

signals. So, read and write control commands as well as interface circuit

characteristics are discussed in this chapter.

6.2 I/O circuits

Input and output circuits brought about communication between a micro-

controller and the rest of the controller hardware. Their main function was

to translate analog and digital signals between the micro-controller and

other circuits. The function of these circuits was to ensure that read signal

magnitudes were kept within safe operation limits of the micro-controller.

The control circuits, for both analog and digital, were powered from on-

board power supply. 5V, 12V and 15V power supply circuit operations

are explained in the following paragraphs.

With reference to figure 6.1, V i was the battery bank voltage, which was

24V. The three power supplies were chain connected, beginning with the

101

15V power supply. They were the same in the way they operated except

that their input and output voltage levels were different.

Three terminal voltage regulators were used to supply 5V, 12V and 15V.

Their current handling abilities were increased with a high current bypass

transistor, Q2 . Q2 protected the regulator from back emf when V i was less

than VO . That was due to the fact that Q2 base-emitter junction presented

feedback path. R2 was connected in series with the voltage regulator such

that VR 2 was proportional to the regulator input current Ireg . Therefore

VR 2 = IregR2 6.1

The value of R2 was determined to ensure that Q2 turns on at any stage

when Ireg tended to exceed a predetermined Ireg . Q2 turns on when

VR 2 ≥ Q2 Vb e = 0.58V (BD436 data sheets page 2)

and output current IO was a sum of Q2 current IQ2 and Ireg . Because Q2 was

outside regulation protection loop, it required its own current limiting, Q1

and R1 . Where R1 is designed to turn Q1 on when

VR 1 ≥ Q1 Vb e = 0.65V (2N3906 data sheets page 2)

This occurred when IQ2 was equal and greater than a predetermined

current value, and for the fact that VR 1 was proportional to IQ2 .

VR 1 = IQ2R1

When Q1 was on, VR 2 dropped to about Q1 Vce sa t , which was 0.2V (2N3906

data sheets page 2). That was because Q1 short circuited R1 . At that

moment, Q2 turned to switch off and thus VR 1 dropped to a value less than

102

Q1 Vb e, hence Q1 switched off and VR 2 was equal to Q2 Vb e though it was

dependent on Ireg . This process repeated itself while IQ2 turned to exceed

its preset value. Therefore, in this way IQ2 was kept within limit.

R2 value was kept the same in all three voltages regulation but R1 value

was determined by the required output current when Ireg was maintained.

RegVo

Vi

0V

R1

R2 C1 C2 C3 C4

Q1

Q2

Figure 6.1 Three terminal voltage regulator with by pass transistor

One other important aspect in I/O circuit was ADC input voltage level

protection. It was achieved by using two diodes as it was indicated in

Figure 6.2. When Van a log i n was greater than Vcc, D1 became forward

biased and thus pulling VADC i n to Vcc. Therefore a micro-controller input

pin was protected from over-voltage. When Van a log i n was below 0V, D2

became forward biased and thus pulled VADC i n to 0V. Because of ADC in

pin protection, Van a log i n could only fluctuate between Vcc and 0V.

R1

C1

D1

D2

Analog in ADC in

Vcc

0V

Figure 6.2 ADC voltage protection

103

Active low pass filter

Because generated voltage was not free of noise, a micro-controller as one

the noise sensitive components was protected against the noise by

introducing active low pass filter circuits in the signal translating circuits.

The filter was a high performance filter. It was built with an op-amp as

active element. It needed few external resistors and capacitors. The filter

circuit was compact, because it did not need inductors, and therefore it

cost less than a passive filter. See figure 6.3.

R1 R2

R3

R4

C1

C2

Vo

Com

Vi

Figure 6.3 Active low pass filter

An op-amp in active filter provided gain so that there was no insertion

loss over the pass band. It also provided good isolation because it has

high input impedance and low output impedance.

A type of filter shown in figure 6.3 was a Sallen-Key filter, and it

consisted of two RC networks in its input circuit and the feedback loop.

Its roll-off was twice as much as of a single pole whose roll-off was

-20db/decade, so the filter roll-off was –40db/decade. R1 and R2 , C1 and

C2 are matched respectively, so the filter cut off frequency was

determined by

FC = __1__ = 60Hz 6.2 (2πRC)

104

A disadvantage was the frequency limitations of an op-amp. General-

purpose op-amps can only be used to filter audible frequencies. In this

application, the disadvantage could be ignored as the cutoff frequency

was within audible limits. See figure 6.4. The figure showed a response

curve that was very similar to that of Butterworth filter. R3 and R4

determined the filter damping factor that was given by

DF = R3 / R4 +1= 1.556 6.3

Low Pass Filter VF Curves

0

0.5

1

1.5

2

2.5

3

3.5

0 30 60 90 120 150 180 210Frequency (Hz)

Vol

tage

(V)

Voltage

Figure 6.4 60Hz active low pass filter characteristics.

Generator speed sensor

A generator speed sensing could have been simpler if the generated

voltage ripple was not dependent on the charging current (load current)

0.707Vmx

105

and the exciting current. An opto-electronic sensor was used to determine

the generator speed from the generator line voltage.

Figure 6.8 was the generator speed sensor schematic diagram. By

monitoring line voltage (V-line), a pulse was generated every time a

revolution was completed. Thus, every time a negative half cycle of the

voltage started a pulse was generated.

Terminals, +RL and +BL, were connected across red line and blue line

such that a fraction of V-line appeared across R2 6 . VR 2 8 appeared to be a

half rectified waveform that was fed through into the circuit by the

differential amplifier. The input pins of a differential amplifier were

protected against high voltage with a zener diode Dz2 . The waveform was

compared with a preset reference voltage that was set within hardware

with VR7. At every instant when the waveform magnitude exceeded the

reference voltage, the comparator generated a pulse as indicated in figure

6.9. The pulses were then fed optically through to a digital counter that

converted them into digital data that could be understood by a

microcontroller.

The counter was synchronized with a micro-controller within software so

that number of pulses read in one-second equals number of hertz. This

was achieved by sending a low pulse to the counter pin 1. The pin is an

active low pin and it cleared the counter outputs every time it received a

low pulse. Then the counter started counting the speed pulses, and after

one second, the pulse count was read with the micro-controller. The

process was repeated every time before reading the speed count.

An octal high voltage high current darlington transistor arrays, ULN2803

was used to interface a counter to a micro-controller.

106

The arrays had freewheeling clamp diodes for transient suppression. This

feature presented an added advantage of using ULN2803. Refer to

ULN2803 data sheet in appendix D.

V-line sensor Low pass filter and the pulse generator

1 2 3 4 5 6

R25

R26

R27

R28

R29

R30

R31

R32 R33

VR6

3

21

411

U6A

5

67

U6B

10

98

U6C

12

1314

U6D

C10

C9

R34

R35

R36

R37R38

VR7

COM

COM +4.5 COM

+4.5

-4.5

+BL

+RL

U7

+5V

-B

PLC

R23

R24

C7

C8

+4.5

-4.5

COM

D3

DZ2

DZ3

Figure 6.8 Speed sensor schematic diagram

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 2 Volt 10 ms 2) Ch 2: 2 Volt 10 ms

Figure 6.9 Speed pulses waveforms

107

Generator line current sensor

Although hall-effect current sensors had no power losses and do not need

secondary power source, they were not easy to source as well as linear

opto couplers for optical voltage coupling.

As an alternative to the hall-effect current sensors, current sensing

resistors were used. The sensing resistors were connected in series in the

generator lines. According to Ohm’s law

V = I*R

The voltage developed across the sensing resistor was proportional to the

current flowing in the line. This voltage was used to determine the line

current.

Figure 6.11 described how a line current was detected. As mentioned in

the preceding paragraph the line current was measured in terms of

voltage, terminals +BL and –BL are connected across the sensing resistor.

The voltage was picked up with a low cost, low power, true RMS-to-DC

converter AD736 configured in a differential input mode. Refer to figure

6.11 for the configuration characteristics.

108

Line current sensor characteristic curve

01020304050607080

0 1 2 3 4 5 6 7 8 9 10

Line current (A)

Vol

tage

(mV

)

sen-voltage (mV)

Figure 6.10 Line current sensor characteristics curve

The output of the AD736 configuration was fed to a non-inverting

amplifier with a voltage follower to drive an opto-coupler.

When AD736 output voltage increased, the opto-coupler diode current IF

increased proportionally with the increasing output voltage. See figure

6.10. Consequently, the coupler collector current increased and thus set

up voltage drop across resistor VVR 1 6 . When VVR 1 6 increased the transistor

VC decreased. See figure 6.12. VVR 1 6 was then amplified with unity gain

differential amplifier. The amplifier resistor ratio conformed to the

optimization requirement of the amplifier as stated in section 3.4.3.

The amplified potential difference across VR16 also conformed to

potential divider rule and therefore could be expressed as:

VVR 1 6 = (V s – Vc) * VR16 6.4 (VR16 + R6 8)

109

Line current sensor L-pass filter Non inverting amp Unity dif. Amp.

3

26

15

74 U12

C13

C16

COM+4.5

+4.5

-4.5

-4.5

1234 5

678U13

+BL

-BL

10

98

U16C

R61R62

R63

R64

R65

R66

VR14 VR15

3

21

411

U19A5

67

U19B

10

98

U19C

C21

C22

+4.5

-4.5COMCOMU20

COM

R67

R68 R69

R70

VR16+15V

-B

IB

R71 -B

Figure 6.11 Line current sensor schematic diagram

Opto IV curves

0

2

4

6

8

10

12

14

16

0 5 10 15 20

I fwd (mA)

V (V

)

Vc I fwd Ic

Figure 6.12 4N25 current and voltage characteristic curves

Owing to the fact that there were some problems with sourcing the linear

opto-coupler LOC110, an opto-isolator 4N25 was tested for linearity. The

test was done to determine whether 4N25 could be used in place of

LOC110. As indicated in figure 6.12, 4N25 presented a linear response

from 2mA to about 10mA IF. When IF was increased further above 10mA,

the increase in IF did no longer generate a proportional increase in VVR 1 6 .

110

The graph indicated that 4N25 LED saturation started at 10mA. Therefore

when it was used in a signal translating circuit it could not be driven into

saturation because the linear response was critical.

Because hall-effect current sensors could not be sourced, it was decided

that the one at hand be used to determine the rotor current. Figure 6.13

showed that an increase in rotor current caused a proportional increase in

the hall sensor output voltage. As a result, a linear response was obtained.

The response obviated a need for complex mathematics within software.

According to y = mx + c the crucial parameter was m, dy/dx.

This parameter defined a relationship between a measured current and the

sensor output. A third parameter, c, was eliminated within hardware with

an op-amp offset adjust, or by setting analog inputs reference so that it

was equal to the analog signal offset. This way, response curve started

from the graph’s point of origin. Therefore y = mx.

BB-100 current test

00.10.20.30.40.50.60.70.80.9

1

0 1 2 3 4 5 6

Load current (A)

V-o

ut (V

)

Figure 6.13 Hall-effect current sensor characteristic curves

With reference to figure 6.14 generated voltage was read as a fraction of

the actual generated voltage. The read value was then amplified with an

111

instrumentation amplifier to voltage level that represented the generated

voltage accurately. The voltage could therefore be expressed as:

Vg = +Vg *dPT1/(R1+PT1+R2) 6.5

R3

+Vg

- VB

PT1

R1

R2

+

-

U1Vo

Vg

Figure 6.14 Voltage detector schematic diagram

6.3 Switching circuit

Interfacing circuit linked one circuit to the other. For instance, it l inked

the generator stator and rotor circuits to the micro-controller. The

interfacing circuit’s main task was to translate analog signals and digital

signals between the micro-controller and controlled parts of the circuit.

An interface circuit performed a number of tasks in the controller

circuitry, and those were:

Buffering data temporarily

Isolating micro-controller from the noisy parts of the controller

circuitry to prevent damage to the micro-controller.

Multiplexing input data

Converting analog inputs to digital inputs

Timing adjustment to ensure synchronized data exchange.

112

Dz R1

R2

R3

R4

Q1

S2

Vbat limits detector

Q2

Figure 6.15 Load disconnect switch

Figure 6.15 showed two switching resistor networks, R1 and R2 for

switching Q1, and R3 and R4 for switching Q2 . When Q2 was on, R1 set up

Q1 switching voltage and when that voltage was greater than Q1 threshold

voltage, Q1 started conducting. The voltage was clamped within limits

with a zener diode DZ. When DZ clamped VR 1 at 12V, R2 served as a DZ

current limiting resistor. To ensure that Q2 turned on and off when an OR

gate output was High and Low respectively, a second potential divider

resistor network, R3 and R4 , was used. The network was such that VR 4 was

equal to Q2 VB E. It was approximately equal 0.85V when the OR gate

output was High, 5V, and Q2 VC E was nearly zero volts. See figure 6.16.

On the other hand, when the gate output was Low, VR 4 turned Q2 off. That

was because the VR 4 was less than the required VB E to keep Q2 on. At that

moment, Q2 VC E was approximately equal to the supply voltage. See figure

6.15.

113

2N3904 Switching voltage test

0.01

0.1

1

10

100

0.01 0.11 0.21 0.31 0.41 0.51 0.61 0.71 0.81 0.91

V BE (V)

V C

E (V

)

Figure 6.16 2N3904 VB E vs VC E .

Isolation techniques

A micro-controller was optically and magnetically linked to the generator

stator and rotor circuit. That was done to ensure safe and reliable

operation of the micro-controller. The two common isolation techniques

were those where relays and opto-isolator were utilized. Relays and opto-

isolators present good isolation between low voltage circuits and high

voltage circuits. In addition, sensitive control circuits were protected

from inadvertent voltage and current spikes generated in the high voltage

circuits.

An electromagnetic switch, relay, was still preferred for isolation between

the generator circuits and the micro-controller. Because relay coils

generated back emf when they were switched off, a freewheeling diode

was used to protect the relay driver. The relay also had frequency

limitations that determined its life expectancy. According to (Loveday,

1992, p 218) electromagnetic switches were suitable for frequencies

below 20Hz.

114

Dv-B

oard

Vcc +V

R1

U1

0V

Load

D1

Q1

Figure 6.17 Relay drive circuit

6.4 Micro-controller

Micro-controllers have ability to store and run unique programs in a

similar manner to that of the computer. For instance, they could be

programmed to perform functions based on the predetermined situation.

Therefore, they defined the intelligence of the control systems.

The programmable nature of the micro-controller enabled it to perform

complex tasks. To get it to change tasks one had to arrange for a change

in software. The micro-controller continuously reads inputs from the

sensors, computes errors and then outputs control commands to drive

external circuitries, one of which was the DC/DC converter.

Micro controller’s ability to control external devices made it very useful

where an automatic control of electronic circuit was required. The

controller eliminated the need for analog to digital converters ADC and

digital to analog converters DAC as it had them built in (Carr, 1991,

p420). One such controller was PIC16F877. It was decided that a

PIC16F877 development system board be used to test software command

functions.

115

6.5 Operation control modes

Operation control modes were selected by setting digital output port bits

either “High” or “Low” depending on an error generated during execution

of the software command functions. The next paragraph describes the

output port bit status.

Digital output control port

MBS LSB S8 S7 S6 S5 S4 S3 S2 S1 Converter DC load disconnect Iex current polarity changer Over voltage load Speed counter enable Analog input select bits

Bit status definition

Bit State Definition

S1: 1 Turns OFF DC / DC converter

0 Turns ON DC /DC converter

S2: 0 Disconnects DC loads from the battery bank

1 Connects DC loads to the battery bank

S3: 0 Selects positive exciting current polarity

1 Selects negative exciting current polarity

S4: 0 Disconnects fault load from the generator and connects

charge control circuit.

116

1 Connects fault load to the generator and disconnects fault

load

S5: 0 Enables generator speed counter

1 Disables generator speed counter

Bit S6, S7 and S8 were used to select an analog quantity to be read when

a single mode operation was selected. The bits status was changed by

writing a hexadecimal number to the LED port, therefore the required

control state was set, for example, passing 2 to the LED port as

Write_LEDS(0X02);

resets the speed counter for a start of a new count. It was mentioned that a

digital counter was used to convert the speed pulses in to digital data that

the microcontroller could understand. The number of pulses read in one-

second represented the generator speed in hertz. The following expression

showes how the speed count was read:

Sp = Read_Switches();

where Sp was the speed variable and Read_Switches() was a function that

read the digital port state. This was because the digital counter was

interfaced to the microcontroller via the 8-bit switches-port.

Select variable bits are described below.

S8 S7 S6 Q Schematic-symbol Selected analog 0 0 0 Vg A DC generated voltage 0 0 1 Irt B Rotor Current 0 1 0 Ireg C Regulator current (charge current) 0 1 1 Vbat D Battery bank voltage 1 0 0 Vrt E Rotor voltage 1 0 1 Ir F Red line current

117

1 1 0 Iy G Yellow line current 1 1 1 Ib H Blue line current

6.5.1 Analog and digital input signals execution

Analog input signals were multiplexed and read in sequence with channel

zero. An average of 50 samples of each analog input signal was taken to

assume an accurate reading. The following lines describe how the analog

input was read:

read_analog() byte i=0; set_adc_channel(0); //Select channel 0 adc_value = 0; //Zero variable for(i=0;i<50;i++) //Setup 50 times loop for 50 ADC samples adc_value = adc_value + Read_ADC_10(); //Add all readings //to variable delay_us(12); //time delay between samples taken //See table 11.1 note 1 in //datasheet of the PIC16F877 p117 // (PIC16F877 Development System) adc_value = adc_value/50; // Assigns adc_average value to // adc_value. Analog signals were continuously read during each mode of operation

although the address bit status was unique for each mode. The bit status

was such that the mode operation conditions were left unchanged when

both analog and digital inputs were read. The LED port bits were bit

masked to make inputs reading simple without changing mode

requirements. See Bit-masking tables in appendix A.

The generator speed was read as a digital input signal to switches port of

the development board. In the same way, the speed was read during all the

118

modes without changing the modes operation conditions. The following

lines show how the speed was read after the speed counter enable pulse

was sent.

Write_LEDS(0x02); // enables generator speed counter Write_LEDS(0x00); delay_ms(1000); // one second delay Sp=Read_Switches(); // Reads speed in hertz Write_LEDS(0x11); // selects Vg & disconnects DC load

6.5.2 Operation mode selection

The operation modes were selected because of continuous monitoring of

generated voltage Vg and generation of appropriate error signals to suit

the selected mode. Control bits in a lower nibble of the LED port were

used to retain the state of the selected mode during execution. The higher

nibble was for inputs selection.

Operation conditions as per mode:

Permanent magnet mode

DC/DC converter was kept off

Exciting current polarity changer was kept off

The balanced star fault load was kept disconnected

Disconnect voltage sensitive DC loads from the battery bank when

its voltage was out of range, 24V – 28V.

Boost mode

DC/DC converter was kept on

Exciting current polarity changer was kept on

The balanced star fault load was kept disconnected

119

Disconnect voltage sensitive DC loads from the battery bank when

its voltage was out of range, 24V – 28V.

Buck mode

DC/DC converter was kept on

Exciting current polarity changer was kept off

The balanced star fault load was kept disconnected

Disconnect voltage sensitive DC loads from the battery bank when

its voltage was out of range, 24V – 28V.

Fault mode

The fault mode was mainly focused on the system protection: line

currents, over speed, exciting current and charge current protection. It

ensured that the line currents were balanced as an imbalance caused

uneven loading of the generator. In this mode, permanent mode settings

are adopted as they presented reduced power losses as opposed to other

modes.

The following lines of code evaluate line currents balance by first

working out what each current should be, and creating a ±5% tolerance

window. That was because the possible line currents imbalance error was

found to be 3.3%. See performance data results in appendix C.

Il=(Ireg*(1000/955))*(707/1000); // Determine line current Ilmx=Il+(Il*(5/100)); // I-line max = I-line + 5% Ilmn=Il-(Il*(5/100)); // I-line min = I-line - 5% sel=0; // initialize current select variable This part of the code evaluated the line currents and determined whether

they were within the ±5% tolerance range. If one line current was not

within range, the line current variable will be set High or otherwise Low

as indicated in the following case definitions.

120

for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilmn<Ir)&&(Ir<Ilmx))) // Is Ir not within tolerance range? Red=1; // If yes note fault in red line else // like wise with other lines REd=0;break; // break indicates end of case case 1: if(!((Ilmn<Iy)&&(Iy<Ilmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilmn<Ib)&&(Ib<Ilmx))) Blue=1; else Blue=0;break; At the end of line currents inspection, the line current’s variables, Red,

Yellow and Blue states were checked whether they were equal. If they

were not equal, the balanced star fault load would be connected to the

generator with the charge circuit disconnected. Then the new line current

average would be determined with the read line current values as shown in

the following lines of code. In the same way as in the preceding paragraph

the ±5% tolerance window was created and the currents were inspected for

the second time. Then the operation would be locked in the fault mode if

the currents were not balanced.

while(!((Red==Yellow)&&(Yellow==Blue))) // Wait untill lines are not balanced // Generator is connected to balanced star sel=0; // network so all the line currents are expected Fault_ana(); // to have close relationship, 5% tolerance. Ilav=(Ir+Iy+Ib)/3; // Determine average line current Ilavmx=Ilav(1+(5/100)); // Determine max average line current Ilavmn=Ilav(1-(5/100)); // Determine min average line current sel=0; The next lines of code continuously monitored the line currents for an

imbalance.

121

for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilavmn<Ir)&&(Ir<Ilavmx))) // if Ir is not within range //and if yes,note the fault. Like wise Red=1; else // with other lines. Red=0;break; case 1: if(!((Ilavmn<Iy)&&(Iy<Ilavmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilavmn<Ib)&&(Ib<Ilavmx)) Blue=1; else Blue=0;break; sel=0; // resets select variable

122

Chapter 7

Conclusion

7.1 Conclusion

The controlled hybrid field generator test results as well as the control

test boards results have proved that introducing a controller for the

generator will optimize the generator performance. The literature survey

also indicated that utilizing low power and temperature compensated

components exhibits reduced on board power losses, and such components

ensure reduced drift in signal translating circuit response hence an

improved linearity and performance.

With an over voltage load included in the input stage of the charging

circuit a wider rage of speed, 149rpm up to 815rpm (10Hz to 54Hz), is

covered whilst still maintaining charging voltage at 28V.

Because an over voltage load was not within the regulation window, its

voltage was not regulated. It reduced stress on charging voltage regulation

components. This was because at speeds above 598rpm, the input charging

voltage started to increase to levels higher than 32.41VDC , resulting in an

excess voltage being diverted to the load. A limiting factor in this case

was the voltage ratings of input stage components like: bridge rectifier

diodes maximum blocking voltage and smoothing capacitor’s rated

voltages, which were 200VDC . That is, excitation could resume at any

desired voltage level below the components rated voltages.

In the case of the hybrid field generator, 24VAC l ine voltage, at 149rpm,

was maintained to keep the charging voltage fixed at 28VDC before

123

excitation. The excitation in this case started at speeds below and above

149rpm, and at voltages above and below 32.41VDC , where 32.41VDC is a

corresponding average voltage when line voltage is 24VAC .

Based on a DC/DC converter test data in appendix C, it is evident that a

1:1 transformer would be suitable for the converter because the generator

rotor needed 22V for maximum excitation whilst the battery bank voltage

is 24V. This would also halve the transformer’s primary copper losses as

opposed to the 1:2 transformer. In addition to reduced losses, the 1:1

transformer would improve the battery bank durability because of reduced

discharge rate.

To ensure an efficient power conversion, switching MOSFETs had to be

matched. Matching the MOSFETs reduced chances of conversion failure

and ringing effects in the converter circuit.

A zener diode and a blocking diode in the converter output stage reduced

effects of back emf mainly when the converter was shutdown because the

effects escalate during the shutdown period. Those were directly

proportional to the generator speed and the load current.

A relationship between the fixed 24V at 600rpm, the load current and the

exciting current is shown in figure 7.1. The 24V was maintained over the

load current range (0A to 10A). At every instant of time when load

current increased, the exciting current decreased resulting in a stable

voltage and vice versa. A decrease in the exciting current due to the

increase in the load current resulted in the reduced armature power losses.

In this way, the excitation control efficiency was improved. That also

implied that the controlled generator loading could be used to stabilize

the generated voltage prior to the armature excitation process resumption.

124

The generator loading process would stop when the full load current is

reached. Then the excitation process would start.

Full load current can therefore be represented as follows:

Ifu l l load = 10A = Ireg + Icon t ro l led load i n g 7.1

Excitation IV curves at 24V V-line 600rpm

01020304050607080

0 2 4 6 8 10 12

DC Load current (A)

Vol

tage

s (V

) AC

&

DC

-6-4-2024681012

Cur

rent

s (A

)

Vline (V)Vdc (V)Vrotor (V)Iline (A)Idc (A)Iexc (A)

Figure 7.1 On load excitation curves at 24V line voltage and at 600rpm

It was found that the efficiency of the system at full load could be

improved by setting the load current to be directly proportional to the

generator speed. For example, if the current at 149rpm could be restricted

to 2A the generator stator copper losses would be reduced by 92.7%. The

controller on board losses would be reduced to 25% of the maximum

losses.

The generated voltage level appeared to have dropped by 7.7% after a

series of repeated generator tests. See figure 7.2. This was because the

magnetism of the permanent magnets embedded in the rotor was reduced.

This was clearly indicated by the generated voltage formula:

125

E = 4.44NΦ fK

where N, f and K are constants. Therefore, a drop in the voltage could be

due to a drop in exciting flux density. The demagnetizing effect could be

reduced by replacing the embedded magnets with the magnets that can

retain their magnetism after the buck and boost modes of operation.

O/C votage characteristics over a range of tests (750rpm)

70

80

90

100

110

120

0 2 4 6 8

Test NO:

Vol

tage

(AC

& D

C) i

n V

olts

V-DC V-Line

Figure 7.2 Open circuit voltage characteristics at 750rpm over a range of

voltage tests

Using a microcontroller also plays an important role in that it

continuously compares parameters like; generated voltage and line

current, with their corresponding references, which are set within

software. It also makes modifications of the controller parameter

references easier than when they are to be done in hardware.

Analog inputs to the micro-controller may be read independently, each

input assigned a channel, or be multiplexed. In the latter case, one

channel is used to read the inputs and only one voltage reference is set for

the inputs. This may be a problem if the inputs response graphs do not all

cross the y-axis at the same point. See figure 2.1 and figure 6.10.

126

However, if a problem of this nature arises the former case may be used

although the reference voltage is required for each channel.

Because the generator characteristics were studied and the controller

specifications could be tailored successfully, a controller to suit the

optimized performance of the hybrid field generator may be designed.

7.2 Recommendations

The generator performance test and the control signal test-board test

results analysis lead to the following recommendations for the controller

hardware design purpose:

• Use of temperature compensated components like hall-effect current

sensors and linear opto-coupler LOC110, and Low temperature

coefficient resistors like metal film type resistors should be used in

the hardware design of the controller. These types of components

ensure very low drift in linearity due to temperature rise.

• Signal translating circuitry has to be powered from a dual power

supply. This guarantees easy control over offset voltages that are

associated with operational amplifiers.

• The generator speed sensing needs to be performed with the disk

method because this method presents a more accurate speed

detection. It can also read a fraction of a revolution e.g 149.3rpm

whereas the electronic one presents ±10% error at cut-in speed

(149rpm) and ±2.5% error at cut-out speed (598rpm).

• Power dissipation in the generator leads to temperature rise in the

generator, therefore the temperature will also have to be taken into

127

account during the design and development of the controller. This

component can lead to the failure of the generator if not mitigated.

• The DC/DC converter transformer needs to be designed and

developed to suit the generator excitation.

7.3 Future work

Present day technology focuses on human machine interface. Therefore, a

further improvement in the development of hybrid field generator control

systems also has to be channeled in the same direction. Figure 7.3 shows a

supervisory control system that would enable the system operator to view

the system performance. It would also make system fault diagnosis and

maintenance easier.

Referring to chapter 7, the causes of line current imbalance could be

located by comparing the line currents before and after connecting the

fault load. If the currents imbalance could be maintained even when the

load is connected, then the problem would be with the generator or

otherwise the charging unit. The fault location could be carried out to the

extent where it could be known whether the problem is with the red,

yellow or blue line by just monitoring the line currents.

128

Generatorexcitation and

power controller

24VBattery bank DC loads

Hybrid fieldgenerator

Operator station

ControlledloadingSynchronizing unitGrid

terminal

Figure 7.3 Hybrid field control station network

129

List of references

ABB. (1998). Effects of AC Drives on Motor Insulation USA: ABB

Industrial Systems Inc

Ahmed, A. (1999). Power Electronics for Technology. New Jersey:

Prentice-Hall-Inc.

Ang, S. S. (1995). Power Switching Converters. New York: Marcel

Dekker, Inc

Beak, J ., Budding, P., & Scaini V. (2001). Reusing and Rerating Older

Rectifiers with New DC/DC Choppers. IEEE Transactions on industry

applications, VOL. 37, NO. 4, pp. 1160 – 1166.

Bell, F. W. (2001).Current Sensors Catalogue USA: Sypris

Carr, J . T. (1991). Microcomputer interfacing New Jersey: Prentice hall

Cripps, M. (1989. Computer interfacing. Great Britain: Routledge

Bergquist, C. (2001). Guide to PICMICRO Microcontrollers. USA:

Prompot.

Billings, K. H. (1999). Switchmode Power Supply Handbook.(2n d ed)

Mexico: McGraw-Hill Inc.

Byers T. J . (1991). Printed Circuit Board Design with Microcomputers.

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130

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135

Appendix A

Flow charts and blocks code

Read Analog Value Functionread_analog()

Start

Initialize Variable

set_adc_channel(x)

initialize acd variable

For int i<50

READ_ADC( )

Assign sum of read values to ADC variable

Assign read values average to ADC variable

End

<50

=>50

136

Reverse / Forward / Permanent Mode Functions Flow chartRM_read_analog() / FM_read_analog() / PM_read_analog()

Start

Initialize selection variableinteger sel

For sel. < 8

If !(24<Vbat<28)

Read speed (Sp) in Hz &Convert Sp to RPM

Read speed (Sp) in Hz &Convert to RPM

Select generatedvoltage with load con.

Select generatedvoltage with load Disc.

read_analog()

Assign read analogvalue to Vg

If !(24<Vbat<28)

Select Irt withload Disc.

Select Irt withload con.

read_analog()

Assign read analogvalue to Irt

If !(24<Vbat<28)

Select Ib withload Disc.

Select Ib withload con.

read_analog()

Assign read analogvalue to Ib

End

case 0

case 1

case 7

switch (sel)

All cases block sequences are similar except that each case selectsa variable to be read. The case sequence of variables for "Fault_ana()" is different from those of other functions, refer to the fault software.

Fault variables read function

Not

Within

Not

Within

Not

Within

<8

=>8

(Fault_ana())

137

Fault Analysis and Operation Function

Fault_ana()

Start

While !(149<Sp < 598)

Fault_ana()

Determine I-line from Ireg(load Current)

Check whether I-lines are within 5% tolerance range

While not within range connectbalanced star network

Determine I-lines averagefrom sum of the line currents

Fault_ana()

Check whether I-lines are within 5% tolerance rangeand display fault message as per current

Not

End

This function analyses the currents and consequently display fault location. It first checkswhether line currents are within 5% tolerance range. If not, it connects a balanced starnetwork and checks whether the currents are within 5% tolerance range. If yes, the fault ison the controllers side and if not it is on the generator's side for that particular line.

Within

NotWithin

138

Main Function

Start

InitializePIC16F877

Setup port A

Setup ADC

Write To LEDS

Select operationmode

While powerON

End

ON

OFF

Select Operation Mode

Start

Permanent mode

If 149<speed <598rpm

If 28<G-voltage <35V

Display PM mode

Permanent mode

If G-voltage< 28V

Display FM mode

Forward mode

Display RM mode

Reverse mode

Fault operation

End

>35V

Yes

NoYes

No

<28V

Opr_Msel()void main (void)

139

/********************************************************************* Compiler: CCS C compliler www.ccsinfo.com Editor: Microchip MPLAB www.microchip.com Filename: HFG_TstC.c Description: The Hybrid Field Generator (HFG) control function simulation blocks

were tested with PIC16F877 development system board. The test was focused on the sequential handling of the functions as per control event occurrence.

Test Device: PIC16F877 development system (Phillips. Adlam. 2001) Author: C.T. 'Moleli Date (2003) *********************************************************************/ #include "D877_INI.C" #include "LCD.C" #include "KEYPAD.C" int sel; unsigned int Sp=374,Vg=35,Irt=3,Ireg,Vbat=26,Vrt,Ir=4,Iy=4,Ib=4,Il, Ilav,Ilavmx,Ilavmn,Ilmx,Ilmn,Red,Yellow,Blue,adc_value=0; /********************************************************************* The function reads and returns a selected analog input value *********************************************************************/ read_analog() byte i=0; set_adc_channel(0); //Select channel 0 adc_value = 0; //Zero variable for(i=0;i<50;i++) //Setup 50 times loop for 50 ADC samples adc_value = adc_value + Read_ADC_10(); //Add all readings //to variable delay_us(12); //time delay between samples taken //See table 11.1 note 1 in //datasheet of the PIC16F877 p117 adc_value = adc_value/50;

140

/********************************************************************* The function measures Speed during calibration *********************************************************************/ Get_Speed() Write_LEDS(0x03); //The Write_LEDS() commands set a pulse Write_LEDS(0x13); //to triger speed counter Sp=Read_Switches(); //Read speed in hertz /******************************************************************** The function allows an operator to calibrates the variables during run-time. The calibration block immediately switches to the permanent magnet except for the rotor current and rotor voltage. ********************************************************************/ Calibrate() while(Read_Keypad()=='*') // Hold cursor in calibration lcd_clrscr(); //Clear LCD module display delay_ms(100); while(Read_Keypad()!='*') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Calibration ON"); //Write message to LCD lcd_gotoxy(0,1); Lcd_putc("Press * to Exit"); switch(Read_Keypad()) case '0':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') //Wait utill # is pressed lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Generator Speed"); //Write message to LCD Get_Speed(); lcd_gotoxy(0,1); printf(lcd_putc,"Speed = %2u",Sp ); lcd_gotoxy(11,1); lcd_putc("Hz"); lcd_clrscr(); //Clear LCD module display break; case '1':lcd_clrscr(); //Clear LCD module display

141

while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Generated Volt"); //Write message to LCD Write_LEDS(0x13); //Selects Vg read_analog(); Vg=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"Voltage = %2u",Vg); lcd_gotoxy(13,1); lcd_putc("V"); lcd_clrscr(); //Clear LCD module display break; case '2':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Exciting Current"); //Write message to LCD Write_LEDS(0x33); read_analog(); Irt=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I ex = %2u",Irt); lcd_gotoxy(11,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '3':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); lcd_putc("Charging Current"); //Write message to LCD Write_LEDS(0x53); read_analog(); Ireg=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I Charge = %2u",Ireg); lcd_gotoxy(14,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '4':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("B. Bank Voltage"); //Write message to LCD

142

Write_LEDS(0x73); read_analog(); Vbat=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"V B Bank = %2u",Vbat); lcd_gotoxy(14,1); lcd_putc("V"); lcd_clrscr(); //Clear LCD module display break; case '5':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Rotor Voltage"); //Write message to LCD Write_LEDS(0x93); read_analog(); Vrt=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"V Rotor = %2u",Vrt); lcd_gotoxy(13,1); lcd_putc("V"); lcd_clrscr(); //Clear LCD module display break; case '6':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Red Line Current"); //Write message to LCD Write_LEDS(0xb3); read_analog(); Ir=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I Red = %2u",Ir); lcd_gotoxy(11,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '7':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Yellow L Current"); //Write message to LCD Write_LEDS(0xd3); read_analog(); Iy=adc_value; lcd_gotoxy(0,1);

143

printf(lcd_putc,"I Yellow = %2u",Iy); lcd_gotoxy(14,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '8':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Blue L Current"); //Write message to LCD Write_LEDS(0xf3); read_analog(); Ib=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I Blue = %2u",Ib); lcd_gotoxy(12,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; lcd_clrscr(); //Clear LCD module display /********************************************************************* The function reads analog input values during reversed rotor excitation, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). *********************************************************************/ RM_read_analog() lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc(" Reversed "); //Write message to LCD lcd_gotoxy(0,1); lcd_putc("Excitation"); Calibrate(); sel=0; for(sel=0;sel<8;sel++) switch(sel)

144

case 0: if(!((24<Vbat)&&(Vbat<28))) // check whether battery bank voltage is // within limts Write_LEDS(0x00); Write_LEDS(0x10); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x10); // selects Vg & disconnects DC load else Write_LEDS(0x02); Write_LEDS(0x12); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x12); // selects Vg (Generated DC voltage) // with DC load connebted delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vg=adc_value; // assigns adc_value to Vg break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x30); // selects Irt & disconnects DC load else Write_LEDS(0x32); // selects Irt (Rotor/excitation current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Irt=adc_value; // assigns adc_value to Irt break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x50); // selects Ireg & disconnects DC load else Write_LEDS(0x52); // selects Ireg (Voltage regulator current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ireg=adc_value; // assigns adc_value to Ireg break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x70); // selects Vbat & disconnects DC load else Write_LEDS(0x72); // selects Vbat (Battery bank voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vbat=adc_value; // assigns adc_value to Vbat break;

145

case 4: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x90); // selects Vrt & disconnects DC load else Write_LEDS(0x92); // selects Vrt (Rotor Voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vrt=adc_value; // assigns adc_value to Vrt break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xb0); // selects Ir & disconnects DC load else Write_LEDS(0xb2); // selects Ir (Red line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ir=adc_value; // assigns adc_value to Ir break; case 6: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd0); // selects Iy & disconnects DC load else Write_LEDS(0xd2); // selects Iy (Yellow line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Iy=adc_value; // assigns adc_value to Iy break; case 7: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf0); // selects Ib & disconnects DC load else Write_LEDS(0xf2); // selects Ib (Blue line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ib=adc_value; // assigns adc_value to Ib break; /********************************************************************* The function reads analog input values during forward rotor excitation, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). *********************************************************************/ FM_read_analog() lcd_gotoxy(0,0); //Set LCD cursor to position

146

lcd_putc("Forward "); //Write message to LCD lcd_gotoxy(0,1); lcd_putc("Excitation"); Calibrate(); sel=0; for(sel=0;sel<8;sel++) switch(sel) case 0: if(!((24<Vbat)&&(Vbat<28))) // check whether battery bank voltage is // within limts Write_LEDS(0x04); Write_LEDS(0x14); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x14); // selects Vg & disconnects DC load else Write_LEDS(0x06); Write_LEDS(0x16); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x16); // selects Vg (Generated DC voltage) // with DC load connebted delay_ms(3000); // delay before taking a reading read_analog(); // reads selected analog input Vg=adc_value; // assigns adc_value to Vg break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x34); // selects Irt & disconnects DC load else Write_LEDS(0x36); // selects Irt (Rotor/excitation current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Irt=adc_value; // assigns adc_value to Irt break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x54); // selects Ireg & disconnects DC load else Write_LEDS(0x56); // selects Ireg (Voltage regulator current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input

147

Ireg=adc_value; // assigns adc_value to Ireg break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x74); // selects Vbat & disconnects DC load else Write_LEDS(0x76); // selects Vbat (Battery bank voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vbat=adc_value; // assigns adc_value to Vbat break; case 4: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x94); // selects Vrt & disconnects DC load else Write_LEDS(0x96); // selects Vrt (Rotor Voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vrt=adc_value; // assigns adc_value to Vrt break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xb4); // selects Ir & disconnects DC load else Write_LEDS(0xb6); // selects Ir (Red line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ir=adc_value; // assigns adc_value to Ir break; case 6: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd4); // selects Iy & disconnects DC load else Write_LEDS(0xd6); // selects Iy (Yellow line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Iy=adc_value; // assigns adc_value to Iy break; case 7: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf4); // selects Ib & disconnects DC load else Write_LEDS(0xf6); // selects Ib (Blue line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ib=adc_value; // assigns adc_value to Ib break;

148

/******************************************************************** The function reads analog input values during no rotor excitation, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). ********************************************************************/ PM_read_analog() lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("P Magnet "); //Write message to LCD lcd_gotoxy(0,1); lcd_putc("Excitation"); Calibrate(); sel=0; for(sel=0;sel<8;sel++) switch(sel) case 0: if(!((24<Vbat)&&(Vbat<28))) // check whether battery bank voltage is within limts Write_LEDS(0x01); // enables generator speed counter Write_LEDS(0x11); delay_ms(1000); // one second delay Sp=Read_Switches(); // converts generator speed in Hz to RPM Write_LEDS(0x11); // selects Vg & disconnects DC load else Write_LEDS(0x03); // enables generator speed counter Write_LEDS(0x13); delay_ms(1000); Sp=Read_Switches(); //converts generator speed in HZ to RPM Write_LEDS(0x13); // selects Vg (Generated DC voltage)

// with DC load connebted delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vg=adc_value; // assigns adc_value to Vg break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x31); // selects Irt & disconnects DC load

149

else Write_LEDS(0x33); // selects Irt (Rotor/excitation current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Irt=adc_value; // assigns adc_value to Irt break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x51); // selects Ireg & disconnects DC load else Write_LEDS(0x53); // selects Ireg (Voltage regulator current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ireg=adc_value; // assigns adc_value to Ireg break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x71); // selects Vbat & disconnects DC load else Write_LEDS(0x73); // selects Vbat (Battery bank voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vbat=adc_value; // assigns adc_value to Vbat break; case 4: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x91); // selects Vrt & disconnects DC load else Write_LEDS(0x93); // selects Vrt (Rotor Voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vrt=adc_value; // assigns adc_value to Vrt break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xb1); // selects Ir & disconnects DC load else Write_LEDS(0xb3); // selects Ir (Red line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ir=adc_value; // assigns adc_value to Ir break; case 6: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd1); // selects Iy & disconnects DC load else Write_LEDS(0xd3); // selects Iy (Yellow line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Iy=adc_value; // assigns adc_value to Iy break;

150

case 7: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf1); // selects Ib & disconnects DC load else Write_LEDS(0xf3); // selects Ib (Blue line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ib=adc_value; // assigns adc_value to Ib break; /******************************************************************** The function reads analog input values during the fault period, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). ********************************************************************/ Fault_ana() Write_LEDS(0x79); // selects battery bank voltage & disconect the load read_analog(); Vbat=adc_value; for(sel=0;sel<6;sel++) switch(sel) case 0: if(!((24<Vbat)&&(Vbat<28))) // Checks whether Vbat is within

// limilts Write_LEDS(0xb9); // selects Red line current with load

// disconnected read_analog(); Ir=adc_value; else Write_LEDS(0xbb); // selects Red line current with load

// connected read_analog(); Ir=adc_value; break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd9); // selects Yellow line current with

151

// load disconnected read_analog(); Iy=adc_value; else Write_LEDS(0xdb); // selects Yellow line current with //load connected read_analog(); Iy=adc_value; break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf9); // selects Blue line current with load // disconnected read_analog(); Ib=adc_value; else Write_LEDS(0xfb); // selects Blue line current with load // connected read_analog(); Ib=adc_value; break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x09); // selects Speed with load

//disconnected Write_LEDS(0x19); // selects Generated voltage delay_us(12); Sp=Read_Switches(); // Converts Hz to RPM read_analog(); // read Generated voltage Vg=adc_value; else Write_LEDS(0x0b); // selects Speed with load connected Write_LEDS(0x1b); // selects Generated voltage delay_ms(1000); Sp=Read_Switches(); // converts Hz to RPM read_analog(); // read Generated voltage Vg=adc_value; break; case 4: if(!((24<Vbat)&&(Vbat<28)))

152

Write_LEDS(0x79); // selects Vbat with load // disconnected read_analog(); Vbat=adc_value; else Write_LEDS(0x7b); // selects Vbat with load connected read_analog(); Vbat=adc_value; break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x39); // selects Irt with load disconnected read_analog(); Vbat=adc_value; else Write_LEDS(0x3b); // selects Irt with load connected read_analog(); Vbat=adc_value; break; /******************************************************************* The function monitors an exciting current. It connects fault load and shutdown the DC/DC converter when the exciting current exceeds 5A. *******************************************************************/ Rotor_Current() lcd_clrscr(); //Clear LCD module display while(Irt>5) lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Over Excitation"); //Write message to LCD Calibrate(); delay_us(12); Fault_ana();

153

/******************************************************************* The function minitors the line currents. It connects fault load and shutdown the DC/DC converter when the currents are not +/- 5% equal to each other. *******************************************************************/ Fault_Opr_Current() lcd_clrscr(); //Clear LCD module display while(!((Red==Yellow)&&(Yellow==Blue))) // Wait utill lines are balanced // Generator is cconnected to balanced star sel=0; // netwok so all the line currents are expected Fault_ana(); // to have close relationship, 5% tolerance. Ilav=(Ir+Iy+Ib)/3; // Determine average line current Ilavmx=Ilav+(Ilav*(5/100)); // Determine max average line current Ilavmn=Ilav-(Ilav*(5/100)); // Determine min average line current sel=0; for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilavmn<Ir)&&(Ir<Ilavmx))) // if Ir is not within range

// and Red=1; // if yes,note the fault. like wise else // with other lines. Red=0;break; case 1: if(!((Ilavmn<Iy)&&(Iy<Ilavmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilavmn<Ib)&&(Ib<Ilavmx))) Blue=1; else Blue=0;break; lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("L Current Imbal"); //Write message to LCD delay_ms(1000); sel=0; // resets select variable

154

/******************************************************************** The function monitors the line currents for the line imbalance ********************************************************************/ Test_Opr_Current() Il=(Ireg*(1000/955))*(707/1000); // Determine line current Ilmx=Il+(Il*(5/100)); // I-line max = I-line + 5% Ilmn=Il-(Il*(5/100)); // I-line min = I-line - 5% sel=0; for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilmn<Ir)&&(Ir<Ilmx))) // Is Ir not within tolerance range? Red=1; // If yes note fault in red line else // like wise with other lines Red=0;break; case 1: if(!((Ilmn<Iy)&&(Iy<Ilmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilmn<Ib)&&(Ib<Ilmx))) Blue=1; else Blue=0;break; Rotor_Current(); Fault_Opr_Current(); /******************************************************************** The function minitors the HFG speed when HFG speed is not within range. It connects fault load and shutdown the DC/DC converter. *********************************************************************/ Fault_Opr_Speed() while(!((10<Sp)&&(Sp<40))) // Checks whether the speed is within range and // wait for as long as it is not. lcd_gotoxy(0,0); //Set LCD cursor to position

155

lcd_putc("Speed Range"); //Write message to LCD Calibrate(); delay_ms(1000); sel=3; Fault_ana(); Test_Opr_Current(); /******************************************************************** The function monitors HFG speed and the generated voltage. It then selects an operation mode accordingly. ********************************************************************/ Opr_Msel() Test_Opr_Current(); if((10<Sp)&&(Sp<40)) // Checks whether speed is within limits if((30<Vg)&&(Vg<35)) PM_read_analog(); else if(Vg<30) FM_read_analog(); else if(Vg>35) RM_read_analog(); Calibrate(); else Fault_Opr_Speed(); Calibrate(); /******************************************************************** The function initializes the PIC16F877 development system and the lcd display module for HMI. ********************************************************************/ void main (void) D877_Init(); //Initialise PIC16F877 development system

156

lcd_Init(); //Initialise LCD display setup_port_a(RA0_RA1_ANALOG_RA3_REF); //Enable channels 0 and 1 - RA3 is //used as reference input setup_adc(ADC_CLOCK_INTERNAL); //Use internal RC clock for ADC //Conversion time is minimum 72uS PM_read_analog(); //Update variables while(1) Calibrate(); Opr_Msel();

157

Operation modes bit masking tables S3 S2 S1 Ct E Dmp Pol L D Con Permanent Forward Reverse Variable MSB LSB mode mode mode 7 6 5 4 3 2 1 0 L L D L L D L L D Speed 0 0 0 0 0 0 1 1 0X03 0X01 0X06 0X04 0X02 0X00 Vg 0 0 0 1 0 0 1 1 0X13 0X11 0X16 0X14 0X12 0X10 Irt 0 0 1 1 0 0 1 1 0X33 0X31 0X36 0X34 0X32 0X30 Ireg 0 1 0 1 0 0 1 1 0X53 0X51 0X56 0X54 0X52 0X50 Vbat 0 1 1 1 0 0 1 1 0X73 0X71 0X76 0X74 0X72 0X70 Vrt 1 0 0 1 0 0 1 1 0X93 0X91 0X96 0X94 0X92 0X90 Ir 1 0 1 1 0 0 1 1 0XB3 0XB1 0XB6 0XB4 0XB2 0XB0 Iy 1 1 0 1 0 0 1 1 0XD3 0XD1 0XD6 0XD4 0XD2 0XD0 Ib 1 1 1 1 0 0 1 1 0XF3 0XF1 0XF6 0XF4 0XF2 0XF0 Fault mode bit masking table S3 S2 S1 Ct E Dmp Pol L D Con Fault Variable MSB LSB mode 7 6 5 4 3 2 1 0 L L D Speed 0 0 0 0 1 0 1 1 0X0B 0X09 Vg 0 0 0 1 1 0 1 1 0X1B 0X19 Irt 0 0 1 1 1 0 1 1 0X3B 0X39 Ireg 0 1 0 1 1 0 1 1 0X5B 0X59 Vbat 0 1 1 1 1 0 1 1 0X7B 0X79 Vrt 1 0 0 1 1 0 1 1 0X9B 0X99 Ir 1 0 1 1 1 0 1 1 0XBB 0XB9 Iy 1 1 0 1 1 0 1 1 0XDB 0XD9 Ib 1 1 1 1 1 0 1 1 0XFB 0XF9

158

Appendix B

Hybrid field generator performance characteristics tests

results

Generator test results

Generator Characteristics At Mid-Range Speed ( 374 rpm ) Excitation V_ Load I_load V_ Line I_ Line Torque Temp Power in DC Power ( A ) DC DC AC AC Nm Dgr. Cel. W W

0 57.1 0 42.6 0 0.8 21 31.332 00 52.2 1 39.8 0.74 2.4 21 93.996 52.20 48.1 2 37.72 1.48 3.4 21 133.162 96.20 45.7 3 36.05 2.24 4.5 21 176.243 137.10 42.4 4 33.75 2.98 5.6 21 219.325 169.60 39.7 5 31.95 3.71 6.3 21 246.741 198.50 36.95 6 29.77 4.49 7.1 21 278.073 221.70 33.67 7 27.63 5.21 7.3 21 285.906 235.690 30.17 8 25.27 5.95 8.5 21 332.904 241.360 26.1 9 22.43 6.69 8.6 24 336.821 234.90 20.64 10 18.64 7.41 8.8 25 344.654 206.4

0 39.7 5 31.95 3.71 6.3 24 246.741 198.51 43.7 5 34.97 4.1 7.4 24 289.822 218.52 47.5 5 37.93 4.47 9.2 24 360.320 237.53 51.1 5 39.9 4.81 10.1 24 395.568 255.54 55 5 42.8 5.18 11.6 24 454.316 2755 58.1 5 45.4 5.49 13.3 25 520.897 290.56 61.3 5 47.7 5.79 14.9 27 583.561 306.57 64.5 5 50.2 6.1 16.9 28 661.892 322.5

0 39.7 5 31.95 3.71 6.3 24 246.741 198.5

-1 35.23 5 28.13 3.3 5.6 24 219.325 176.15-2 31.03 5 24 2.91 4.5 24 176.243 155.15-3 26.65 5 21.33 2.49 3.5 24 137.078 133.25-4 22.35 5 18.25 2.07 2.7 25 105.746 111.75-5 18.06 5 14.96 1.68 2 25 78.330 90.3-6 13.95 5 11.82 1.26 1.5 26 58.748 69.75-7 9.51 5 8.44 0.88 1.1 28 43.082 47.55

159

Fwd Ex. Torque Power Curves @ 374rpm

0

200

400

600

800

0 5 10 15 20

Torque (Nm)

Pow

er (W

)

Pow er in W DC Pow er W

Torque & Current Curves @ 374rpm

0

2

4

6

8

10

12

14

16

18

0 5 10 15

DC Current (A)

Torq

ue (N

m)

No Ex I_DC Fwd Ex @ 5A I DCRev Ex @ 5A I DC

N0 Excitation Load IV Curve @ 374rpm

0

10

20

30

40

50

60

0 2 4 6 8 10 12

Current (A)

Vol

tage

(V)

Torque & Power Curves @ 374rpm

050

100150200250300350400

0 5 10

Torque (Nm)

Pow

er (W

)

Pow er in W DC Pow er W

160

V I Excitation Curves @ 374rpm

0

10

20

30

40

50

60

70

0 5 10 15

DC Current (A)

DC

Vol

tage

(V)

No Ex V_ DCFwd Ex V_DC @ 5A DCRev Ex V_DC @5A_DC

161

Static air gap flux density measured with digital millitesla meter per pole Pole 1

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0001 -8.43 -10.09 -10.75 -9.74 -7.96 -1 5.59 -7.52 -8.97 -9.55 -8.44 -6.87 -2 8.97 -6.71 -8.11 -8.44 -7.59 -6.09 -3 12.83 -5.82 -6.95 -7.29 -6.62 -5.46 -4 16.45 -5.15 -6.01 -6.25 -5.69 -4.63 -5 21.2 -4.12 -4.85 -5.05 -4.61 -3.82

1 -4.61 -9.13 -11.33 -11.9 -10.82 -8.79 2 -8.97 -10.13 -12.47 -13.15 -11.88 -9.59 3 -12.75 -11.05 -13.64 -14.43 -13.04 -10.48 4 -16.76 -12.06 -14.97 -15.78 -14.12 -11.41 5 -20.7 -12.59 -15.98 -17.21 -15.43 -12.29

Pole 2

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0001 8.44 10.25 10.88 9.78 8.07 -1 5.25 7.52 9.22 9.84 8.85 7.19 -2 9.92 6.38 7.97 8.34 7.62 6.21 -3 13.39 5.63 6.97 7.34 6.73 5.45 -4 17.34 4.79 5.94 6.21 5.64 4.62 -5 21.5 4.08 5.02 5.23 4.78 3.87

1 -4.88 9.01 11.31 11.89 10.92 8.35 2 -9.27 9.97 12.49 13.21 11.75 9.71 3 -13.15 11.02 13.57 14.49 13.02 10.62 4 -17.4 12.03 15.01 15.91 13.98 11.7 5 -21.4 12.43 16.06 17.27 15.48 12.54

Pole 3

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0002 -8.21 -10 -10.29 -9.19 -7.65 -1 5.79 -7.13 -8.61 -8.84 -8.15 -6.84 -2 9.51 -6.31 -7.64 -7.78 -6.96 -5.75 -3 13.39 -5.31 -6.43 -6.54 -5.91 -4.85 -4 17.37 -4.31 -5.33 -5.42 -4.87 -4.06 -5 21.5 -3.05 -3.87 -4.31 -3.82 -3.17

1 -5.05 -8.31 -10.27 -10.48 -9.56 -7.23 2 -9.13 -9.23 -11.36 -11.71 -10.41 -8.01 3 -13.06 -10.18 -12.33 -12.91 -11.52 -9.98 4 -17.09 -11.1 -13.89 -14.25 -12.62 -10.92 5 -21.1 -12.01 -15.23 -15.61 -13.75 -11.87

162

Pole 4

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0002 6.92 8.76 8.85 7.96 6.64 -1 5.42 5.87 7.31 7.46 6.75 5.39 -2 9.63 4.95 6.08 6.17 5.54 4.51 -3 13.61 3.96 5 5.04 4.48 3.68 -4 17.8 3.08 3.8 3.84 3.49 2.75 -5 21.8 2.16 2.64 2.77 2.54 2.08

1 -5.03 7.94 9.92 9.98 9.04 7.56 2 -9.16 9.08 11.23 11.36 10.11 8.56 3 -13.58 10.32 12.81 12.84 11.45 9.59 4 -17.51 11.34 13.88 14.38 12.66 10.89 5 -21.6 12.02 15.84 16.14 14.11 11.95

Pole 5

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 0.0002 -7.11 -8.54 -9.12 -8.41 -7.04 -1 5.66 -6.02 -7.14 -7.63 -7.12 -5.71 -2 9.86 -4.97 -6.1 -6.31 -5.82 -4.79 -3 13.82 -4.02 -4.79 -5.05 -4.71 -3.8 -4 17.55 -3.21 -3.71 -3.99 -3.58 -2.95 -5 21.9 -2.24 -2.58 -2.67 -2.56 -2.11

1 -4.84 -8.09 -9.72 -10.34 -9.43 -7.86 2 -9.23 -9.11 -10.96 -11.76 -10.81 -8.83 3 -13.14 -9.89 -12.33 -13.21 -12.05 -9.91 4 -17.29 -11.01 -13.81 -14.82 -13.51 -11.04 5 -21.6 -12.26 -15.43 -16.68 -15.21 -12.25

Pole 6

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0003 7.47 8.87 8.91 7.84 6.57 -1 5.63 6.33 7.38 7.54 6.71 5.58 -2 9.38 5.35 6.46 6.55 5.77 4.76 -3 13.7 4.36 5.21 5.31 4.76 3.67 -4 17.68 3.58 4.17 4.24 3.77 2.92 -5 21.4 2.75 3.19 3.31 2.93 2.49

1 -4.83 8.23 9.91 9.97 8.98 7.48 2 -9.35 9.33 11.18 11.51 10.08 8.27 3 -13.25 10.35 12.49 12.65 11.35 9.15 4 -17.26 11.43 13.59 14.08 12.34 10.15 5 -21.6 12.36 15.32 15.39 13.59 11.32

163

Pole 7 Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0002 -7.42 -9.18 -9.48 -8.63 -7.25 -1 6.08 -6.39 -7.85 -8.14 -7.29 -6.15 -2 9.88 -5.61 -6.79 -7.04 -6.32 -5.33 -3 13.96 -4.69 -5.71 -5.86 -5.29 -4.41 -4 17.99 -3.92 -4.65 -4.8 -4.29 -3.59 -5 22.2 -3.03 -3.57 -3.76 -3.41 -2.94

1 -4.8 -8.35 -10.32 -10.42 -9.69 -8.15 2 -9.91 -9.41 -11.54 -11.92 -10.81 -8.86 3 -13.25 -10.3 -12.54 -12.98 -11.8 -9.64 4 -17.65 -11.36 -13.95 -14.4 -12.91 -10.67 5 -21.9 -12.03 -15.45 -15.89 -14.29 -11.72

Pole 8

Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5

0 -0.0003 7.05 8.67 8.98 8.24 6.69 -1 6.23 6.05 7.43 7.71 6.92 5.76 -2 9.83 5.29 6.51 6.71 6.03 4.98 -3 14.33 4.34 5.26 5.48 4.92 4.23 -4 17.95 3.74 4.41 4.52 4.08 3.43 -5 22.3 2.87 3.36 3.48 3.19 2.69

1 -5.15 7.82 9.71 10.16 9.25 7.62 2 -9.91 8.86 10.97 11.51 10.38 8.62 3 -13.61 9.76 12.08 12.64 11.43 9.42 4 -17.96 10.75 13.33 13.98 12.66 10.07 5 -22.2 11.76 14.54 15.49 13.94 10.98

Average pole flux density in milli Tesla Iex Pt1 Av Pt2 Av Pt3 Av Pt4 Av Pt5 Av

0 7.63 9.30 9.66 8.72 7.23-1 6.60 7.99 8.34 7.53 6.19-2 5.70 6.96 7.17 6.46 5.30-3 4.77 5.79 5.99 5.43 4.44-4 3.97 4.75 4.91 4.43 3.62-5 3.04 3.64 3.82 3.48 2.901 8.36 10.31 10.64 9.71 7.882 9.39 11.53 12.02 10.78 8.813 10.36 12.72 13.27 11.96 9.854 11.39 14.05 14.70 13.10 10.865 12.18 15.48 16.21 14.48 11.87

164

Average permanent mde flux density Curve

0.002.004.006.008.00

10.0012.00

0 1 2 3 4 5 6

Pole teeth poistion

Flux

den

sity

(mT)

0 A

Average Buck mode flux density

0.00

2.00

4.00

6.00

8.00

10.00

12.00

0 1 2 3 4 5 6

Pole teeth position

Flux

den

sity

(mT) 0A

-1 A-2 A-3 A-4 A-5 A

165

Average Boost mode flux density

0.002.004.006.008.00

10.0012.0014.0016.0018.00

0 1 2 3 4 5 6

Pole teeth position

Flux

den

sity

(mT) 0A

1 A2 A3 A4 A5 A

Buck & Boost static average flux density

0.00

2.00

4.00

6.00

8.00

10.00

12.00

14.00

16.00

18.00

0 1 2 3 4 5 6

Pole teeth positions

Flux

den

sity

(mT)

0 A-1 A-2 A-3 A-4 A-5 A1 A2 A3 A4 A5 A

166

Static air gap flux density

0.00

5.00

10.00

15.00

20.00

-6 -4 -2 0 2 4 6

Exciting current (A)

Air

gap

flux

dens

ity (m

T)

Pt1 AvPt2 AvPt3 AvPt4 AvPt5 Av

Open circuit dynamic flux density

0

2

4

6

8

10

12

-6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6

Exciting current (A)

Flux

den

sity

(mT)

149 rpm

264 rpm

374 rpm

598 rpm

750 rpm

167

Open Circuit test ( V_dc ) Iexc Speed In RPM Cut_in Mid_range Cut_out Sync Above Sync Speed 149 374 598 750 1000 1200 1500 1785

0 23 56 91 115 152 187 231.7 2771 26.6 63 101 127 171 207 257.2 3082 28 69 111 140 187.5227.5 283 3383 31 75 121 152 203.8244.7 307 3684 33 81 131 164 220264.8 331.6 3985 35 87 141 176 236.9283.8 355.5 4286 38 93 150 187 250.7302.2 379.4 4557 40 98 159 200 267322.3 402 4850 23 56 91 115 152 187 231.7 277

-1 21 51 83 104 139.8 167 208.8 250-2 18 45 72 90 122.5146.7 182.9 218-3 16 35 62 78 105.2126.4 155.5 187-4 13 32 52 66 88.8104.5 130.9 157-5 10 26 42 52 70.8 85 105.6 127-6 8 20 32 40 54 64.5 79.5 93-7 6 14 23 28 38.6 44.5 54 58

Open circuit buck & boost voltage curves

0

100

200

300

400

500

600

0 500 1000 1500 2000

Speed (rpm)

Vol

tage

(V)

0 A1 A2 A3 A4 A5 A6 A7 A-1 A-2 A-3 A-4 A-5 A-6 A-7 A

168

Short Circuit test ( I_dc ) Excitation Current Speed In RPM Cut_in Mid_range Cut_out Sync 149 374 598 750

0 10 12 12.5 12.51 11 13 13.5 142 11.5 14.7 15 153 13 16 16.5 16.54 14.7 16.5 17.5 17.55 16 18 19 196 17 19.5 20.5 20.57 21 22 220 10 12 12.5 12.5

-1 7 10.7 11 11.5-2 7.5 9.5 9.7 9.5-3 6.5 7.8 8 8-4 6 6.8 6.5 6.5-5 5 5.5 6 5.5-6 4.5 4 4.5 4.5-7 2.5 3.5 3.5 3.5

Short circuit buck & boost current curves

0

5

10

15

20

25

0 200 400 600 800

Speed (rpm)

Cur

rent

(A)

0 A

1 A

2 A

3 A

4 A

5 A

6 A

7 A

-1 A

-2 A

-3 A

-4 A

-5 A

-6 A

-7 A

169

Excitation power loses ( I*I*R) Ex Current Armture Power in (A) resistance in (W)

0 4.18 0 1 4.18 4.18 2 4.18 16.72 3 4.18 37.62 4 4.18 66.88 5 4.18 104.5 6 4.18 150.48 7 4.18 204.82 8 4.18 267.52 9 4.18 338.58

10 4.18 418

Armature Excitation Power loses

0100200300400500

0 2 4 6 8 10 12Exciting current (A)

Pow

er lo

ses

(W)

Maintaining 24V on the lines(Open circuit test) Iexc (A) Vline (V) Vdc (V) Vrotor (V) Flux Speed Torque

4.55 24 33 18.1 3.91 149 0.8 1.55 24.8 33.7 6.82 3.83 200 0.8

-0.78 24.2 32.8 4.11 3.55 250 0.8 -1.23 24.2 32.8 5.88 3.47 264 0.8 -2.18 24.1 32.7 9.42 3.29 300 0.8 -3.15 24.1 32.7 12.95 3.03 350 0.8 -3.64 24 32.4 14.79 2.94 374 0.8 -3.97 24 32.5 16.12 2.83 400 0.8 -4.47 24.1 32.6 18.02 2.73 450 0.8 -4.88 24.4 33 19.74 2.58 500 0.8 -5.21 24 32.5 21.3 2.37 550 0.8 -5.54 23.9 32.4 22.4 2.25 600 0.9

-5 31.2 42.3 20.3 2.79 650 0.9 -5 33.3 45.3 20.4 2.82 700 0.9

-4.98 35.4 48.1 20.3 2.9 750 0.9

170

Open circuit Excitation IV curves at 24V V-line

0

20

40

60

100 200 300 400 500 600 700 800

Speed (rpm)

Vol

tage

s (V

) A

C &

DC

-10-50510

Exc

iting

cu

rren

t (A

)

Vline (V) Vdc (V) Vrotor (V) Iexc (A)

Load test maintaining 24V on the lines at 600rpm

Iexc (A) Iline (A) Idc (A) Vline (V) Vdc (V) Vrotor (V) Flux Torque

0.05 0.57 0.6 60.9 80.5 -0.159 6.37 1.80.05 0.57 0.6 60.9 80.5 -0.159 6.37 1.8

-5.04 0.72 1 24.4 30.9 20.2 2.77 1.4-4.6 1.47 2 24.3 30 18.67 3.03 1.7

-4.21 2.2 3 24 29.2 17.17 3.23 2.5-3.71 2.94 4 24.6 30.1 15.3 3.53 3.2-3.28 3.77 5 24.1 29.2 13.61 3.75 3.7-2.67 4.51 6 25 30.3 11.15 4.07 4.2-2.38 5.05 7 24.5 28.6 9.41 4.14 5-1.53 5.96 8 24.5 28.8 7.02 4.39 5.4-1.12 6.48 9 24.3 29.5 4.71 4.75 6.6-0.08 7.69 10 24.1 29.1 0.518 5.28 7.2

Excitation IV curves at 24V V-line 600rpm

0

10

20

30

40

50

60

70

80

0 2 4 6 8 10 12

DC Load current (A)

Vol

tage

s (V

) AC

& D

C

-6-4-2024681012

Cur

rent

s (A

)

Vline (V)

Vdc (V)

Vrotor (V)

Iline (A)

Idc (A)

Iexc (A)

171

Load fixed at Idc = 5.8A & I line 4A and also speed 750rpm V ex Vdc Vline Iex bbd Iex Torque

5.32 64.7 51.3 -1.05 -1.05 4.98.74 52.9 42.3 -1.99 -2 4.8

12.33 40.4 32.7 -3.01 -3.02 4.415.85 22.2 18.6 -3.96 -3.98 319.8 1.89 2.97 -4.97 -5 1.54.59 88.7 69.3 1 1.01 7.98.22 101.4 78.7 1.99 2 8.911.9 112.4 87.4 2.99 3 9.5

15.66 124.2 95.9 3.97 4 10.419.47 136.8 105.2 4.97 5 10.6

Load test at 5.8A and 750rpm

0255075

100125150

-24 -18 -12 -6 0 6 12 18 24

Exciting voltage (V)

Vol

tage

s (V

) A

C &

DC

-10

-5

0

5

10

15

Exc

iting

C

urre

nt (A

)

Vdc Vline Iex Torque

Open circuit test at different speeds without excitation

Speed V DC Back EMF V AC Torque Flux

149 22.1 0.506 16.82 0.7 0.28 200 29.3 0.536 22.37 0.7 0.28 250 36.8 0.569 28.02 0.8 0.28 264 38.5 0.577 29.38 0.8 0.28 300 43.8 0.606 33.33 0.8 0.28 350 51.1 0.651 38.82 0.9 0.28 374 54.7 0.671 40.5 0.9 0.28 400 58.6 0.687 43.5 0.9 0.28 450 66.1 0.734 49 0.9 0.28 500 73.6 0.776 54.6 0.9 0.28 550 80.6 0.815 59.8 0.9 0.28 598 87.4 0.856 64.9 0.9 0.28

172

650 95.2 0.904 70.6 0.99 0.28 700 103.1 0.953 76.4 0.99 0.28 750 110.5 1 82 1 0.28

Speed VS open circuit Voltages

0

20

40

60

80

100

120

100 300 500 700 900Speed (rpm)

Sta

tor v

olta

ges

(V)

V

AC

& V

DC

0

0.2

0.4

0.6

0.8

1

1.2

Bac

k em

f (V

)

V DC V AC Back EMF

Open circuit test maintaining 42Vline for 48V battery bank

Speed Iexc Vrotor Vline Vdc Torque flux

149 5.53 23.1 25.4 34.8 0.8 5.79 200 5.44 22.6 33.9 46.1 0.9 7.81 250 5.33 22.4 42.1 57.1 0.9 9.31 264 4.51 19.04 42.2 57.3 0.9 9.08 300 2.93 12.71 42.4 57.7 0.91 8.42 350 1.14 5.72 42.4 57.7 0.9 7.69 374 0.42 2.66 42.1 57.1 0.9 7.28 400 -0.18 2.01 42.1 56.9 0.86 7.02 450 -1.16 5.72 42 56.8 0.85 6.56 500 -1.95 8.76 42 56.8 0.81 6.10 550 -2.58 11.12 42.1 57 0.8 5.77 600 -3.07 12.95 42.1 57 0.85 5.46 650 -3.5 14.59 42.2 57.2 0.78 5.13 700 -3.92 16.11 42.1 57.1 0.8 4.84 750 -4.23 17.28 42.1 57 0.81 4.64 800 -4.51 18.42 42.2 57.2 0.75 4.47 850 -4.78 19.47 42.1 57.2 0.8 4.21 900 -5.02 20.5 42 57 0.81 4.05 950 -5.2 21.2 42.2 57.4 0.8 3.88

1000 -5.4 21.9 42 57 0.8 3.68 1050 -5.4 21.8 44.6 60.7 0.8 3.73 1100 -5.4 21.7 46.7 63.6 0.8 3.79 1150 -5.4 21.7 48.9 66.5 0.7 3.76 1200 -5.4 21.7 51 69.5 0.75 3.80

173

Open circuit test maintaining 42V Vline for 48V battery bank

01020304050607080

0 200 400 600 800 1000 1200 1400

Speed (rpm)

Indu

ced

volta

ges

(Vro

tor,

Vlin

e &

Vdc

)

-8

-3

2

7

Iexc

(A),

flux(

mT)

&

torq

ue(N

m)

Vrotor

Vline

Vdc

Torque

flux

Iexc

Open circuit test at different speeds without excitation Speed Back emf Vdc Vline Flux Torque

149 0.196 21.5 15.85 2.90 0.7200 0.254 28.8 21.5 3.83 0.78250 0.31 35.7 26.4 4.87 0.85264 0.324 37.8 28.1 5.06 0.9300 0.367 43.1 31.9 5.53 0.85350 0.423 50.6 37.2 6.10 0.9374 0.451 54.1 39.8 6.42 0.8400 0.481 57.9 42.8 6.66 0.9450 0.533 64.9 47.9 7.04 0.9500 0.59 72.3 53.3 7.37 0.9550 0.647 79.7 58.7 7.81 0.9598 0.699 86.7 63.8 8.05 0.99650 0.762 95.1 70 8.33 0.91700 0.814 102.1 75.1 8.54 1750 0.867 109.1 80.2 8.67 1

174

Generator open-circuit voltage magnitudes as per test at 750rpm

over a range of tests Test NO: V-DC V-Line % Drop

1 117.5 88 0.002 115 86.14 2.113 113.65 85.14 3.254 110 82.44 6.325 109.1 81.77 7.086 108.4 81.25 7.67

O/C votage characteristics over a range of tests (750rpm)

70

80

90

100

110

120

0 2 4 6 8

Test NO:

Vol

tage

(AC

& D

C) i

n V

olts

V-DC V-Line

Synchronous line current error chart Line current error comparison chart Ir-Iy Iy-Ib Ib-Ir

0.01 0.05 0.06 0.2 0.21 0.01

0.16 0.17 0.01 0.18 0.12 0.06 0.11 0.04 0.07 0.08 0.05 0.03 0.07 0.03 0.04 0.01 0.03 0.04 0.01 0.02 0.01 0.01 0 0.01 0.02 0.02 0.04

Maximum percentage error is 3.3%

175

Iexc V12 V23 V31 Vline Vrotor Ia Ib Ic IR IY IB Flux (mT) P (kW) S (kVA) Pf T (Nm) 5 86 87 87 87.7 20.3 7 7 7 7.21 7.21 7.28 10.75 -1 1 0.98 15.7

4.5 86 86 88 86.9 19 7 7 7 6.95 6.94 7.05 10.35 -1 1 0.98 15.54 84 84 86 86.3 16.55 7 6 7 6.92 6.91 6.98 10.03 -1 1 0.99 15.453 84 83 85 85.4 12.65 7 6 7 6.73 6.71 6.73 9.45 0 1 1 15.32 83 82 84 83.6 8.53 7 6 7 6.83 6.78 6.89 8.79 -1 1 -0.98 151 80 81 82 82.3 4.54 7 7 7 7.16 7.08 7.23 8.19 -1 1 -0.93 14.90 79 80 81 80.9 0.26 8 8 8 7.81 7.71 7.98 7.57 -1 1 -0.87 14.8

-1 77 77 78 79.2 4.7 8 8 8 8.51 8.32 8.56 6.66 -1 1 -0.78 14.7-2 76 77 78 76.9 8.71 9 9 9 9.76 9.54 9.76 5.45 -1 1 -0.67 14.5

-5 74 74 75 75.8 20.5 8 8 8 8.15 8.16 8.21 4.19 0 1 -0.49 8.1-4 76 76 78 78.1 16.7 6 6 6 6.26 6.06 6.27 4.69 0 1 -0.57 8-3 78 77 78 79.2 12.68 6 6 6 5.84 5.68 5.85 5.32 0 1 -0.66 8-2 78 78 79 79.9 8.51 6 5 5 5.45 5.27 5.39 5.95 0 1 -0.76 8.2-1 79 79 81 81.2 4.35 5 4 4 4.62 4.51 4.55 6.47 0 1 -0.84 8.10 81 80 82 82 0.43 4 4 4 4.23 4.15 4.2 6.82 0 1 -0.92 8.11 81 82 83 83.2 4.57 4 4 4 4.01 3.94 3.97 7.26 0 1 -0.99 8.12 83 83 84 84.2 8.52 4 4 4 3.94 3.93 3.9 7.7 0 1 1 8.13 84 85 85 85.7 12.51 4 4 4 4.14 4.13 4.15 8.15 0 1 0.95 8.14 85 85 86 86.7 16.31 4 4 4 4.39 4.38 4.38 8.49 0 1 0.84 8.25 86 87 87 87.8 20.7 5 5 5 4.92 4.94 4.96 9.04 0 1 0.73 8

Synchronous load test

176

Appendix C

Tests boards-data

Line voltage harmonics

Harmonic magnitude as a % of the fundamental amplitude2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40 42 44 46 48 50

0.0%

0.4%

0.8%

1.2%

1.7%

2.1%

2.5%

2.9%

3.3%

3.7%

4.2%

Line voltage harmonic chart at 598rpm.

Harmonic magnitude as a % of the fundamental amplitude2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40 42 44 46 48 50

0.00%

0.08%

0.16%

0.24%

0.31%

0.39%

0.47%

0.55%

0.63%

0.71%

0.78%

Line current harmonic chart at 598rpm.

177

Load current influence on ripple voltage at zero

exciting current

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75000 Ripple waveform at 0A load current and at 0A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75010 Ripple waveform at 1A load current and at 0A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75020 Ripple waveform at 2A load current and at 0A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75030 Ripple waveform at 3A load current and at 0A exciting current when speed is 750rpm.

178

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75040 Ripple waveform at 4A load current and at 0A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75050 Ripple waveform at 5A load current and at 0A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 1 ms RPL75090 Ripple waveform at 9A load current and at 0A exciting current when speed is 750rpm. From waveform RPL75000 up to waveform RPL75090, it is clear that the ripple voltage decreases with the increasing load current. Exciting current influence on ripple voltage at 2A load current when the generator speed is maintained at 750 rpm

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RPL75021 Ripple waveform at 2A load current and at 1A exciting current when speed is 750rpm.

179

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RPL75022 Ripple waveform at 2A load current and at 2A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RPL75023 Ripple waveform at 2A load current and at 3A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RPL75024 Ripple waveform at 2A load current and at 4A exciting current when speed is 750rpm.

T1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RPL75025 Ripple waveform at 2A load current and at 5A exciting current when speed is 750rpm. With reference to waveform RPL75021 up to waveform RPL75025, it is clear that the ripple voltage decreases with the increasing exciting current.

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 50 Volt 5 ms 2) Ch 2: 10 mVolt 5 ms

Generated voltage waveform (CH1) and current waveform (CH2) with unity star load.

T

2 >2 >2 >2 >

2) Ch 2: 1 Volt 5 ms

WL2750NX Line current 2A with 6-pulse bridge

180

T

2 >2 >2 >2 >

2) Ch 2: 1 Volt 5 ms

WL5750NX Line current 5A with 6-pulse bridge

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 2 Volt 10 ms 2) Ch 2: 2 Volt 10 ms

Generator speed pulses waveforms Rotor excitation control waveforms

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 5 Volt 250 us 2) Ch 2: 2 Volt 250 us

Ramp and PWM waveforms

TT

TT

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us

DC/DC converter no load drive signals

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us

DC/DC converter on load drive signals

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us

DC/DC converter transformer primary waveforms

181

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us

DC/DC converter’s miss matched MOSFETs drive signals

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us

Miss matched MOSFETs effect on DC/DC transformer primary waveforms

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us

Escalated MOSFETs miss match effect on the converter drive signals.

T

T

1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us

Escalated miss matched MOSFETs effect on DC/DC transformer primary waveform

T1 >1 >1 >1 >

1) Ch 1: 20 Volt 2.5 us DC/DC transformer secondary waveform for matched MOSFETs Primary load Secondary load 20V 2A 39.2V 0.83A 20V 2.21A 39.2V 1A 20V 11.93A 38.81 5A

T

1 >1 >1 >1 >

1) Ch 1: 20 Volt 2.5 us Converter 40V DC voltage waveform

182

Rotor back emf waveforms without and with freewheeling diode and zerner diode

T

1 >1 >1 >1 >

1) Ch 1: 10 Volt 2 ms

RNX75020 Open circuit back emf at 750rpm, 2A load current and 0A exciting current

T

1 >1 >1 >1 >

1) Ch 1: 10 Volt 2 ms

RNX75030 Open circuit back emf at 750rpm, 3A load current and 0A exciting current

T

1 >1 >1 >1 >

1) Ch 1: 10 Volt 2 ms

RNX75040 Open circuit back emf at 750rpm, 4A load current and 0A exciting current

TT1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RWX75010 The waveform RWX75010 is of the induced back EMF when a freewheeling diode is connected across the rotor terminals. The negative pulses of the EMF are clipped off with a freewheeling diode. This is because every pulse forward biases the diode. During the clipping period, the diode current flows in to the rotor. It flows for as long as the excitation source voltage is less than the back EMF. The sharp positive pulses reverse bias both the freewheeling diode and the blocking diode. Although the positive pulses are blocked by the two diodes, their magnitudes still need to be limited to magnitudes below freewheeling diode avalanche voltage Vdav. Thus, zener-clipping diode is introduced by connecting a zener diode across the free wheeling diode. The zener voltage Vz magnitude has to be greater than that of the excitation voltage Vex to avoid zener continous conduction as it increases losses on board. That is, Vex < Vz < Vdav (diode avalanche voltage).

183

TT1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RWX75020 Back emf at 750rpm, 2A load current, 0A exciting current and with freewheeling diode.

TT1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RWX75030 Back emf at 750rpm, 2A load current, 0A exciting current and with freewheeling diode.

TT1 >1 >1 >1 >

1) Ch 1: 10 Volt 2.5 ms RWX75040 Back emf at 750rpm, 4A load current, 0A exciting current and with freewheeling diode.

T

1 >1 >1 >1 >

1) Ch 1: 5 Volt 1 ms ONZ75020 No exciting current waveform with freewheeling diode and no zerner diode at 750rpm, 2A load current

T1 >1 >1 >1 >

1) Ch 1: 5 Volt 1 ms OWZ75020 Back emf at 750rpm, 2A load current, 0A exciting current and with freewheeling diode and zerner diode

T1 >1 >1 >1 >

1) Ch 1: 5 Volt 1 ms ONZ75040 No exciting current waveform with freewheeling diode and no zerner diode at 750rpm, 4A load current

184

T1 >1 >1 >1 >

1) Ch 1: 5 Volt 1 ms OWZ75040 Back emf at 750rpm, 4A load current, 0A exciting current and with freewheeling diode and zerner diode

185

Switched exciting voltage and current waveforms

T1 >1 >1 >1 >

1) Ch 1: 20 Volt 2.5 ms LWX75051 Exciting voltage at 750rpm, 5A load current and 1A exciting current

T

T1 >1 >1 >1 >

2 >2 >2 >2 >

1) Ch 1: 100 mVolt 250 us 2) Ch 2: 5 Volt 250 us

FGPWM Exciting current ripple at 30% duty cycle

186

Appendix D

Schematics test results

187

188

189

190

191

192

193

Figure D.1 ULN2803 representative schematic diagram

Table D.1

Ferrite geometries offer a wide selection in shapes and sizes. When choosing a core for power applications, parameters shown in Table D.1 should be evaluated.

Figure D.2 shows the reduction in flux levels for MAGNETICS “P” ferrite material necessary to maintain constant 100mW/cm³ core losses at various frequencies, with a maximum temperature rise of 25°C.

194

Figure D.2

195

196

Flexible PVC Insulated Stranded Copper Panel Wire.

Conductor Cross Section Current Rating

0.2 mm² 0.5 Amperes

0.5 mm² 3 Amperes

0.75 mm² 6 Amperes

1.0 mm² 10 Amperes

1.5 mm² 16 Amperes

2.5 mm² 25 Amperes

4.0 mm² 32 Amperes

6.0 mm² 41 Amperes

10.0 mm² 55 Amperes

Manufactured to SABS 1574 1992 Last Updated on 10th of February, 2000

197

V-BE V- CE

0 15.030.001 15.030.05 15.030.1 15.03

0.15 15.030.2 15.01

0.25 15.010.3 15.01

0.35 15.010.4 15.01

0.45 150.5 14.99

0.55 14.970.6 14.79

0.65 13.530.7 4.41

0.75 0.090.8 0.06

0.85 0.060.9 0.06

0.95 0.061 0.06

1.05 0.061.1 0.06

1K 1K

2K

+15V

VV

2N3904 Switching voltage test

0.01

0.1

1

10

100

0.01 0.11 0.21 0.31 0.41 0.51 0.61 0.71 0.81 0.91

V BE (V)

V C

E (V

)

198

Opto Isolator 4N25 (IV)F Curves Test I fwd V fwd Step mA V

12 0 0.82 12 0.02 0.91

11.98 0.1 0.97 11.38 0.3 1.02 11.24 0.5 1.05 11.09 0.7 1.06 10.94 0.9 1.07 10.76 1.1 1.08 10.6 1.3 1.085 10.4 1.5 1.09

10.19 1.7 1.096 10 1.9 1.1

9.81 2.1 1.104 9.61 2.3 1.108 9.39 2.5 1.111 9.18 2.7 1.114 8.95 2.9 1.117 8.7 3.1 1.119

8.45 3.3 1.122 8.3 3.5 1.124

8.08 3.7 1.126 7.81 3.9 1.128 7.62 4.1 1.131 7.41 4.3 1.132 7.18 4.5 1.134 6.92 4.7 1.136 6.76 4.9 1.138 6.5 5.1 1.139

6.31 5.3 1.141 6.06 5.5 1.142 5.86 5.7 1.144 5.66 5.9 1.145 5.44 6.1 1.146 5.26 6.3 1 5.06 6.5 1.149 4.84 6.7 1.15 4.66 6.9 1.152 4.5 7.1 1.153 4.4 7.3 1.154

4.21 7.5 1.155 3.85 7.7 1.156 3.74 7.9 1.158

3.52 8.1 1.159 3.44 8.3 1.16 3.28 8.5 1.161 3.15 8.7 1.162 2.97 8.9 1.163 2.88 9.1 1.164

4N25 I fwd Vc Curve

02468

101214

0 1 2 3 4 5 6 7 8 9 10

I fwd

Vc

(V)

I fwd mA

4N25 IV fwd Curve

0.85

0.9

0.95

1

1.05

1.1

1.15

1.2

0 1 2 3 4 5 6 7 8 9 10

I fwd

V fw

d

V fwd V

199

I ac Vo

0.085 1.60.1 1.650.5 1.67

1 1.681.5 1.72

2 1.772.5 1.82

3 1.883.5 1.94

4 1.994.5 2

5 2

AD736 IV Curves

1.5

1.6

1.7

1.8

1.9

2

2.1

0 2 4 6

Iac in (mA)

Vdc

out

(V)

V o @ 10Hz

I-line sen-voltage (mV)

0.01 11.30.51 141.03 17.51.51 21.32.02 25.42.5 28.8

3.04 33.83.54 37.64.07 41.64.51 44.75.01 48.85.44 51.65.96 55.66.51 59.67.04 63.77.42 66.58.01 71.18.59 75.5

Line current sensor characteristic curve

01020304050607080

0 1 2 3 4 5 6 7 8 9 10

Line current (A)

Vol

tage

(mV

)

sen-voltage (mV)

200

Vgs (v) Vds (v)

0 29.81 1 29.8 3 28.94

3.1 27.86 3.2 22.87 3.3 10.8

3.31 10.41 3.32 7.9 3.34 4.64 3.35 3.69 3.36 1.007 3.38 0.231 3.42 0.0537 3.43 0.0476 3.45 0.0354 3.47 0.028 3.48 0.025 3.5 0.021

3.54 0.014 3.58 0.0114 3.6 0.0099

3.68 0.0062 3.69 0.006 3.82 0.0037 3.9 0.0029

3.96 0.0024 4.04 0.002 4.14 0.0018 4.2 0.0016 4.4 0.0014 4.5 0.0013 4.7 0.0012 4.8 0.0011 5.5 0.001 6.3 0.0009

7 0.0009 8 0.0009 9 0.0008

10 0.0008 11 0.0008 12 0.0008 13 0.0008 14 0.0008 15 0.0008

IRF9640 Vgs vs Vds

0.0001

0.001

0.01

0.1

1

10

100

0 1 2 3 4 5 6 7 8 9 10 11 12

Vgs (v)

Vds

(v)

Vds

Vgs

Vin Vo

201

Opto Isolator 4N25 VI Curve I fwd Vc V rc IC MA V V mA

0.7 9.15 0.15 0.381 9.07 0.23 0.59

1.3 8.95 0.35 0.901.6 8.81 0.49 1.261.9 8.67 0.63 1.622.2 8.49 0.81 2.082.5 8.35 0.95 2.442.8 8.21 1.09 2.793.1 8.06 1.24 3.183.4 7.91 1.39 3.563.7 7.75 1.55 3.97

4 7.54 1.76 4.514.3 7.38 1.92 4.924.6 7.24 2.06 5.284.9 7.08 2.22 5.695.2 6.91 2.39 6.135.5 6.74 2.56 6.565.8 6.62 2.68 6.876.1 6.45 2.85 7.316.4 6.3 3 7.696.7 6.14 3.16 8.10

7 6.01 3.29 8.447.3 5.85 3.45 8.857.6 5.72 3.58 9.187.9 5.56 3.74 9.598.2 5.46 3.84 9.858.5 5.32 3.98 10.218.8 5.16 4.14 10.629.1 5.07 4.23 10.859.4 4.94 4.36 11.189.7 4.86 4.44 11.3810 4.77 4.53 11.62

10.3 4.63 4.67 11.9710.6 4.56 4.74 12.1510.9 4.51 4.79 12.2811.2 4.45 4.85 12.4411.5 4.4 4.9 12.5611.8 4.36 4.94 12.6712.1 4.31 4.99 12.7912.4 4.27 5.03 12.9012.7 4.23 5.07 13.00

13 4.2 5.1 13.0813.3 4.15 5.15 13.2113.6 4.12 5.18 13.2813.9 4.08 5.22 13.38

14.2 4.05 5.25 13.4614.5 4.01 5.29 13.5614.8 3.97 5.33 13.6715.1 3.94 5.36 13.7415.4 3.91 5.39 13.8215.7 3.88 5.42 13.90

16 3.85 5.45 13.97

Opto IV curves

0

2

4

6

8

10

12

14

16

0 5 10 15 20

I fwd (mA)

V (V

)

Vc I fwd Ic

+

_

0V

15V

V

+ 4.5V

- 4.5V

A

V

1K

10K

10K

1K

202

Freq Voltage Hz V

5 310 315 320 325 330 335 2.940 2.745 2.650 2.555 2.360 2.265 2.0570 1.975 1.780 1.685 1.590 1.395 1.25

100 1.2105 1.1110 1.05120 0.9130 0.8140 0.7150 0.6160 0.5170 0.45180 0.42190 0.4200 0.35

Low Pass Filter VF Curves

0

0.5

1

1.5

2

2.5

3

3.5

0 30 60 90 120 150 180 210Frequency (Hz)

Vol

tage

(V)

Voltage

203

BB 100 hall effect current sensor `+/- 7.5V supplied V-out I-rot

0.0362 0.18 0.0405 0.2 0.0585 0.3 0.0782 0.4 0.0956 0.5 0.1873 1 0.2845 1.5 0.3732 2 0.4646 2.5 0.551 3 0.639 3.5 0.742 4 0.816 4.5 0.923 5

BB-100 current test

00.10.20.30.40.50.60.70.80.9

1

0 1 2 3 4 5 6

Load current (A)

V-o

ut (V

)

BB-100 in circuit test V-out I-load

0.544 0.180.545 0.20.57 0.4

0.596 0.60.619 0.80.646 10.67 1.2

0.696 1.40.723 1.60.74 1.74

0.779 2.020.809 2.250.839 2.450.886 2.780.951 3.210.979 3.421.036 3.811.105 4.311.15 4.56

1.196 4.861.206 5

BB-100 circuit test

0.5

0.6

0.7

0.8

0.9

1

1.1

1.2

1.3

0 2 4 6

Load current (A)

V-o

ut (V

)

204

Appendix E

Project Test boards and schematics

Analog signals test board

I/O interface test board

205

Power test board

Power converter test board

206

Analog signals test board schematic diagram

1 2 3 4 5 6 7 8

A

B

C

D

87654321

D

C

B

A

Title

Number RevisionSize

A3

Date: 17-Nov-2003 Sheet of File: D:\Proschm&pcb\resaerch_A.ddb Drawn By:

CTL5

GN

D1

OUT 3

RST4

THR6

TRIG2 DISC 7

U1

R1

R2

Vin1

GN

D2

Vout 3RG1

Vin1 GN

D2

Vout 3

RG2

C1

C2 C3

C4

+B

-B -10

GND

VCC

-B

GND

R3

R4

R5

VR1

Q1

C5

C6

VCC

GND

GND

VCC

VR2

R6

3

21

411

U2A

5

67

U2B

10

98

U2C

12

1314

U2D

VCC

-10R7

R8

U5

Q2

GNDR9

VR3

VCC

GND

R22

VR4

VR5

R10

R11

R12

R13 R14

R15

R16

R17

R18

R19

R20

R21 3

21

411

U3A

5

67

U3B

10

98

U3C

12

1314

U4D

3

21

411

U4A

5

67

U4B

GND

GNDGND

-10

VCC

VCC

-10

LK

LK

-B

+15V

VgP

R25

R26

R27

R28

R29

R30

R31

R32 R33

VR6

3

21

411

U6A

5

67

U6B

10

98

U6C

12

1314

U6D

C10

C9

R34

R35

R36

R37R38

VR7

COM

COM +4.5 COM

+4.5

-4.5

+BL

+RL

U7

+5V

-B

PLC

R23

R24

C7

C8

+4.5

-4.5

COM

3

26

15

74 U8

3

26

15

74 U10

3

26

15

74 U12

C11

C12

C13

C14

C15

C16

COM

COM

COM

+4.5

+4.5

+4.5

+4.5

+4.5

+4.5

-4.5

-4.5

-4.5

-4.5

-4.5

-4.5

1234 5

678U9

1234 5

678U11

1234 5

678U13

+RL

-RL

+YL

-YL

+BL

-BL

R39R40

R41

R42

R43

R44

VR8 VR9

3

21

411

U14A5

67

U14B

10

98

U14C

C17

C18+4.5

-4.5COMCOMU15

COM

R45

R46 R47

R48

R49

3

21

411

U16A

5

67

U16B

10

98

U16C

12

1314

U16D

-B

+15V

VR10+15V

-B

IR

-B

+PWM

-PWM

R50R51

R52

R53

R54

R55

VR11 VR12

3

21

411

U17A5

67

U17B

10

98

U17C

C19

C20

+4.5

-4.5COMCOMU18

COM

R56

R57 R58

R59

VR13+15V

-B

IY

R61R62

R63

R64

R65

R66

VR14 VR15

3

21

411

U19A5

67

U19B

10

98

U19C

C21

C22

+4.5

-4.5COMCOMU20

COM

R67

R68 R69

R70

VR16+15V

-B

IB

R60

R71

-B

-B

U21-R

R73

R72 R74

R75

VR17+15V

-B

VRt

R76 -B

VRti

+Vcc 1-Vcc 2GND 3

GND 4Vo 5HE

OR1

OR2

OR3

3

26

15

74 OU1

3

26

15

74

OU21234 5

678OU3

-R

+9VVRti

-R

+9V

-B

+15V

VRt

3

26

1 5

74

U22R77

R78

VR18

C23

IRt

+15V

+15V-B

-B

-B

-B

123456

CON1

123456

CON2

123456

CON4

+B

-B

+15V+5V

+9V-R

+4.5-4.5

+RL-RL+YL-YL+BL-BL

VgP

S1S2S3S4+PWM-PWM

1 23 45 67 89 10

HD1

1 23 45 67 89 10

HD2

IRt VRtIR IYIB -B

PLC

S1S2S3S4-B

VRti

123456

CON3

Control Analog Signals Board

C.T. 'Moleli

-R

1 2

J2

R8a

D1

D2

D3

DZ2

DZ3

1

2

J1

12

OJ

1 2

OJ1

DZ1

S1a+BS_+B

PWM Generator

Line current sensors

Generator speed sensor

External component

-10V

VoVo

207

I/O interface test board schematic

1 2 3 4 5 6 7 8

A

B

C

D

87654321

D

C

B

A

Title

Number RevisionSize

A3

Date: 25-Oct-2003 Sheet of File: D:\Proschm&pcb\Research.ddb Drawn By:

2

31

411

U1A

5

67

411

U1B

9

108

411

U1C

1413

12

411

U1D

2

31

411

U2A

5

67

411

U2B

9

108

411

U2C

1413

12

411

U2D

2

31

411

U3A

5

67

411

U3B

9

108

411

U3C

1413

12

411

U3D

R1

R2

R3

R4

R5

R6

R7

R8

R9

R10

R11

R12

R13

R14

R15

R16

R17

R18

VR1

VR2

VR3

-B

+15V

-Vg

+Vg

-IRg

+IRg

-VB

+VB

a1

a3

a4

+5Vd

+5Vd

+5Vd

+5Vd +5Vd

+5Vd +5Vd

+5Vd

-Bd

-Bd

-Bd -Bd

-Bd -Bd

-Bd

-Bd

-B

-B

+15V

+15V

A

C

D

E

F G

H

BIRt VRt

IR IY

IB

1

2

3

4

5

Ain1 2

3 4

5

Ain

Ain

Ain

1234

CON1

1234

CON2

1234

CON3

+15V-B+5Vd-Bd

-Vg+Vg-IRg+IRg

-VB+VB

1 23 45 67 89 10

HD1

1 23 45 67 89 10

HD2

IRt VRtIR IYIB

-BPLC

a1a IRta3 a4VRt IRIY IB

S1S2S3S4S5S6S7S8

O1O2O3O4O5O6O7O8

2

36

74

8

DIS U4

2

36

74

8

DIS U5

2

36

74

8DIS U6

2

36

74

8DIS

U72

36

74

8DIS

U8

2

36

74

8DIS

U92

36

74

8DIS

U10

2

36

74

8DIS

U11

a1a

1 23 45 67 89 1011 1213 1415 1617 1819 20

HD6

A1

B2

C3

G16

G2A4

G2B5

Y0 15

Y1 14

Y2 13

Y3 12

Y4 11

Y5 10

Y6 9

Y7 7

U12

CLR1

LOAD9

ENT10

ENP7

CLK2

RCO 15

A3

QA 14

B4

QB 13

C5

QC 12

D6

QD 11

U13

CLR1

LOAD9

ENT10

ENP7

CLK2

RCO 15

A3

QA 14

B4

QB 13

C5

QC 12

D6

QD 11

U14

-B

-B

-B

IN 11

IN 22

IN 33

IN 44

IN 55

IN 66

IN 77

IN 88

DIODE CLAMP10 OUT 8 11OUT 7 12OUT 6 13OUT 5 14OUT 4 15OUT 3 16OUT 2 17OUT 1 18U18

+5Vd

O0O1O2O3O4O5O6O7

ABCDEFGH

S6S7S8

-Bd

R19

R20

R21

R22

R23

R24

R25

R26

R27

R28

R29

R30

R31

R32

R33

R34

-Bd

+5Vd

+5Vd

-Bd

-Bd

Qc0Qc1Qc2Qc3

TC

PLC

S5

S5

TCQc4Qc5Qc6Qc7

-Bd

-Bd

+5Vd

-Bd

C1

C2

C3-Bd

-Bd

S1 S2S3 S4S5 S6S7 S8

O1 O2O3 O4O5 O6O7 O8

VCCGND

1 23 45 67 89 10

HD5S1S2S3S4

Control I/O Interface Board

C.T. 'Moleli

Qc0Qc1Qc2Qc3Qc4Qc5Qc6Qc7

1 23 45 67 89 1011 1213 1415 16

HD3

1 23 45 67 89 1011 1213 1415 16

HD4

1 2

J1

1 2

J2

a1a

D1

D2

D3

D4

D5

D6

D7

D8

When single channel read analog mode is used jumber J2 has to be disconnected and J1 be connected.On the contrary, J2 has to be connected and J1 disconnected when multi channel (8 channel)read analog mode is used.

208

Power test board schematic diagram

1 2 3 4 5 6 7 8

A

B

C

D

87654321

D

C

B

A

Title

Number RevisionSize

A3

Date: 9-Nov-2003 Sheet of File: D:\Proschm&pcb\researchb.ddb Drawn By:

R7

PT1

PT2

R8

R9

Q1

R11

R12

R13

R14

PT3

Tp1a Tp2 Tp3

C4C7

C5C6

+15V

-B

NC1

CL2

CS3

Inv4

Ninv5

Vref6

-V7 NC 8Vz 9Vout 10Vc 11+V 12FC 13NC 14U2

R15

R16

R17

R18

R19 R20

R21

VR1

C8Tp5 Tp4Q3Tp3Tp6

C9 C10

R23

R24

R25

R26

R27

R28

R29

R30

R31

R32

R33

PT4

PT5

PT6

+B

+15V

Q5

Q6

+5V

Tp7 Tp8

-Vg +Vg

VgP

+VB -VB

R34

R35

R36

R37

Q8

Q9

R38

U4 U5

Rly1

+15V

-B

S4

+5V

-BS2

+5V

+5V

+15V

-B

R39

R40

R41

R42

R43

R44

R45 R46

VR2 1234 5

678U7

C11 C13

C15

C17

C19

C12C14

C16

C18

1 3

2

V V

GNDIN OUT

Reg1

1 3

2

V V

GNDIN OUT

Reg2

1 3

2

V V

GNDIN OUT

Reg3

1 3

2

V V

GNDIN OUT

Reg4

Q16

Q10

Q11

Q12

Q13

Q14

Q15

R47

U6

S1 +5V

+12V

+15V

+12V

+5V

Tp9 Tp10

+B

+15V

+12V

-B

-B

-B

C20C22

C21

R48

R49

R50

R59

R60

PT7

Q19

Q22

Q23

Q21

R51

R52R53

R54

C26

C25

U9R61

+PWM-PWM

+9V

-R

-R

S3-B

Rly2

+15V+5V

R55

R56

R57

R58

R62

R63

R64

C23

C24

VR3

3

21

411

U8A

5

67

U8B

10

98

U8C

12

1314

U8D

+9V

-R

VRti

LK

LK

+R Tp11 Tp12

+B

123456

CON1

123456

CON2

123456

CON7

1234

CON6

1234

CON5

Tp1

Tp2Tp3Tp4Tp5Tp6

Tp7Tp8

Tp9Tp10

Tp11Tp12

+B

+15V

+5V

-B

+R-R

+Vg-Vg

+9V

+PWM-PWM

Rly1Rly2

S1

S2S3S4

VgP

+VB

-VB

Control Power Board

C.T. 'Moleli

1

2

J1

VRti

1234

CON3

1234

CON4

+12V

VRx

3

21

411

U1A

5

67

U1B

10

98

U1C

12

1314

U1D

Vref

Vref X

X

R11a

R28a

R32a

Rx1

Rx2

Dx

12

3

U3A

DZ1

DZ2

DZ3

DZ4

D8

D9

D10

D11

D16

D17

D18

Q18

Q17

Q2 Q7

Q20

Q4

D7

R22

R10

Tp9

T10

dQ18

dQ17

-B

F1

S1

F2

FQ17

FQ18

Tp1

Tp1a

Tp8

+Load

All dotted line enclosed componets are external to the control power board

R22a

209

Power converter test board schematic

1 2 3 4

A

B

C

D

4321

D

C

B

A Title

Number RevisionSize

A4

Date: 25-Oct-2003 Sheet of File: D:\Proschm&pcb\converters.ddb Drawn By:

12

RL

12

YL

12

BL

12

Rly1

12

D7 Con

12

Rot12

Rtpol

12

Rly2

123

Trans

12

Hv1

12

Hv2

R1

R2

R3

R4

R5

R6

Rrly1

Rrly2

D1

D2

D3

D4

D5

D6

D12 D13

D14 D15

D7

Dbd

D16a

L1

L2

HVc1

HVc2C_Trans

Rly2

C1

C2

C3

Tp1

-B

+RL-RL

+YL-YL

+BL

-BL

+15V

Rly1

Q4E

+B

dQ17

dQ18

+B

dQ20

-R

-R

+R

+15vRly2

Rly1 is formed with 3_relays:

Rly1a Rly1b Rly1c

Rly2 is formed with 2_relays:

Rly2a Rly2b

All coils connected in parallel

All coils connected in parallel

Converter Board

C.T. 'Moleli

Rly1

A balanced star load is biult with R4, R5 and R6 as indicated by the schematic.

210

Generator test workstation

Test boards arrangement