E5BYTQ3

8/13/2019 E5BYTQ3

http://slidepdf.com/reader/full/e5bytq3 1/8

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 57, NO. 2, FEBRUARY 2009 331

Broadband Microstrip AntennaWith Left-Handed Metamaterials

Merih Palandoken, Andre Grede, and Heino Henke

Abstract—A novel type of microstrip antenna is proposed forcompact wideband wireless applications. The antenna is composedof six unit cells of left-handed metamaterial (LHM) and a dipoleelement. The dipole is directly connected to three of six LHM unitcells, which are arranged in a 2 3 antenna array form. In thisaspect, the proposed antenna is regarded as LHM loaded dipoleantenna. The antenna is matched with a stepped impedance trans-former and rectangular slot in the truncated ground plane. Thecoupled LH resonances and simultaneous excitation of differentsections of unit cells and dipole result into broad bandwidth. Theproposed antenna has a maximum gain of 1 dBi at 2.5 GHz.The measured return loss indicates 63% bandwidth for

over the band of 1.3–2.5 GHz. The overall size of LHM

loadedantenna is

at thecenterfrequency. The radiation of the electrically small LHM unit cells isalso demonstrated by the simulated radiation pattern, which is animportant concept for the antenna miniaturization.

Index Terms—Dipole antennas, left-handed metamaterials(LHMs), loaded antennas, periodic structure.

I. INTRODUCTION

T

HE demand on the portable mobile devices is increasing

progressively with the development of novel wireless

communication techniques. In that respect, compact size, lightweight, low profile and low cost are now quite important chal-

lenges to be accomplished by the designers for every wireless

mobile component. One of the most important components

of any wireless system is its radiating element. In addition

to the physical requirements, the emerging requirements of

the wireless systems are high directivity, large gain, efficient

and broadband operability of the antennas. Many broadband

techniques have been investigated in order to overcome the

trade off between the size of the antenna and minimum achiev-

able quality factor, which is dictated by Chu formulations

[1]. These techniques are mainly increasing the thickness of

the substrate, using different shaped slots or radiating patches[2]–[4], stacking different radiating elements or loading of the

antenna laterally or vertically [5]–[7], utilizing magnetodielec-

tric substrates [8] and engineering the ground plane as in the

case of EBG metamaterials [9].

Manuscript received January 28, 2008; revised August 19, 2008. Current ver-sion published March 20, 2009.

The authors are with the Technische Universitaet Berlin, 10587 Berlin, Ger-many (e-mail: [email protected]; [email protected];[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2008.2011230

Metamaterials are artificially structured materials providing

electromagnetic properties not encountered in nature. The elec-

trodynamics of hypothetical materials having simultaneously

negative permittivity and permeability was first theoretically

predicted by Veselago [10]. These materials are termed as “left-

handed materials (LHM)” due to the left-handedness of electric,

magnetic field and wave vector. There are quite interesting elec-

tromagnetic phenomena related with LHM including reversal of

Doppler effect, reversal of Goos-Hänchen effect, inverse Snells

law and reverse Cherenkov radiation [10].

A left-handed material was first implemented in a two di-

mensional periodic array of split ring resonators and long wirestrips by Smith [11]. The logical approach was to excite the split

ring resonators and wire strips in order to force the structure

to behave like magnetic and electric dipoles, respectively [12].

Since then, there have been large numbers of experimental in-

vestigations on the observation of this phenomenon. The effec-

tive electromagnetic parameters were also retrieved experimen-

tally and numerically from the transmission and reflection data

[15]–[19]. Rather than split ring resonators and wire strips, the

left-handed feature can also be realized with periodic loading

of conventional microstrip transmission lines with series capac-

itors and shunt inductors [12], [20]. Many microwave circuits

have been implemented by using this strategy such as com-pact broadband couplers [12], [21], broadband phase shifters

[20], compact wideband filters [22], compact resonator antennas

[23]–[26]. In terms of traveling wave antennas, a leaky wave an-

tenna consisting of a microstrip transmission line has also been

realized [12], [27]. Because its radiation principle depends on

the leakage from the transmission line, this type of antenna has

the length comparable to the wavelength. However, this kind

of LH leaky wave antenna has a very unique property of back-

fire-to-endfire frequency scanning capability with broadside ra-

diation, which is not possible for RH leaky wave antennas [12].

Recently, an unsymmetrically fed LH dipole antenna has been

demonstrated as an example for an electrically small antenna

[28].In this paper, a broadband planar antenna, which is composed

of a dipole and six LHM unit cells, is demonstrated. The antenna

is matched to 50 with the stepped impedance transformer and

rectangular slot in the truncated ground plane. By the utiliza-

tion of phase compensation and coupled resonance feature of

LHMs, the narrowband dipole antenna is operated at broader

bandwidth.

The structure of the electrically small LHM unit cell is de-

scribed together with the geometrical parameters. A one dimen-

sional dispersion diagram is numerically calculated by finite ele-

ment method (FEM) in Section II to prove the lefthandedness of

the proposed unit cell. The effective permittivity, permeability

0018-926X/$25.00 © 2009 IEEE

8/13/2019 E5BYTQ3


332 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 57, NO. 2, FEBRUARY 2009

Fig. 1. LHM unit cell geometry. (a) Front and (b) back side of one LHM unit cell with indicated geometrical parameters in Table I.

and refractive index parameters are also retrieved from the re-

flection and transmission data of one unit cell. In Section III,the configuration and operation principle of the proposed an-

tenna are explained. The simulated and measured reflection co-

efficients, radiation pattern and numerically computed radiation

parameters are presented.

The proposed broadband antenna is not electrically small.

The overall size is at the center

frequency 1.9 GHz. However, in addition to the broadband op-

eration of the proposed antenna, an important contribution of

the paper is to prove the resonance and radiation of the elec-

trically small unit cells, a key issue for the design of miniatur-

ized antennas. The efficiency of the antenna at max gain is 40%,

which is appreciably higher than that of conventional andedge excited dipoles.

II. LHM UNIT CELL DESIGN

The negative material parameters are synthesized by the si-

multaneous excitation of electric and magnetic dipoles in the

LHM unit cell. The original structure proposed in [11] consists

of a bulky combination of metal wires and split ring resonators

(SRR) disposed in alternating rows. The excited wires and SRRs

are electric and magnetic dipoles, thus creating the left-handed

behavior. Because the typical LHM designs are inherently in-

homogeneous, novel strategies to miniaturize the unit cell with

different topological and geometrical methods are important. A. Description of the Structure

LHM behavior implies small unit cells as compared to the

free space wavelength . The upper limit of the unit cell size

is one fourth of the guided wavelength [12]. One well-known

method of miniaturization is to increase the coupling between

the resonators. This strategy was chosen for the proposed struc-

ture, Fig. 1 with geometrical parameters given in Table I, where

the wire strips and spiral resonators (SR) are directly connected

with each other, on both sides of the substrate. Further,instead of

SRRs as in the original proposals, SRs are used [13], which have

half the resonance frequency of SRRs [14]. In the design, the

geometrical parameters of the front and back side unit cells arethe same, except a 0.6 mm shorter wire strip length on the front

TABLE I

GEOMETRICAL PARAMETERS OF LHM UNIT CELL (mm)

side. Different strip wire lengths lead to a smaller resonance fre-

quency and larger bandwidth, as discussed next in Section II-B.

The substrate material is nonmagnetic FR4-Epoxy with a rela-

tive permittivity of 4.4 and loss tangent of 0.02. The frequency

dispersive property of FR4 has not been taken into account in

the simulation. As indicated in Fig. 1(a), the inner ring of the

SR is connected to the outer one with a 0.2 mm wide line. Thewidth of the line is selected so as not to cause unwanted over-

lapping of the corner points of the inner and outer rings of SRs

because of poor printing resolution in the fabrication process.

Also, attention has to be paid to the change of dimensions of

the copper layers after the etching process, which might effect

the low loss transmission through the LHM. The validity of the

model is shown by retrieving the effective constitutive parame-

ters from S parameters and by the opposite direction of group

and phase velocity.

B. Simulation Results

1) One Dimensional Brillouin Diagram: To determine the

frequency interval of lefthandedness, a one dimensional Bril-louin diagram is studied. In order to obtain the dispersion rela-

8/13/2019 E5BYTQ3


PALANDOKEN et al.: BROADBAND MICROSTRIP ANTENNA WITH LHM 333

Fig. 2. Dispersion diagram of the proposed LHM structure.

tion of the infinite periodic structure, the cells must be excited

with the magnetic field perpendicular to the SR plane (z-direc-

tion), and the electric field in the direction of strip wires(x-di-

rection), Fig. 1. Therefore, the eigenfrequencies of a unit cell

are calculated with perfect magnetic boundaries (PMC) in z-di-

rection and perfect electric boundaries (PEC) in x-direction.

Periodic boundary conditions (PBC) are imposed in y-direc-

tion. The simulation was done with the FEM based commer-

cial software HFSS and is shown in Fig. 2. Oppositely directed

phase and group velocities are observed in the LH band between

2.15–2.56 GHz with 410 MHz bandwidth.

Alternatively, the same unit cell structure but with longer

strip wires on the front side leads to higher cutoff frequencies

(2.58–2.65 GHz) and a narrow LH passband (69.7 MHz). Also,

if the front and back side are chosen identically, the LH pass-

band is between 3.45–3.51 GHz, which is relatively narrow andwhich is at higher frequencies than for the proposed design. This

explains the use of a shorter wire strip on the front side of the

substrate, which reduces the resonance frequency and increases

the bandwidth.

2) Retrieval of Effective Parameter: In addition to the dis-

persion diagram, the effective constitutive parameters are also

retrieved from the scattering parameters of a one cell thick LHM

sample. Therefore, the lefthandedness of the unit cell is not only

proved with the opposite phase and group velocities as in Fig. 2,

but also with the values and the sign of the retrieved param-

eters. The reflection and transmission parameters are numeri-

cally calculated for x polarized and in y-direction propagatingplane waves. PEC and PMC boundary conditions are imposed in

x- and z- direction. The effective permittivity, permeability and

refractive index are retrieved from the simulated S parameters

and shown in Fig. 3. Due to the asymmetry of the unit cell, the

modified S parameter retrieval procedure [16] is implemented.

The effective parameters show similar frequency dispersion as

in [16]. There are important issues to be discussed about the fre-

quency dispersion of retrieved parameters. First of all, as shown

in Fig. 3, the imaginary part of the refractive index is negative.

It is due to the engineering convention for the time

harmonic fields in contrast to the proposed retrieval procedures

in [15]–[18], in which is selected. Second, the re-

trieval procedure leads in general to satisfying results—an ex-pected Lorentzian type magnetic resonance for —but unphys-

Fig. 3. Real and imaginary

part of retrieved effective parameters

of LHM: (a) complex refractive index, (b) complex permittivity, (c) complexpermeability.

ical artifacts also occur such as a positive imaginary part of .

The anti-resonance of the real part of near 1.8 GHz leads con-

sistently to a positive imaginary part [16]. This is an inherent ar-

tifact for inhomogeneous, periodic structures because of the fi-

nite unit cell size [29]. As a last remark, there is an LHresonance

near 1.94 GHz in S parameter simulation, which is smaller than

the lower cutoff frequency in the Brillouin diagram and is at-

tributed to the single cell simulation. The Bloch impedance of

the infinitely periodic LHM is no longer valid for an isolatedsingle cell.

8/13/2019 E5BYTQ3



Fig. 4. Surface eigencurrent distribution on the (a) front and (b) back side of one unit cell.

The LH band for retrieved parameters extends from 1.75 up

to 2.55 GHz. It is in good correspondence with simulated band

in the range from 2.17–2.53 GHz in terms of refractive indicies

calculated directly from the dispersion diagram in Fig. 2 [29].

The size of a unit cell is approximately 1/43 of at 2 GHz,

which is directly connected, in first approximation and ne-

glecting all coupling, to the total metallic length from the

open circuited SR to the short circuited wire strip. The varying

degree of coupling between the resonators shifts and broadens

the transmission band. If the electrically small unit cells are

excited by their eigencurrents, they represent effective radiating

elements and are key elements for the future aspects in the

antenna miniaturization.

III. ANTENNA DESIGN

A. Operation Principle

The operation principle of the antenna depends on the radia-

tion of the dipole antenna and the excitation of LHM unit cells

with the dipole field. The excitation of LH cells in their eigen-

modes causes the individual electric and magnetic dipoles to be

coupled in the same way as in the eigenmode simulation. These

unit cell dipoles are also radiation sources in addition to the ex-

citing dipole antenna even though they are designed as loads for

the dipole. The magnetic and electric dipole moments are ex-

pressed by the surface current density as in [30]

(1)

(2)

where is the displacement vector directed from the surface

current element to calculation point; is the current

element position; is the differential current carrying surface

element.

The current distribution in one unit cell is inhomogeneous as

shown in Fig. 4. For each unit cell, the electric and magnetic

dipoles are simultaneously excited in principle. However, the

magnetic dipoles are more effective than the electric ones. At

first, magnetic dipole fields do not cancel in the far field be-cause of inplane electrical coupling among the cells on the front

Fig. 5. (a)Top, (b)bottom geometry, and(c) prototypeof theproposed antenna.

and back side as in the case of same direction imaging of the

horizontal magnetic dipole on perfect electric plane. However,

electric dipoles tangential to the electric coupling plane are ac-

tually suppressed as in the case of reverse direction imaging of

horizontal electric dipole over perfect electric plane. The second

reason is that the current on the back side strip wire has partially

opposite directions and do not to excite the electric dipole as ef-

fectively as the magnetic dipole. As a last reason, the surface

current on the back side unit cell spirals in the same direction

as the surface current on the front side unit cell, thus enhancing

the effective magnetic dipole moment. In that respect, front and

back side unit cells are mainly magnetically coupled and the

back side unit cells can be considered as the artificial magnetic

ground plane for the front side unit cells. It also follows from

the Lorentzian type magnetic resonance in Fig. 3(c), which is

the dominating resonance in the retrieved effective parameters.However, the antenna radiates mainly in the dipole mode, which

is the reason why we call it as an LHM loaded dipole antenna.

B. Antenna Design

As a first step in the antenna design, the front and back side

unit cells were connected symmetrically with adjacent cells

in x-direction and periodically in the y-direction, see Fig. 1.

These requirements follow from the boundary condition in the

eigenmode simulation. Six unit cells were used without vertical

stacking and arranged in a 2 3 array, Fig. 5. As can be seen

from the figure, the front sides of unit cells are directly con-

nected to the dipole in order to increase the coupling from thedipole to the LH load. In that way, the impedance of the LH load

8/13/2019 E5BYTQ3



TABLE IIGEOMETRICAL PARAMETERS OF ANTENNA (mm)

Fig. 6. Measured (solid line) and simulated (dashed line) reflection coefficientof the proposed antenna.

is transformed by the dipole. The truncated ground plane leadsto a decreased stored energy because of lower field components

near the metallic interfaces (decreased effective permittivity).

A rectangular slot in the truncated ground plane allows for one

more degree of freedom for the matching network. The effect

of the slot can be modeled by a shunt element consisting of a

parallel LC resonator in series with the capacitance [22]. The

width of the slot is appreciably smaller than half a wavelength

in the substrate and is optimized together with the length. Geo-

metrical parameters are tabulated in Table II and a prototype is

shown in Fig. 5. All the design parameters have been optimized

for best return loss.

C. Experimental and Simulation Results

The reflection coefficient of the antenna was measured with

the vector network analyzer HP 8722C and is shown in Fig. 6

together with the simulation result. As can be seen from the

data, the bandwidth of 63.16% extends from approximately

1.3–2.5 GHz with the center frequency of 1.9 GHz. Unit cell

resonances can be clearly observed in the passband. The low

frequency ripples are attributed to the inaccurate modeling of

the coax-microstrip line transition. In summary, the measured

and simulated reflection coefficients are in good agreement.

There are nevertheless some issues to be discussed from themeasured and simulated results. First of all, in the experimental

Fig. 7. The footpoint impedance of LHM loaded dipole antenna (top line: sus-ceptance, bottom line: resistance).

TABLE IIIOVERALL SIZE AND RADIATION PARAMETERS OF CONVENTIONAL AND

LHM LOADED DIPOLE ANTENNAS

result, there are lower resonance frequencies than those of the

LH passband in Fig. 2, which is also the case in the simulated re-

flection coefficient. These lower resonance frequencies are due

to the direct coupling between the dipole antenna and LHM unit

cells and are not emerging from the LHM resonances. In order

to prove this reasoning, the current distribution in LHM unit

cells and dipole is examined. At 1.7 GHz, the dipole is stronger

excited than the LHM unit cells, which is obvious because of

out-of.-band resonance. In other words, the LH load impedance

is transformed by the dipole to match at this lower frequency.

Secondly, the bandwidth is enhanced by the fact that different

sections of the LHM cells and the dipole are excited at differentfrequencies. Still, the effect of the LH load is quite important

for broadband operation. In order to figure out the unique prop-

erty for broadbandness, the foot-point impedance of LH loaded

antenna is plotted in Fig. 7. The unit cell resonances are closer

to each other at the lower frequencies than at higher frequen-

cies. This unique property results in a broadband behavior at low

frequencies, which is not the case for RH operation. The same

reasoning can also be deduced from the dispersion diagram in

Fig. 2. Therefore, the coupled resonance feature of LHM unit

cells results in an antenna input impedance as smooth as in the

case of tapering. It is the main reason why the antenna is broad-

band [31], [32]. The topology of matching network is as impor-tant as the broadband load for the wideband operation [33].

8/13/2019 E5BYTQ3



(a) (b)

(c) (d)

Fig. 8. Normalized radiation patterns cross-polarization (o - light line) and co-polarization (+ - dark line) at 1.7 GHz in (a) y-z and (c) x-z plane, and at 2.3 GHzin (b) y-z and (d) x-z plane.

The third important issue is the radiation of electrically small

LHM cells. It could be verified not only from the current dis-

tribution and the return loss but also from the radiation pattern,

which is explained next. To prove the subwavelength radiation,

we simulated also a stub matched antenna with an electrically

very short feeding line and LHM unit cells, and confirmed the

radiation However, we do not include the results due to lim-

ited space. The antenna matching can be explained by the phasecompensation feature of LHM as for instance in the case for the

length independent subwavelength resonators [9] and antennas

[34].

The normalized radiation patterns of the antenna in the y-z

and x-z planes at 1.7 GHz and 2.3 GHz are shown in Fig. 8.

They are mainly dipole-like radiation patterns in E and H planes,

which is the reason to call the antenna an LHM loaded dipole

antenna. The radiation of the electrically small LHM unit cells

is also observed from the radiation pattern at 2.3 GHz. As it is

shown in Fig. 8(b), the more effective excitation of the LHM

cells at 2.3 GHz than at 1.7 GHz results in an asymmetric radi-

ation pattern because of the structure asymmetry along y axis.The cross polarization in the y-z plane is 8 dB higher at 2.3 GHz

than that at 1.7 GHz, see Fig. 2, because of LH passband res-

onance. The gain of the broadband antenna is unfortunately

small. The maximum gain and directivity are 1 dBi and 3 dB

with 40% efficiency at 2.5 GHz, respectively. The gain, direc-

tivity and efficiency are plotted in Fig. 9. For the comparison of

the overall size and radiation parameters of the proposed design

with the conventional dipole antennas, two edge excited and

dipole antennas are designed. The simulated antenna pa-rameters are tabulated in Table III. The proposed antenna has

relatively better radiation performance than the conventional

dipole antennas. In addition, the gain of proposed antenna is

higher than different kinds of miniaturized and narrow band an-

tennas in literature [23], [25], [28], [35].

IV. CONCLUSION

A broadband planar antenna is proposed by loading a narrow-

band dipole antenna with six LHM unit cells. The antenna pos-

sesses broad impedance bandwidth of 63% overthe band of 1.3–2.5 GHz.

8/13/2019 E5BYTQ3



Fig. 9. (a) Directivity, (b) efficiency, and (c) gain versus frequency.

The radiation of electrically small LHM unit cells is also

demonstrated, which is a quite important concept for the antenna

miniaturization. The overall size of the antenna is smaller than

at the center operation frequency. It has superior radiationperformance in comparison to the conventional dipole antennas.

The radiation pattern suggests the use of the antenna in compact,

broadband, wireless applications. It is obvious that the concept

of lefthandedness shows new opportunities, and better antenna

designs can be obtained from the development of left-handed

antennas than in normal right handed structures.

REFERENCES

[1] L. J. Chu, “Physical limitations on omnidirectional antennas,” J. Appl.

Phys., vol. 19, pp. 1163–1175, Dec. 1948.[2] K. F. Lee, K. M. Luk, K. F. Tong, S. M. Shum, T. Huynh, and R. Q.

Lee, “Experimental and simulation studies of the coaxially fed U-slot

rectangular patch antenna,” Proc. Inst. Elect. Eng. Microw. AntennasPropag., vol. 144, no. 5, pp. 354–358, Oct. 1997.

[3] K. L. Lau, K. C. Kong, and K. M. Luk, “A miniature folded shortedpatch antenna for dual-band operation,” IEEE Trans. Antennas

Propag., vol. 55, no. 8, pp. 2391–2398, Aug. 2007.[4] K.-F. Tong and T.-P. Wong, “Circularly polarized U-slot antenna,”

IEEE Trans. Antenn as Propag., vol. 55, no. 8, pp. 2383–2385, Aug.2007.

[5] B.-L. Ooi, S. Qin, and M.-S. Leong, “Novel design of broad-bandstacked patch antenna,” IEEE Trans. Antennas Propag., vol. 50, no.

10, pp. 1391–1395, Oct. 2002.[6] C. L. Lee, B. L. Ooi, and X. D. Zhou, “A broadband air filled stacked

U-slot patch antenna,” Elect. Lett., vol. 35, no. 7, pp. 515–517, Apr.1999.

[7] M. A. Matin, B. S. Sharif, and C. C. Tsimenidis, “Probe fed stackedpatchantenna for widebandapplications,” IEEE Antennas Propag., vol.55, no. 8, pp. 2385–2388, Aug. 2007.

[8] K. Sarabandi, R. Azadegan, H. Mosallaei, and J. Harvey, “Antennaminiaturization techniques for applications in compact wireless trans-ceivers,” in Proc. URSI , 2002, pp. 2037–2040.

[9] Metamaterials Physics and Engineering Explorations, N. Engheta andR. W. Ziolkowski, Eds. New York: Wiley/IEEE, 2006.

[10] V. G. Veselago, “The electrodynamics of substance with simultane-ously negative values of and ,” Sov. Phys.-Usp., vol. 10, no. 4, pp.509–514, Jan.–Feb. 1968.

[11] D. R. Smith, W. J. Padilla, D. C. Vier, S. C. Nemat-Nasser, and

S. Schultz, “Composite medium with simultaneously negative per-meability and permittivity,” Phys. Rev. Lett., vol. 84, no. 18, pp.4184–4187, 2000.

[12] C. Caloz and T. Itoh , Electromagnetic Metamaterials: Transmission

Line Theory and Microwave Applications. Piscataway, NJ: Wiley-IEEE, 2005.

[13] J. D. Baena, J. Bonache, F. Martín, R. Marqués, F. Falcone, T.Lopetegi, M. A. G. Laso, J. García-García, I. Gil, M. F. Portillo,and M. Sorolla, “Equivalent-circuit models for split-ring resonatorsand complementary split-ring resonators coupled to planar transmis-sion lines,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 4, pp.1451–1461, Apr. 2005.

[14] J. Baena, R. Marqués, and F. Medina, “Artificial magnetic metama-terial design by using spiral resonators,” Phy. Rev. B, vol. 69, pp.0144021–0144025, 2004.

[15] H. Chen, J. Zhang, Y. Bai, Y. Luo, L. Ran, Q. Jiang, and J. A. Kong,“Experimental retrieval of the effective parameters of metamaterialsbased on a waveguide method,” Opt. Express, vol. 14, no. 26, pp.12944–12949, Dec. 2006.

[16] D. R. Smith, D. C. Vier, T. Koschny, and C. M. Soukoulis, “Elec-tromagnetic parameter retrieval from inhomogeneous metamaterials,”Phys. Rev. E , vol. 71, pp. 0366171–03661711, 2005.

[17] N. G. Alexopoulos, C. A. Kyriazidou, and H. F. Contopanagos, “Effec-tive parameters for metamorphic materials and metamaterials througha resonant inverse scattering approach,” IEEE Trans. Microw. Theory

Tech., vol. 55, no. 2, pp. 254–267, Feb. 2007.[18] X. Chen, T. M. Grzegorczyk, B.-I. Wu, J. Pacheco, Jr., and J. A. Kong,

“Robust method to retrieve the constitutive effective parameters of metamaterials,” Phys. Rev. Lett. E , vol. 70, pp. 0166081–0166087,2004.

[19] D. R. Smith, D. C. Vier, N. Kroll, and S. Schultz, “Direct calculation

of permeability and permittivity for a left- handed metamaterial,” App.Phys. Lett., vol. 77, no. 14, pp. 2246–2248, Oct. 2000.

[20] G. V. Eleftheriades and K. G. Balmain , Negative Refraction Metama-

terials: Fundamental Principles and Applications. New York: WileyInterscience, 2005.

[21] H. V. Nguyen and C. Caloz, “Generalized coupled-mode approach of metamaterial coupled line couplers: Coupling theory, phenomenolog-ical explanation and experimental demonstration,” IEEE Trans. Mi-

crow. Theory Tech., vol. 55, no. 5, pp. 1029–1039, May 2007.[22] M. Gil, J. Bonache, J. Garcia-Garcia, J. Martel, and F. Martin, “Com-

posite right/left-handed metamaterial transmission linesbased on com-plementary split-rings resonators and their applications to very wide-band and compact filter design,” IEEE Trans. Microw. Theory Tech.,vol. 55, no. 6, pp. 1296–1304, Jun. 2007.

[23] C.-J. Lee, K. M.K. H. Leong,andT. Itoh,“Compositeright/left-handedtransmission line based compact resonant antennas for RF module inte-

gration,” IEEE Trans. Antennas Propag., vol. 54,no. 8, pp.2283–2291,Aug. 2006.

8/13/2019 E5BYTQ3



[24] M. Schüßler, J. Freese, and R. Jakoby, “Design of compact planar an-tennas using LH- transmission lines,” in Proc. IEEE MTT-S Int. Mi-

crow. Symp., 2004, vol. 1, pp. 209–212.[25] C.-J. Lee, K. M. K. H. Leong, and T. Itoh, “Design of resonant small

antenna using composite right/left handed transmission line,” in An-

tennas Propag. Soc. Int. Symp., 2005, vol. 2B.[26] A. Sanada, M. Kimura, I. Awai, C. Caloz, and T. Itoh, “A planar ze-

roth-order resonator antenna using a left handed transmission line,”

in 34th Eur. Microw. Conf., Amsterdam, The Netherlands, 2004, pp.1341–1344.

[27] C. A. Allen, K. M. K. H. Leong, C. Caloz, and T. Itoh, “A two-dimen-sional edge excited metamaterial based leaky wave antenna,” in Proc.

Antennas Propag. Soc. Int. Symp., 2005, vol. 2B, pp. 320–323.[28] H. Iizuka and P. S. Hall, “Left-handed dipole antennas and their

implementations,” IEEE Trans. Antennas Propag., vol. 55, no. 5, pp.1246–1253, May 2007.

[29] T. Koschny, P. Markos, D. R. Smith, and C. M. Soukoulis, “Resonantand antiresonant frequency dependence of the effective parameters of metamaterials,” Phys. Rev. E , pp. 0656021–0656024, 2003.

[30] L. Li, H. Yao, Q. Wu, and Z. Chen, “Broad-bandwidth and low-lossmetamaterials: Theory, design and realization,” J. Zhejiang Univ. Sci-

ence A, vol. 7, no. 1, pp. 5–23, Jan. 2006.[31] W. Geyi, P. Jarmuszewski, and Y. Qi, “The Foster reactance theorem

for antenna radiation Q,” IEEE Trans. Antennas Propag., vol. 48, no.

3, pp. 401–408, Mar. 2000.[32] K. Y. Yazdandoost and R. Kohno, “Ultra-wideband L-loop antenna,”

in Proc. IEEE Int. Conf. on Ultrawideband , 2005, pp. 201–205.[33] R. C. Johnson , Antenna Engineering Handbook , 3rd ed. New York:

McGraw-Hill.[34] T. Jiang, Y. Yuan, D. Wang, L. Ran, and J. A. Kong, “High directive

cavity antenna based on 1D LHM-RHM resonator,” in PIERS , 2007,vol. 3, no. 3, pp. 2054–2056.

[35] A. K. Skrivervik, J. -F. Zürcher, O. Staub, and J. R. Mosig, “PCSantenna design: The challenge of miniaturization,” IEEE Antennas

Propag. Mag., vol. 43, no. 4, pp. 12–27, Aug. 2001.

Merih Palandoken received the B.S. degree inelectrical engineering (with honors) from CukurovaUniversity, Turkey, in 2002, the M.S. degree inmicroelectronics and microsystems from the Tech-

nical University of Hamburg, Hamburg, Germany,in 2005, and is currently working toward the Ph.D.degree at the Technical University of Berlin, Berlin,Germany.

He has worked in the Power Electronics ResearchGroup, Cukurova University, of Prof. Dr. M. Tumayin the field of active power filter design for one year.

Since 2006, he hasbeenworkingat theTechnicalUniversityof Berlinin thefieldof metamaterial based antennas and microwave filters in the research group of Prof. Dr.-Ing. H. Henke.

Andre Grede was born in 1979. He received the de-gree in electrical engineering in 2006 and is currentlyworking toward the Ph.D. at the Technical Universityof Berlin (TU-Berlin), Berlin, Germany, under Prof.Dr.-Ing. H. Henke.

From 1999 to 2006, he was a Visiting Scholarat TU-Berlin. From 2001 to 2006, he was with theFraunhofer Heinrich-Hertz-Institut where he was

responsible for the development of RF-componentsin the Optical Time Division Multiplexing Groupof Dr. Weber. Since 2006, he has been working in

the field of electron vacuum tubes with the aim to find new ways to generatemedium and high power microwaves. Parallel to this work, he is involved inprojects like short range radar and wireless energy transfer.

Heino Henke received the Dr.-Ing. degree in elec-trical engineering from the Technische HochschuleDarmstadt, Darmstadt, Germany, in 1976.

He has worked in bio-cybernetics at the Universityof Frankfurt, Frankfurt, Germany, and for 13 yearsin the European Laboratory for High Energy Physics(CERN), Geneva, Switzerland, where his activitywas in accelerator physics, radio-frequency equip-ment and electromagnetic interferences. In 1990,he became a Professor in electromagnetics at theTechnical University of Berlin, Berlin, Germany. His

research is focused on electromagnetic fields and waves, particle acceleratingstructures, passive microwave components, metamaterials and millimeter-wavetubes. He has published two textbooks and in excess of 200 papers.

E5BYTQ3

Documents

Transcript of E5BYTQ3