Design of RF MEMS Phase S hifter using Capacitive Shunt ...Design of RF MEMS Phase S hifter using...

Design of RF MEMS Phase Shifter using

Capacitive Shunt Switch 1B. Nataraj,

2K.R. Prabha,

3S. Surya Sri,

4G. Suguna and

5K.A. Swathi

Associate Professor, 1Department of Electronics and Communication Engineering,

Sri Ramakrishna Engineering College,

Coimbatore, Tamil Nadu, India.

Assistant Professor 2Department of Electronics and Communication Engineering,


Coimbatore, Tamil Nadu, India. 3Department of Electronics and Communication Engineering,






Coimbatore, Tamil Nadu, India.

Abstract This paper presents the design and analysis of RF MEMS phase shifter

using capacitive shunt switches for broadband applications in microwave

and millimeter wave devices. The equivalent circuit of the phase shifter

have been examined with the capacitance of MEMS switches in both up

and down states in bilateral inter-digital coplanar waveguide. A control

voltage is applied between the center conductor and the switch’s upper

surface which actuates it and pulls down the surface, which creates a slow-

wave transmission line. The loading capacitance is summed up with the

International Journal of Pure and Applied MathematicsVolume 119 No. 10 2018, 1053-1066ISSN: 1311-8080 (printed version); ISSN: 1314-3395 (on-line version)url: http://www.ijpam.euSpecial Issue ijpam.eu

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line capacitance, thus varying the transmission line’s characteristic

impedance. The phase velocity of the signal is varied by this change in

impedance which produces a phase shift. In order to overcome the defects

of conventional waveguide, tapered coplanar waveguide is used and this

results in an increase in phase shift per unit length with a small decrease in

insertion loss. By further design implementation of the taper sections, the

losses can be reduced to a large extent.

Key Words:MMIC, MEMS, CPW, DMTL, quasi-TEM, phase shifter,

switches.

International Journal of Pure and Applied Mathematics Special Issue

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1. Introduction

In modern communication technology, antennas play a crucial role. It’s size is

the most demanding factor and therefore, it is to be integrated with common

very-large-scale integration (VLSI) technology. Phase shifter is an important

device in communication system. It should be designed in such a way that it

reduces electromagnetic interference and size should be small to consume less

power. In order to overcome the defects of conventional electronics beam

steering (Garver, 1972), Phase shifters are widely used in phased array

antennas. Phase Shifters can either be analog or digital. Continuous phase shift

is provided by analog phase shifter whereas digital phase shifter gives discrete

phase shift. Analog phase shifter has a very low insertion loss compared to

digital phase shifter. For low-cost microwave applications, many monolithic

microwave integrated circuit (MMIC) phase shifters are developed.

In recent years, RF Micro electromechanical Systems (MEMS) devices have

undergone enormous development and it provides many solutions for novel

components and system implementation. MEM is truly an enabling technology

allowing the development of smart products by augmenting the computational

ability of microelectronics with the perception and control capabilities of micro

sensors and micro actuators. The three characteristic features of MEMS

fabrication technologies are miniaturization, multiplicity, and microelectronics.

This technology has gained potential in defence and commercial

communication systems over a wide range of frequency. The development of

radio frequency Micro Electro Mechanical Systems (RF MEMS) technology

lead to miniaturization, low power consumption applications, low insertion loss

and wide bandwidth operation features at high frequency. It has most promising

role in applications like reconfigurable components such as switches, filters,

varactor diodes and phase shifters with low losses, low power consumption,

lesser inter-modulation products and high linearity are achievable using this

technology. To overcome huge size and losses that conventional phase shifters

exhibit, RFMEMS phase shifters are used in phased-array radar applications.

Phased array antennas are actually an electronically scanned array, from which

the beam of radio waves are projected in different directions without moving the

antennas. Feed network (Transmitter), phase shifters and antennas compute a

typical phased-array antenna. Hybrid topology of these components increases

the network’s size and it results in parasitic capacitive effects, package costs

and increased losses. These complications can be prevented by putting these

individual components on a single substrate, producing monolithic phased

arrays, possible through MEMS technology. Design analysis of conventional

coplanar waveguide (CPW) and tapered CPW for the use in phase shifter design

using RF MEMS technology is discussed in this paper. In this study, phase

shifter is designed to operate at 10 GHz and employs analog distributed

transmission line phase shifters. The phase shifters presented in this study are

used to obtain maximum phase shift with minimum wavelength using various


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types of CPW circuits.

2. Coplanar Waveguide

Nowadays, High power electronic applications can be operated at frequencies of

100 GHz and beyond the frequency ranges. At such frequencies, for device are

connected and signal is distributed through transmission lines. Comparing

various transmission lines, coplanar waveguides are preferred due to its

structure in which all the conductors supporting wave propagation are located

on the same plane. CPWs play a vital role for signal characterization and

MMICs. The high impedance CPW transmission line and its equivalent circuit

are shown in Figure. 1 a and b.

The first analytic formulas were proposed by Wen (1969) for the calculation of

quasi-static wave parameters of CPW’s using conformal mapping, based on the

thickness of the substrate and the ground wires of the CPW’s are infinitely

extensive. A mathematical study by Veyres and Fouad Hanna (1969) expanded

the implementation of conformal mapping to CPW’s with definable dimensions

and thickness of the substrate. The analyzed results are accurate only if the

thickness of substrates are greater than the line dimensions. Ghione et al. (1984)

have discovered more widely applicable formulas that the phase velocities of

complementary lines are equal, using the duality principle.

Figure 1a): Layout of the CPW b): Equivalent Circuit of the CPW

There exist two types of coplanar lines: the first, called coplanar waveguide

(CPW) which has a center conductor strip and two ground conductor planes


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placed on the same side of a dielectric substrate (Quartz) whose widths can be

varied, as shown in Fig.1. The surface of the substrate is often in contact with

the center conductor, that is coated with metal, or metalized. Here the ground is

at the same side of the surface as the center conductor, therefore the inductance

coupled with accessing ground is remarkably reduced. The width and area of

the center conductor determines the characteristic impedance of the

transmission line.

The conformal mapping is applied to obtain the characteristics of transmission

lines using the assumption that the propagation mode in the CPW transmission

lines is quasi-static, i.e., it is a pure TEM mode. The effective dielectric

constant, velocity of the phase, and characteristic impedance of a CPW

transmission line are given as (Gupta, 1979):

eff =

=

Zo =

where CCPW is the line capacitance of the CPW transmission line, C0 is the line

capacitance of the transmission line when no dielectrics exist, and c is the speed

of light in free space. To obtain the quasi-static wave parameters of a

transmission line, the capacitances CCPW and C0 -is to be found.

The Veyres and Fouad Hanna (1969) approximation (superposition of partial

capacitances) is used, in which the line capacitance of the CPW is the sum of

two line capacitances, i.e,

=

= 4 o

where K is the elliptical integral of the 1st kind, and K’(k) = K(k’). The

variables k and k’ are given as

k =

=

C1= capacitance in which the electrical field exists only in a dielectric layer

with h1(thickness) and effective dielectric constant of -1.

= 2 o )

Where

=

=


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The absolute elliptical integrals of the 1st kind using the approximations given

by Hilberg (1969) is given as

≈ ln(2 ) for 1 ≤ ≤ ∞ and

≤ k ≤ 1

for 0 ≤ ≤ 1 and 0≤ k ≤

3. Phase Shifter Design

The circuit suggested here is based on a CPW distributed MEMS transmission

lines were phase velocity is varied by using a single control voltage and the

height is varied by the MEMS loading capacitors, and the capacitive load

distributed to the transmission line and its propagation characteristics, as shown

in Figure. 2a. This results in the analog control of the phase velocity and,

therefore a true time delay phase shifter. The design requires a small value of

loading capacitance per unit length, which results in very high actuation

voltage. The topology of the CPW transmission line presented here, which

varies the impedance, helps to increase the phase shift per unit length, resulting

in a reduced physical line length, reduced pull down voltage and high

capacitance ratio. The impedance and propagation velocity of the slow-wave

trans-mission line are determined by the size of the MEMS bridges and their

periodic spacing. The equivalent circuit of the loaded distributed MEMS

transmission line is shown in Figure 2b. The shunt capacitance associated with

the MEMS bridges is in parallel with the distributed capacitance of the

transmission line, shown in Figure 2b.

From the analysis of CPW using conformal mapping, the per unit length

capacitance is obtained. i.e. Ct=Ccpw. The unloaded lines per unit length

capacitance and inductance are given by (Barker, 1998)

= and =

where eeff is the effective dielectric constant of the unloaded CPW transmission

line, Z0 is the characteristics impedance of the unloaded CPW line, and c is the

free space velocity. The MEMS bridge only loads the transmission line with a

parallel capacitance Cb, the loaded line impedance Z1 and phase velocity Vl of

the loaded line, become

= and =

where s is the periodic spacing of the MEMS bridges and Cb/s is the distributed

MEMS capacitance on the loaded CPW line.

The MEMS bridge becomes unstable at 2g0/3, where g0 is the zero-bias bridge

height. The voltage at which this instability occurs is the “pull-down” voltage


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and is given by

= V

The relative phase between the two states or the net phase shift is found from

the change in the phase constant given by

= ( - )

The design consists of a 19365µm long CPW trans-mission line whose center

conductor width (W) is 100µm and the gap is 100µm is fabricated on a 100µm

silicon substrate with a dielectric constant of 3.8 and loss tangent=0.001 and

with 15 shunt MEMS bridge capacitors placed periodically over the

transmission line, shown in Figure 3. The height of the bridge above the center

conductor is computed for up state(3µm) and down states(1µm) respectively.

The effective dielectric constant ( eff) of the unloaded CPW line has an average

value of 6.25 and is linearly invariant with frequency. The width and span of the

MEMS bridges are 100µm and 100µm, respectively. The same parameters are

used for designing bilateral inter-digital CPW with wings and without wings.

Figure 2: a) Layout of the ConventionalCPW with MEMS Switch

b) Equivalent Circuit of the ConventionalCPW with MEMS Switch

Figure 3: Layout of Conventional CPW loaded with 11 MEMS bridges


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Results

(a)

(b)

(c)

(d)

Figure 4(a): Layout of bilateral inter-digital CPW Design-I (without wings) having 15

MEMS bridges. (b) (dB)in 1µm(DOWN) and 3µm(UP) states. (c) (dB)in

1µm(DOWN) and 3µm(UP) states.(d) (phase)in 1µm(DOWN) and 3µm(UP) states.

6 7 8 9 10 11 12 13 145 15

-60

-40

-20

-80

0

freq, GHz

dB(S

(1,1

))dB

(bic

pw1_

mom

_1_a

..S(1

,1))

6 7 8 9 10 11 12 13 145 15

-1.5

-1.0

-0.5

-2.0

0.0

freq, GHz

dB(S

(1,2

))dB

(bic

pw1_

mom

_1_a

..S(1

,2))

6 7 8 9 10 11 12 13 145 15

-100

0

100

-200

200

freq, GHz

phas

e(S

(1,2

))ph

ase(

bicp

w1_

mom

_1_a

..S(1

,2))


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(a)

(b)

(c)

(d)

Figure 5 (a): Layout of bilateral inter-digital CPW Design-II (with wings) having 15

MEMS bridges. (b) (dB) in 1µm(DOWN) and 3µm(UP) states. (c) (dB) in



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(a)

(b)

(c)

(d)

Figure 6: (a) Layout of tapered bilateral inter-digital CPW Design-II(without wings) having

15 MEMS bridges. (b) (dB) in 1µm(DOWN) and 3µm(UP) states. (c) (dB) in

1µm(DOWN) and 3µm(UP) states. (d) (phase) in 1µm(DOWN) and 3µm(UP) states.

6 7 8 9 10 11 12 13 145 15

-50

-40

-30

-20

-10

-60

0

freq, GHz

dB(S

(1,1

))dB

(_12

34b_

mom

_3_a

..S(1

,1))

6 7 8 9 10 11 12 13 145 15

-2.0

-1.5

-1.0

-0.5

-2.5

0.0

freq, GHz

dB

(S(1

,2))

dB

(_1

23

4b

_m

om

_3

_a

..S

(1,2

))

6 7 8 9 10 11 12 13 145 15

-100

0

100

-200

200

freq, GHz

phas

e(S

(1,2

))ph

ase(

_123

4b_m

om_3

_a..S

(1,2

))


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(a)

(b)

(c)

(d)

Figure 7: (a) Layout of tapered bilateral inter-digital CPW Design-II (with wings) having

15 MEMS bridges. (b) (dB) in 1µm(DOWN) and 3µm(UP) states. (c) (dB) in



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4. Conclusion

Design-II of tapered bilateral inter-digital coplanar waveguide produce more

phase shift among the three designs with and without wings. Phase shift

changes with respect to number of bridges and the gap between center

conductor and the bridge (bridge gap). So tapping the center conductor gives

rise to the generation of six capacitances in between the center conductor and

bridges, namely , , on the upper surface and , in the lower surface

and thus resulting in relatively low insertion loss. Therefore a maximum phase

shift is obtained with minimum wavelength. The future work is to increase the

isolation, reduce the insertion loss and mainly to increase the phase shift per

unit length.

References

[1] Jacopo Iannaci, RF MEMS for high performance and widely reconfigurable Passive components- A Review with focus on future telecommunications, IoT and 5G applications, Journal of King Saud University-Science, centre for materials and Microsystems (2015).

[2] Laxma Reddya B., Shanmuganantham T., Design of Novel Capacitive RF MEMS Shunt Switch with Aluminum Nitride (AlN) Dielectric, 3rd International Conference on Materials Processing and Characterization (2014).

[3] Che-Heung Kim, Mechanically coupled low-voltage electrostatic resistive RF multi throw switch, IEEE Transactions on Industrial Electronics 59(2) (2012).

[4] MonFernandez-Bolanos Badia, Elizabeth Buitrago, Adrian Mihai Ionesco, RF MEMS Shunt capacitive switches using AIN, IEEE Journal of MEMS 21(5) (2012).

[5] Xiaobin Yuan, Zhen Peng, James, Hwang C.M., David Forehand, Cahrles L. Goldsmith, Acceleration of Dielectric charging in RFMEMS capacitive switches, IEEE Trans. On Device and Materials Reliability 6(4) (2013).

[6] Rainee N. Simon, Coplanar Waveguide Circuits components and systems, IEEE Transaction on Microwave Theory and Techniques 44(245) (2007).

[7] Sterner M., Roxhed N., Stemme G., Oberhammer J., Static zero –power –consumption coplanar waveguide embedded DC-to –RF metal contact MEMS switches in 2 –port and 3-port configuration, IEEE Trans. Electron devices 57(7) (2010), 1659-1669.


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[8] Kevin D. Leedy, Richard E. Strawser, Thin-Film Encapsulated RFMEMS switches, IEEE Journal of MEMS 16(2) (2007).

[9] Gabriel M. Rebeiz, RF MEMS Theory, Design and Technology, 1st Ed., Wiley & Sons Inc., (2007).

[10] Jin Yalin, Nguyen Cam, Ultra-Compact High –Linearity High-Power fully integrated DC -20 GHz 0.18 micrometer CMOS T/R, IEEE, Transactions on microwave theory and techniques 55 (2007), 30-36.


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