AC Current Monitor by LM358

7/21/2019 AC Current Monitor by LM358

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AC Current Monitor

Senses high current-flow into power cables No wire-cutting, three versions available

Circuit diagram:

Parts:R1,R2,R8____________1K 1/4W ResistorsR3,R4_____________220K 1/4W ResistorsR5________________100R 1/4W Resistor (See Notes)R6_________________10K 1/2W Trimmer CermetR7,R10______________1M 1/4W ResistorsR9_________________22K 1/2W ResistorR11 to R17__________1K 1/4W Resistors

C1,C3_____________100F 25V Electrolytic CapacitorsC2,C4_______________1F 63V Electrolytic Capacitors

D1________________5mm. Red LEDD3,D4___________1N4002 100V 1A DiodesD2,D5,D6,D7_______LEDs (Any color and size)

Q1_______________BC327 45V 800mA PNP Transistor

IC1______________TL061 Low current BIFET Op-Amp (First version)IC1______________LM358 Low Power Dual Op-amp (Second version)IC1______________LM324 Low Power Quad Op-amp (Third version)

L1________________10mH miniature Inductor (See Notes)

RL1______________Relay with SPDT 2A @ 220V switchCoil Voltage 12V. Coil resistance 200-300 Ohm

J1_______________Two ways output socket

Device purpose:

This circuit was designed on request, to remotely monitor when a couple of electric heaters have been left on. Its sensor must be placed in contact with the feeder to be

able to monitor when the power cable is drawing current, thus causing the circuit to switch-on a LED.

The circuit and its sensor coil can be placed very far from the actual load, provided an easy access to the power cable is available.

Any type of high-current load or group of loads can be monitored, e.g. heaters, motors, washing machines, dish-washers, electric ovens etc., provided they dissipate a

power comprised at least in the 0.5 - 1KW range.

This design features three versions. The basic one illuminates a LED when the load is on. The second version activates a Rela y when a pre-set current value flows into


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While there are a plethora of similar 2-transistor schematics available, this one is above average. A major improvement over the little 1-transistor circuit that weve

discussed previously, this transmitter consist of two stages. The first transistor is used to amplify audio, which means that the microphone is now much more sensitive

to sound. The second one acts as an oscillator.

All in all, this is an interesting project for beginners and more experienced hobbyists alike. Use any stiff wire or te lescope whip antenna, just make sure that its not too

long. Range should be about 100m or even more in the open. Stability is still a problem, though.

PartR1 270

R2, R5, R6 4k7

R3 10k

R4 100kC1 1n

C2 5.6p

C3, C4 10u

C5 3-18p

L1, L2 5 turns of enamel coated magnet wire with an inside diameter of about 4mm

Mic Electret microphone

Q1, Q2 2N2222, 2N3904 or any other general-purpose NPN

Dew sensor by LM358

Dew (condensed moisture) ad- versely affects the normal performance of sensitive electronic devices. A low-cost circuit described here can be used to switch off any

gadget automatically in case of excessive humidity. At the heart of the circuit is an inexpensive (resistor type) dew sensor element.

Solar panel voltage regulator by LM358

For regulating a solar panels output, there are several possible ways. A linear series regulator can be used, but has the disadvantage of causing some voltage drop and

having some internal power consumption at times when the sun is weak and the load is heavy. Its much better to use a shunt regulator, which is inactive at such times,

and springs to life only when there is excess energy. For this reason, most solar panel regulators use the shunt scheme, the one presented here being no exception.

D1 can be any diode that can safely survive the panels current. If the panel has a very low voltage output (less than 33 cells in series), it is an advantage to employ a

Schottky diode in this place.
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Q1 and Q2 are common power Darlington transistors. They need to be heatsinked for safe long- term operation at the 12 Watt dissipation level. Thats easy enough to

do, but many newcomers misjudge how much thermal resistance is introduced by a mica insulator! Plan on 1K/W thermal resistance inside each transistor, two times as

much in the insulator (if you use any), and 370K safe junction temperature. For typical environmental conditions, this makes you need a heatsink having a thermal

resistance of about 1.3K/W. If it is larger, you get more safety margin.

R1 and R2 will have to be made by combining a number of power resistors in parallel. Yes, you need to make two resistor arrays of 4 Ohm, 80W each! This 80W figure

includes a reasonable safety margin. These resistors will produce a lot of heat, and you may cook your coffee on them! Be sure to mount them in such a way that they

have lots of ventilation, and that the heat from them will not reach the other components.

R3 and R4 may to have be built from parallel combinations too, because of the low value of only 0.15 Ohm.

U2 is a voltage reference IC. You cannot replace it by a standard Zener diode! Zeners are much too unstable! If you cant find this chip locally, you may use the

ubiquitous 7805 regulator instead, but the power drain from the battery will be higher. In this case, of course you dont need R8, but you would need a 1uF capacitor at

the 7805 output.

Q3 is a power MOSFET that has a very low Rds(on). You may use a different one, provided that it has a resistance thats low e nough for your application. You may use

several in parallel. The one I used has low loss even at loads of 20A, and can handle much more!

ECM microphone preamplifier

Notes:Both transistors are low noise types. In the original circuit, I used BC650C which is an ultra low noise device. These are hard to find so I now useBC109C. The circuit is very device tolerant and will set its quiescent point at roughly half the supply voltage at the emitter of the last transistor.

The ECM is a mic insert, designed to operate at around 2 - 10 Volts DC. The 1k resistor limits the current to the mic. The output impedance is verylow and well suited to driving cables over distances up to 50 meters. Screened cable therefore is not necessary.

The frequency response measured across the 10k load resistor is shown below:


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The noise response of the amplifier measured across the 10k load is shown below. Please note that was measured with the mic insert replaced by a

signal generator.

This circuit has an excellant overload margin, and can cope with anything from a whisper to a loud shout, however the amplifier youconnect this too can be overloaded, so care should be taken.

HI-FI AF Preamplifier

Notes:This circuit has an exceptionally good high frequency response, as demonstrated by applying an 100kHz squarewave to the input. I have produced

some response graphs using Tina Pro to highlight these characteristics.


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The Preamp's Frequency and Phase Response

Squarewave Response with 100kHz Input Applied

Total Noise at Output Measured with 600R Load

Signal to Noise Ratio at Output


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Battery Monitor

The schematic

Click here forPIC software andthehex file can be found here.

Finesse Voltage Regulator Noise!

System designers often find themselves battling power supply hum, noise, transients, and various perturbations wreaking havoc with low noise

amplifiers, oscillators, and other sensitive devices. Many voltage regulators have excessive levels of output noise including voltage spikes from switchingcircuits and high flicker noise levels from unfiltered references. Ordinary three-terminal regulators will have several hundred nanovolts per root-hertz ofwhite noise and some reference devices exceed one microvolt per root-hertz. DC to DC converters and switching regulators may have switchingproducts ranging into the millivolt range covering a wide frequency spectrum. And many systems have offending devices that "d irty up" otherwise clean

supply rails.
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The traditional approach to reducing such noise products to acceptable levels could be called the "brute force" approach - a large-value inductorcombined with a capacitor or a clean-up regulator inserted between the noisy regulator and load. In either case, the clean-up circuit is handling theentire load current in order to "get at" the noise. The approach described in this paper uses a bit of finesse to remove the undesired noise withoutdirectly handling the supply's high current.

The key to understanding the "finesse" approach is to realize that the noise voltage is many orders of magnitude below the regulated voltage, evenwhen integrated over a fairly wide bandwidth. For example, a 10 volt regulator might exhibit 10 uV of noise in a 10 kHz bandwidth - six orders ofmagnitude below 10 volts. Naturally, the noise current that flows in a resistive load due to this noise voltage is also six o rders of magnitude below the

DC. By adding a tiny resistor, R, in series with the output of the regulator and assuming that a circuit somehow manages to reduce the noise voltage atthe load to zero, the noise current from the regulator may be calculated as Vn/R. If the resistor is 1 ohm then, in this example, the noise current will be10uV/1ohm = 10uA - a very tiny current! If a current-sink can be designed to sink this amount of AC noise current to ground at the load, no noise

current will flow in the load. By amplifying the noise with an inverting transconductance amplifier with the right amount of gain, the required currentsink may be realized. The required transconductance is simply -1/R where R is the tiny series resistor.

Consider the low power version shown in fig. 1 which might be suitable for cleaning up the supply to a low current device. A 15 ohm resistor is insertedin series with the regulator's output giving a 150 millivolt drop when the load draws 10 mA - typical for a low-noise preamplifier or oscillator circuit. The

single transistor amplifier has an emitter resistor which combines with the emitter diode's resistance to give a value near 15 ohms. The regulator's noisevoltage appears across this resistor so the noise current is shunted to ground through the transistor's collector. The noise reduction can be over 20dBwithout trimming the resistor values and the intrinsic noise of the 2N4401 is only about 1 nanovolt per root-hertz. Trimming the emitter resistor canachieve noise reduction greater than 40 dB.

For higher current loads it is desirable to have a much lower series resistor. For such applications more gain is needed and one approach is to replace

the single transistor in fig. 1 with a compound transistor as shown in fig. 2. The effective emitter resistance is on the order of 0.25 ohms so an emitterresistor near 0.75 ohms is needed for a series-pass 1 ohm resistor. The circuit is biased with a bit more current by the 470 ohm resistor and it canhandle 10mV glitches of either polarity. The darlington may be replaced with a 2N4403 but the effective emitter resistance wi ll be slightly above 1 ohm.

The simplicity of the one-transistor circuit is attractive and it is interesting to investigate the possibility of using this circuit for higher currents. Onelimiting factor is the intrinsic resistance of the emitter which limits the gain of the single st age. Choose a large die device or a device rated for high

collector current. A power transistor is a good choice even though the power dissipation will be low. The emitter resistor in figure 1 is set to zero andthe bias resistor is reduced to about 5 or 10k. The collector resistor is selected to achieve the desired gain: as this resistor drops in value, the emitterresistance drops by about .025 / Ic, not including the intrinsic resistance. A 2N5192 with a 270 ohm collector resistor and a 4.7k bias resistor wi ll work

well with a 1 ohm current sense resistor and will consume about 40mA. The transistor gain is obviously sensitive to temperature with no emitterdegeneration but good noise reduction will be maintained over a wide temperature range.

The experimentally inclined may wish to try a TL431 shunt regulator in place of the single transistor. The flicker noise will be a bit h igh but the circuitcould be useful for eliminating switching regulator spikes. The high gain of the TL431 should allow the use of very low series resistance. Another

interesting device is the CA3094 which has a built-in darlington transistor capable of handling up to 100 mA and the op-amp noise is a respectable 18nV at 10Hz.


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These two circuits are representative of many possible versions using the same basic technique. A three-transistor version has been constructed for usewith a 0.05 ohm resistor and a couple of op-amp versions have been constructed with the LM833. Although these versions work quite well thecomplexity begins to rival low noise voltage regulators. One advantage, however, is that no high-current pass element is needed so the circuit can bequite small.

The following circuit is designed for filtering 15 volt supplies like those typically found in instrumentation. The shunt will greatly reduce white noise,

spurious signals, and line-related signals on the power supply; the attenuation can exceed 40 dB with careful construction. The values are not criticalexcept that the gain of the amplifier should be very near the ratio of the transistor emitter resistor to the series shunt resistor. In this case the gain is

15/0.05 = 300. Actually the gain is 301 with the indicated values so a 299k resistor would be theoretically better but the resistor tolerances and theactual resistance in the 0.05 ohm shunt path will cause more variation. One of the gain resistors may be made variable to allow the performance to betweaked for the deepest null, if desired. Choose a low noise metal foil or wirewound potentiometer for best results. Standard fixed values will giveexcellent noise reduction sufficient for most applications. The LM833 is an excellent choice but many other low noise op-amps will work well. Choose anop-amp with a high bandwidth and low input noise voltage. A higher value shunt resistor may be used if the voltage drop can be tolerated; adjust the

gain of the amplifier to match as described above. The LM833 is a dual op-amp so two shunts may be implemented with the one package for filteringtwo different supplies or for cascading two shunts for additional line rejection and noise reduction. The noise shunt provides no load rejection beyondthe rejection provided by the source regulator through the 0.05 ohm resistor.

The following graph shows the performance of the noise shunt when powered by a three-terminal regulator. The regulator's noise is 330 nV per root-hertz at 100 hertz and the circuit reduces this noise to 20 nV. This 24 dB reduction is achieved without any selected values and with no particular

attention paid to the layout. The one potential problem area is the grounding; heavy ground traces or even ground plane are recommended. Theultimate rejection of the circuit is better than the apparent rejection in the plot; low frequency performance is impacted by the size of the couplingcapacitors and the noise floor is limited by the performance of the LM833 and resistor noise.

13.8 V / 15 A from a PC Power Supply


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Safety Instructions

Caution mortal danger: The following circuit operates at a mains voltage of 230 Vac. Because of rectification some of the components conduct dcvoltage of more than 322 V. Work has to be carried out only if the circuit is disconnected from the mains and de-energized. Note that capacitors locatedto the primary side can be charged with high voltage for several seconds even after switching of the mains voltage.

The major disadvantages of usual linear power supplies are high power dissipation, the size and the appropriated weight. When looking for analternative solution, I decided to use a switch mode power supply (SMPS). The efficiency of such power supplies is around 70 % to 90 % at a power

density of 0.2 W / cm. Because homebrewing was out of the question due to lack of time, I tried the modification of a PC swi tch mode power supply.

Fig.1: Block diagram of a primary switching power supply

Brief description of PC SMPS Features

Depending on the PC model, these are rated anywhere between 150 and 240 W. For supplying socket 7 main boards they have four different outputvoltages of +5 V, +12 V, -12 V and -5 V. They are mainly primary switching power supplies with power switches arranged in a half-bridge configuration.

The outputs can drive the usual 20 A (+5 V), 8 A (+12 V) and 0,5 A (-12 V, -5 V). At approx. 205 W output power and a typical efficiency of 75 % thismeans a dissipation of only 68 W. I had acquired an unbranded PC power supply, measuring 140 x 100 x 50 mm (W, D, H) and weighing 350 g. Most

power supply units are designed according to the same principle (half-bridge configuration) and hence the following described modification should beapplicable also to power supplies from other producers.

Fig.2: Half-bridge configuration of power switches

Regulation

After switching on the mains voltage the circuit operates for a short duration as a free-running oscillator. This behavior is caused by a feedback windingat the output transformer T2. As soon as the auxiliary voltage Uaux is present the pulse width modulator IC TL494CN from Texas-Instruments takes

over the control function and synchronizes the "oscillator".The error amplifier in the TL494 compares the voltage at the +5 V output (actual value) with a reference voltage (set value), calculates the analoguecontrol variable according to the PI algorithm and adjusts the pulse width modulator (see Fig. 6). The modulator sends alternate pulses to the drivertransistors Q5 and Q6. The pulse duration is reverse proportional to the control variable rating. Increasing loading on the +5 V output makes for wider

pulses, lighter loading causes narrower pulses. As there is a finite minimum pulse width, a minimum load of 0.1 A is required . Without this load the


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power supply may be destroyed. The switching frequency is approx. 33 kHz as usual for PC power supplies. It is defined by a resistor and a capacitorlocated at pin 5 and 6 of IC1.

Fig. 3: Primary side mains f ilter, rectifier, power switches and drivers

Monitoring Circuit

Several protection circuits are included in the original power supply. Excessive primary current due to a very high secondary current leads to a highalternating voltage at the T3 output. If this voltage is above a fixed threshold the TL494 stops immediately generating cyclically pulses and changes tothe intermitted mode (on / off). The circuit and the load are protected likewise against over-voltage at the +5 V output or short-circuit at the -12 V and

-5 V outputs. Switching off is executed via H-signal to the IC1 protection input (pin 4) too.

If you see a KA7500 or IR3MO2 PWM regulator IC on the board, each one is a pin compatible second source to the TL494CN. IC3 is a dual comparatorfrom LM339 type. Some power supplies are not equipped with this IC, but with a two transistor discrete monitoring circuit, offering the samefunctionality.

Mods to the Secondary Rectification

The intent is for all of the available power at the 12 V secondary of T1 to be rectified, regulated, protected and filtered to provide a single output of13.8 V DC at 205 W, or more if possible. A first check indicates that the +12 V wire was of the same diameter as the +5 V wire.First unsolder and remove all components on the secondary side of T1 which are provided for rectification, filtering and regulation of the four outputvoltages. On that part of the board are only remaining three RC members RC1 to RC3 and the components for providing the auxiliary power supplyUaux.

Fig.4: Secondary rectification as found in the original PC power supply

Reconstruction of the secondary side.


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Break the PCB tracks between the RC members RC1 / RC2 and both 5 V taps of the T1 secondary winding.

Modify L4 for 12 V at 20 A. Remove windings L4a, L4b and L4c from the toroid (counting turns of L4c). Rewind

the toroid L4* with a single winding, turn count as old L4c but with 2.5 times the thickness. Take two wires with1 mm diameter each, bifilar wounded.

Install two low ESR electrolytic capacitors of 2200 uF each and the 100 Ohm bleeder resistor as permanent load.

Use the old PCB tracks from the +5 V section and GND tracks as terminals for L4* , the 100 Ohm resistor and thetwo 2200 uF capacitors. Insert L4* at the same place onto the PCB component side where the L4b winding wasconnected before.

The original cooling of the rectifier diode D5 is insufficient. Adequate cooling is achieved by a finned heat sink

measuring 70 x 50 x 30 mm (W, D, H) instead of the old aluminium sheet metal.

Fasten D5 to the heat sink and extend the three leads by 40 mm long wires. Use isolation material and thermal

compound. D5 carries on some boards the abbreviation SKD.

Place the finned heat sink approx. 40 mm above the "stripped" secondary (see photo) with plastic spacers andlong M3 screws (avoid short-circuit to common).

Connect the anode leads of D5a and D5b with one RC member RC1 / RC2 each. The cathodes have to beconnected to the nodal point of RC1, RC2 and L4.

Establish two links between the 12 V terminals of T1 and the RC members by two thick wires. D5 will be fed fromthe 12 V winding.

A simple and clear structure of the secondary rectification was achieved after "stripping" and "reconstruction".

Fig. 5: New designed secondary for Ua = 13,8 V

Mods to the Regulation and Protection Circuit

The part of the circuit responsible for regulation and monitoring has to be modified at three places. Arrange additional components free standing ontothe component side of the PCB.

R24* is calculated for 13.8 V output voltage. The voltage at the (+) input of the error amplifier must be equal to2.5 V after control loop stabilization, i.e. half the 5 V reference voltage when the output is at 13.8 V.

R24* = 20 kOhm = 2 x 10 kOhm in series

Arrange a second universal diode 1N4148 and a 8,2 V Zener diode in series to D16.

Usum = 8,2 V + 2 x 0,7 V = 9,6 V

Simplify the voltage divider (R36, R42, R45 and D14) in the short-circuit protection circuit. For this remove R36

and D14. Connect the free end of R42 to common (GND) and replace R45 with one of higher value to ensure noshut-down at normal operation. The voltage across R42 must be less than 1,7 V (I chose 1,2 V).

R45* = 15 kOhm

The areas marked with dotted frames, show the modified or additional components that are necessary for 13.8 V output voltage.


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Fig. 6: Regulation and protection circuits incl. all modifications

Further Modifications

After commissioning the modified board, the situation regarding to interferences looks very bad. The whole reception range from 3,5 MHz to 30 MHzwas disturbed by harmonics of the 33 kHz switching frequency. S-meter readings showed S5 on 80 m down to S2 on 10 m. As I was testing the board

in a metal box, the HF radiation could only get out on the mains cable and/or DC output leads. The insertion of an additional standard 230 VAC mainsfilter and a home-brewed pi-fil ter in the output rendered the interference inaudible.

Insert an additional 230V / 2A mains filter to the primary side, close to the place where the mains cable entersthe enclosure rear wall.

Insert a 20 A pi-filter to the DC output , behind the +/- DC terminals at the rear wall.

The power supply enclosure must absolutely consist of iron sheet metal to screen magnetic fields. Aluminumplates protect only against electrical fields.

Optional on the primary: Replace the 220 uF smoothing capacitors C1 and C2 by 470 uF capacitors. This reduces

primary ripple, which helps output regulation at full load.

Testing the Power Supply

Phase 1: These tests have to be carried out at a low DC supply voltage in order to avoid component destruction in case of possible errors. The 13.8 Voutput is loaded with a 12 V / 50 W car headlight bulb and a 15 V / 1 A lab power supply is connected to GND and Uaux. The TL494 IC gets its

operating voltage and generates control pulses with maximum pulse duration. Check the signals at Q5 and Q6.

Phase 2: During the second test phase the galvanic isolated primary side of the circuit is supplied by the lab supply too. For this purpose make a shortcable link between Uaux and U+ as well as between GND and U-. The PWM controller tries to offer 13.8 V at the output at maximum pulse duration.

The later cannot be successful due to the low 15 Vdc input voltage and the present transformer ratio. With an oscilloscope measured signals at themeasuring points TP1 (emitter Q1 against emitter Q2) and TP2 (cathode D5 against GND) must look l ike as shown in figure 7.


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Fig. 7: Signal shape at TP1 and TP2

Phase 3: Nor disconnect the lab supply from the primary side only. Instead connect a 48 V / 1 A mains transformer to the L1 a nd N terminal in order tofeed the board with a galvanic isolated Ac voltage. 60 Vdc at C1 and C2 is in Europe defined as a non-dangerous voltage rate. 48 VAC at the input

causes a rise of the output voltage up to +6 V.

If everything is all right up to now, one can proceed with the exciting test at 230 Vac. The laboratory power supply, the 48 V transformer, the measuringinstruments and all provisional cable links attached for the test etc. must obviously be removed. The car bulb are further needed as a load and for thefunctional checks. If after applying of the 230 Vac mains voltage the lamps light up brightly, the output voltage amounts to 13.8 V and no undefined

noises or smells are noticeable one has won the first round. If a non recognizable error has passed the pre-testing the two switching transistors andcopper tracks say good-bye with a more or less loud bang.

For the following load test some high power resistors with resistance 1 Ohm and sufficent power rating are required. The current flowing wi th this loadshould not cause excessive heating of the rectifier diode and the switching transistors during a 5 minutes test periode.

Warning: Check temperature of components only if the mains voltage is switched off

Cooling of the switching transistors Q1 and Q 2 at a continuos current of 15 A has to be improved in any case. When exchanging the small heat sinks,note that they form an electrical connection between coper tracks on some boards. Replace the missing connection by wire links. As one can see on thephoto, I did not taken this measures for further power improvement.

Operation Experience

The modified board was permanently installed in the speaker cabinet SP120 that matches my transceiver. The mains lead exit fr om its back, which also

carries the DC terminals, an on-off switch, the additional mains filter and a small 12 V blower. A green LED power-on indicator was inserted in the frontpanel into a 5 mm hole drill. I had installed the small blower just in case, but found it superfluous; at the low duty cycle of CW and SSB, none of thecomponents is getting hot. The power supply has been used for several years and has given no problems.

Fig. 8: Modified power supply board in the SP120 speaker cabinet

AC Current Monitor by LM358

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