Abstract - NTNU...TET 4190 Mini-Project 2012 1 | P a g e Abstract In nowadays, the application of...

TET 4190 Mini-Project 2012

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Abstract

In nowadays, the application of DC-DC converter exponentially increases. Staring from

Mobile charger to Aerospace Power supply systems. We designed and simulate a DC-DC

buck converter to meet the specifications set by our professor and Wårtsilå. Essentially, our

buck converter is used to charge 300V battery by stepping down 400V DC voltage input

from micro grid. Our buck converter consists of a MOSFETs with diode, capacitors and

inductors. The MOSFET acts as a switch and is turned on and off by a Proportional Integral

(PI) Pulse Width Modulation (PWM) controller. This means that by controlling the duration

when the switch is on, the voltage or current at the output can be regulated.


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Contents

Abstract............................................................................................................................. 1 1. Introduction.................................................................................................................. 4 1.1 Background.................................................................................................... 4 1.2 Switching regulator........................................................................................ 4 2. Theory ........................................................................................................................... 5 2.1. Input DC voltage source.............................................................................. 5 2.2. Switching Devices....................................................................................... 5 2.3. Low Pass Filter (LPF)................................................................................. 5 2.4. Control of a DC - DC Converter................................................................. 6 3. Circuit Parameters and Loss Analysis .......................................................................... 8 3.1. Circuit parameters optimization.................................................................. 8 3.1.1.Inductance selection........................................................................... 8 3.1.2 Snubber capacitor optimization....................................................... 9 3.2 Switching Losses with Linear MOSFET Output Capacitance.................... 10 3.3 Power Losses and Efficiency of Buck Converter....................................... 13 4. Circuit Design and Simulation................................................................................. . 19 4.1 Main Converter Circuit design.................................................................... 19 4.2 MESFET Gate signal (PWM) .................................................................... 19 4.3 Ripple Inductor Current ............................................................................ 20 4.4 Ripple Output Voltage................................................................................ 21 5. Conclusion .................................................................................................................... 23 6. References...................................................................................................................... 23


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List of figures

Fig 1 Reversible power flow with reversible of the output current........ 6 Fig 2 Sample design for DC-DC buck converter................................... 13 Fig 3 Main DC- DC Converter circuit design............................................ 19 Fig 4 PWM Triggering circuit...................................................................... 19 Fig 5 Output Waveform of PWM .............................................................. 20 Fig 6 Ripple Inductor current waveform......................................................... 21 Fig 7 ripple voltage output.............................................................................. 22

1. Introduction


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1.1 Background

The switched mode dc-dc converters are some of the simplest power electronic

circuits which convert one level of electrical DC voltage into another level by

switching action. These converters have received an increasing deal of interest

in many areas. This is due to their wide applications like power supplies for

personal computers, office equipments, appliance control, telecommunication

equipments, DC motor drives, automotive, aircraft, etc. The analysis, control

and stabilization of switching converters are the main factors that need to be

considered. Many control methods are used for control of switch mode dc-dc

converters and the simple and low cost controller structure is always in demand

for most industrial and high performance applications. Every control method has

some advantages and drawbacks due to that particular control method consider

as a suitable control method under specific conditions, compared to other control

methods. The control method that gives the best performances under any

conditions is always in demand.

1.2 Switching regulator

Voltage regulation conventionally has been done by Linear Regulators but

slowly is being replaced with Switching Regulators. To realize the importance

of a switching regulator we will first compare its efficiency with a linear

regulator. The resistance of the linear regulator varies in accordance with the

load resulting in a constant output voltage.

If we consider an example, where V in = 24 and we want to have a V out = 12. In

this case we need to drop 12 volts across the regulator. This results in a mere

50% efficiency for the linear regulator and a lot of wasted power which is

normally transformed into heat. Providing for heat sinks for cooling makes the

regulator bulky and large. Hence, where size and efficiency are critical, linear

voltage regulators cannot be used. When we come to the switching regulator

these problems doesn't face.

The switching regulator is a simple switch (and hence ideally no resistance or

very low resistance). This switch goes on and off at a fixed rate (usually

between 50 KHz to 100 KHz). The time that the switch remains closed during

each switch cycle is varied to maintain a constant output voltage. The switching

regulator is much more efficient than the linear regulator achieving efficiencies

as high as 80% to 95% in some circuits. In contrast, the linear regulator usually

exhibits only 50% to 60% efficiency. With higher efficiency smaller heat sinks

will be required because lesser heat is dissipated.

There is also another advantage of Switching Regulators and that is the energy

stored by inductor & capacitor can be transformed to output voltages that can be

greater than input (boost), or can be transferred through a transformer to provide


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electrical isolation with respect to the input. Unlike linear regulators, switched

power supplies can step up or step down the input voltage.

2. Theory

2.1 Input DC voltage source:

The input voltage for step down (buck) converter can be unregulated DC

voltage such as solar energy or wind power. Also we can have regulated DC

voltage from battery. For our mini project, the source of DC voltage is 400V DC

from micro grid bus for a battery application, typically 300 V battery and 20

kW.

2.2 Switching Devices:

MOSFETs and IGBTs are being heavily utilized in most of today's modern

power electronics equipments and systems. The MOSFET is a device that is

voltage- and not current-controlled. MOSFETs have a positive temperature

coefficient, stopping thermal runaway. The on-state-resistance has no theoretical

limit, hence on-state losses can be far lower. All these advantages and the

comparative elimination of the current tail soon meant that the MOSFET

became the device of choice for power switch designs. The IGBT has the output

switching and conduction characteristics of a bipolar transistor but is voltage-

controlled like a MOSFET. In general, this means it has the advantages of high-

current handling capability of a bipolar with the ease of control of a MOSFET.

However, the IGBT has lower switching speeds, and higher switching losses as

compared to MOSFET.

For the input voltage up to 200 V, MOSFETs are better than IGBT. From 200 V

up to 400 V, we can choose either of the two. However, if the input voltage is

above 400 V it is strongly recommended to use IGBTs than MOSFET due to

their high power handling capacity.

2.3 Low Pass Filter (LPF):

Selecting the proper value of inductors and capacitors has a great impact on the

ripple output voltage. The converter ripple voltage is dependent upon ripple

current. The size of the inductor determine the conduction mode, continuous or

discontinuous of our converter and it is dependent upon switching frequency.

The magnitude of switching ripple in the output voltage in a properly designed

DC supply is much less than the dc component. The peak inductor current which

is equal to the DC component plus the peak to average ripple ΔI L/2, flows

through the semiconductor switches and is necessary when specifying device


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ratings. To reduce the peak current a larger value of inductor is required. A

secondary benefit in lowering the ripple current is that it reduces inductor, ESR

and load losses.

Selection of the output capacitors depend on the capacitance and output ripple

current. The only steady state component of output capacitor current is that

arising from the inductor current ripple. Hence inductor current cannot be

neglected when calculating the output voltage ripple. The inductor current

contains both a DC and ripple current component. The DC component must

flow entirely through the load resistance R. While the ac switching ripple

divides between the load resistance R and the filter capacitor C.

Fig 1. Reversible power flow with reversible of the output current

2.4 Control of a DC - DC Converter :

Pulse Width Modulation (PWM) is the most widely used method for controlling

the output voltage. It maintains a constant switching frequency and varies the

duty cycle. Duty cycle is defined as the ratio of switch on time to reciprocal of

the switching frequency (fS). Since the switching frequency is fixed, this

modulation scheme has a relatively narrow noise spectrum allowing a simple

low pass filter to sharply reduce peak-to-peak ripple at output voltage. This

requirement is achieved by arranging an inductor and capacitor in the converter

in such a manner as to form a low pass filter network. This requires the

frequency of low pass filter (fc) to be much less than switching frequency (fs),

fc<<fs.

The following are commonly used control methods in Pulse Width Modulated

(PWM) : PWM voltage mode control, PWM current mode control with

Proportional (P), Proportional Integral (PI), and Proportional Integral Derivative

(PID) controller. These conventional control methods like P, PI, and PID are

unable to perform satisfactorily under large parameter or load variation.

Therefore, nonlinear controllers come into picture for controlling dc-dc

converters. The advantages of these nonlinear controllers are their ability to


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react suddenly to a transient condition. Different types of nonlinear controllers

are hysteresis controller, sliding mode controller, boundary controller, etc.

The hysteresis control methods for power converters are also gaining a lot of

interest due to its fast response and robust with simple design and

implementation. The hysteresis controllers react immediately after the load

transient takes place. Hence the advantages of hysteretic control over other

control technique include simplicity, do not require feedback loop

compensation, fast response to load transient. However, the main factors need to

be considered in case of hysteresis control are variable switching frequency

operation and stability analysis.

Most of the feedback controllers for dc-to-dc converters generally employ a

high-gain operational amplifier as an error amplifier in order to regulate the

output voltage. Phase compensation circuits are inevitably used not only to

improve

the dynamic performance of the converter but also to compensate the effect of

phase lag in the operational amplifier itself. It is known that a current mode

control together with the voltage feedback is one of solution to improve the

stability

and the dynamic performance . In this case, a current sensing resistor which may

cause the increase of power loss and a number of components in the control

circuit are necessary.

In voltage mode, a hysteretic PWM control for buck converters using output

ripple voltage shows inherently stable performances since the feedback signal

includes inductor current information . However, the switching frequency and

the dynamic performances strongly depend on the Equivalent Series Resistance

(ESR) of the output capacitor in this method. We have reported the buck

converter employing an RC network directly connecting to the inductor winding

instead of using the output ripple voltage and it provides quick response to the

output voltage variations and excellent dynamic performances without using the

error amplifier.


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3. Circuit Parameters and Loss Analysis

3.1 Circuit parameters optimization

3.1.1 Inductance selection

The inductor design has a significant impact on the system performance such as

device switching loss, inductor power loss, system volume etc. it is necessary to

optimize the inductance with all the design considerations.

The relationship with inductor peak current Ipeak, minimum current Imin and

inductor RMS current ILRMS can be Zmin equation. Where TS is the switching

frequency ILoad is load current, P is the load power and ∆I is inductor current

ripple.

∆I =

IO = ILoad =

IPeak = ILoad + ∆I

Imin = ILoad - ∆I

IRMS =

The inductance allows the converter operating under the boundary condition

with DCM & CCM

Lcr =

it’s the critical inductor point boundary condition

For CCM Lcr > L or for DCM Lcr < L

Ipeak increases with decrease of inductor value

Ipeak decreases with increase of inductor value

The relationship with inductor peak current Ipeak minimum current Imin are as

follows and we can also calculate the value of inductor for our specific designs

as follow:

∆I =

ILoad =


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IPeak = ILoad + ∆I/2

Imin = ILoad - ∆I/2

IRMS =

∆I = 30%

∆I =

ILoad =

∆I =

Ipeak = 66.67+10 = 76.67 A

Imin = 66.67-10 = 56.67 A

∆I =

3kHz < fs < 10kHz

For fs = 3kHz

L =

For fs = 10kHz

L =

3.1.2 Snubber capacitor optimization

The snubber capacitor design can be simply design (done) with charge balance

of CV2 and ½*Li

2

C*V2=1/2*Li

2

V is capacitor voltage

I is inductor current

Calculating the snubber capacitor selection consideration

CV2 = 1/2 *L

Where V is voltage of capacitor


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iL is inductor current

C is snubber capacitor

V = Voutput

CV2 = 1/2 *L

C =

iL = 66.67 A Vout = 300 Volt

For fs=10kHz

C =

For fs = 3kHz

C =

3.2 Switching Losses with Linear MOSFET Output Capacitance

Let us assume that the MOSFET output capacitance Co is linear. First, we shall

consider the transistor turn-off transition. During this time interval, the transistor

is OFF, the drain-to-source voltage vDS increases from nearly zero to VI , and

the transistor output capacitance is charged. Because, the charge transferred

from the input voltage source VI to the transistor output capacitance Co during

the turn-off transition is:

dQ = CodvDS

= CoVI

Where Co= 2Cds

yielding the energy transferred from the input voltage source VI to the converter

during the turn-off transition,

= VIQ = CoVI2

Using

dWs = QdvDS/2

the energy stored in the transistor output capacitance Co at the end of the

transistor turn-off transition when vDS = VI is given by

= 1/2VIQ = 1/2CoVI

2


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Thus, the energy lost in the parasitic resistance of the capacitor charging path is

the turn-off switching energy loss described by:

Wturn-off = WVI - Ws = CoVI2- 1/2CoVI

2= 1/2CoVI

2

which results in the turn-off switching power loss in the resistance of the

charging path

Pturn-off = Wturn-off /TS =1/2fSCOVI2

For our min-project we take such type of MOSFET: An IRF510 power

MOSFET with VB = 0.774158 V, Crss = 25 pF, and Coss = 100 pF is

operated in the buck PWM converter

The capacitances Crss and Coss are measured at VDS = 25V and VGS = 0V.

Hence, Cds25 = Coss − Crss

Cds25 = Coss − Crss

Cds25 = 100-25 =75pF

Cds =

= 75pF*

= 19,019pF

CO = 2*19,019pF

= 38,038pF

Pturn-off =0,5*10000*38,038*10-12

*(400)2

=0,304W

=30,4mW

After turn-off, the transistor remains in the off-state for some time interval and

the charge Ws is stored in the output capacitance Co.

Now consider the transistor turn-on transition. When the transistor is turned

on, its output capacitance Co is shorted out through the transistor on-resistance

rDS, the charge stored in Co decreases, and the drain-to-source voltage

decreases from VI to nearly zero. As a result, all the energy stored in the

transistor output capacitance is dissipated as heat in the transistor on-resistance

rDS. Therefore, the turn-on switching energy loss is:

Wturn-on=1/2CoVI2

resulting in the turn-on switching power loss in the MOSFET

Pturn-on =1/2fSCOVI2


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Similarly to the calculation of the turn-off switch power loss we can also

calculate for power loss on turn on switch.

therefore

Pturn-on =1/2fSCOVI2

=0,5*10000*38,038*10

-12*(400)

2

=0,304W

=30,4mW

The turn-on loss is independent of the transistor on-resistance rDS as long as the

transistor output capacitance is fully discharged before the turn-off transition

begins.

The total switching energy loss in every cycle of the switching frequency during

the process of first charging and then discharging the output capacitance is given

by

Wsw = Wturn-off + Wturn-on = COVI2

and the total switching power loss in the converter is

Ploss = fSCOVI2

= 10000*38,038*10-12

*(400)2

=0,612W

=61,2mW

For a linear capacitance, one-half of the switching power is lost in the MOSFET

and the other half in the resistance of the charging path of the transistor output

capacitance.

But in diode it is different from MOSFET a diode cannot discharge its parallel

capacitance through its forward resistance. This is because a diode does not turn

on until its voltage drops to the threshold voltage. However, the junction diodes

suffer from the reverse recovery at turn-off.


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3.3 Power Losses and Efficiency of Buck Converter

An equivalent circuit of the buck converter with parasitic resistances is shown in

Figure 1. In this figure, rDS is the MOSFET on-resistance, RF is the diode

forward resistance, VF is the diode threshold voltage, rL is the ESR of the

inductor L, and rC is the ESR of the filter capacitor C.

The conduction losses will be evaluated assuming that the inductor current iL is

ripple-free and equals the dc output current IO. Hence, the switch current can be

approximated by

Fig 2 sample design for DC-DC buck converter

iS =

which results in its rms value:

ISrms =

=

= IO

and the MOSFET conduction loss can be calculated as follow

PrDS = rDS ISrms2 = DrDSIo

2 =

PO

for our parameters =0,1mOhm, D = 0,75 RL = 4,5 Ohm

PrDS = 0,1*0,75*20000/4,5

= 333W

The transistor conduction loss PrDS is proportional to the duty cycle D at a fixed

load current IO. At D = 0, the switch is OFF during the entire cycle and therefore

the conduction loss is zero. At D = 1, the switch is ON during the entire cycle,


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resulting in a maximum conduction loss. Assuming that Dmax = VO/VImin as for

the lossless converter, the maximum MOSFET conduction power is

PrDSmax = DmaxrDSIOmax2 =

POmax =

POmax

Assuming that the transistor output capacitance Co is linear, the switching loss is

expressed by

Psw = fsCoVI2 =

=

Po

Mvdc is nothing but it stands for D it is the transfer function. The maximum switching loss is

Psw(max) = fsCoVImax2 =

=

Po

Excluding the MOSFET gate-drive power, the total power dissipation in the

MOSFET is

PFET = PrDS + Psw/2 = DrDSIo2 + 1/2fsCoVI

2

= (DrDS/RL + fsCoRL/2MVDC

2)POmax

Similarly, the diode current can be approximated by

iO =

yielding its rms value,

IDrms =

=

= IO

and the power loss in RF ,

PRF = RF. IDrms2 = ( ) RFIO

2 =

PO

for our design for the given specification of the parameters PRF can

given as:

PRF = 0,25*0, 1*20000/4,5

= 111W


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The average value of the diode current is

ID

(1-D)Io

which gives the power loss associated with the voltage VF ,

PVF = VF ID = (1 − D)VF IO

= (1 − D)VF PO/VO

= 0,25*0,78*20000/300

= 13 W

Thus, the overall diode conduction loss is

PD = PVF + PRF

=(1 − D)VF PO/VO +

PO

= 13+111 = 124,11W

The diode conduction loss PD decreases, when the duty cycle D increases at a

fixed load current IO. At D = 0, the diode is ON during the entire cycle, resulting

in a maximum conduction loss. At D = 1, the diode is OFF during the entire

cycle and therefore the conduction loss is zero. The maximum diode conduction

loss is

PDmax = (1 − Dmin)

POmax

=

POmax

Typically, the power loss in the inductor core can be ignored and only the

copper loss in the inductor winding should be considered. The inductor current

can be approximated by:

iL = IO

leading to its rms value


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ILrms = IO

and the inductor conduction loss

PrL = rLI 2

Lrms= rLI2

O =

PO

= 0, 1*20000/4,5

= 444 W

The maximum power loss in the inductor is

PrLmax = rLI 2

Lrms= rLI2

Omax =

PO

The rms current through the filter capacitor is found to be

ICrms =

=

=

and the power loss in the filter capacitor can be computed as

PrC =rC ICrms2 =

=

=

typically for our design the power loss through parasite resistor of capacitor is

PrC = 4,5*0,25^2*20000*1/(12*10000^2*(2,22*10^-3)^2)

= 968mW

The maximum power loss in the capacitor is

PrC =

=


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The overall power loss in the buck converter is computed as:

PLS = PrDS + Psw + PD + PrL + PrC

= DrDSIo2 + fSCOVI

2 + (1 − D)VF PO/VO +

PO

+ rLI2

O +

PLS =

RL 2rC12fs2L2 PO

therefore the overall power loss numerically can be computed as

PLS = 968mW + 444 W +124,1W +333W +61,2mW

= 902,03W

Therefore the converter can be evaluated it efficiency using

η =

=

=

η =

for our mini-project its efficiency is computed numerically as:

η = (20000/20902,03)*100%

= 95,67%


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If the inductor peak-to-peak current ripple Δ is taken into account, the rms

values of the switch is given by

ISrms =

= IO Δ

where ISmin = IO − /2 and ISmax = IO + /2. Similarly, the rms value of the

diode

IDrms =

= IO Δ

where ILmin = IO − /2 and ILmax = IO + /2. Similarly, the rms value of the

inductor is

ILrms =

= IO Δ

The conduction power loss in the MOSFET is given by:

PrDS = rDSI 2Srms= rDSDI

2o

Δ

= 0,1*0,75*66,67^2(1+20^2/(12*66,67^2))

=335,87W

The conduction power loss in the diode forward resistance is

PRF = RF I2Drms= RF (1 − D)I

2O

Δ

= 111,95W Assuming that the inductor resistance rL is independent of frequency, the power

loss in the inductor winding is given by:

PrL = rLI 2

Lrms= rLI2

O Δ

=447.8W

there for total power loss can be calculated as :

PLS =447.8 +111.95+335.87+13+0.986+0.062

=909,69W

η = 20000/20909.69*100%

=95,649%


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4. Circuit Design and Simulation Results

4.1 Main Converter Circuit design

Fig 3. Main DC- DC Converter circuit design

4.2 MESFET Gate signal (PWM)

This refers to PWM which trigger the gates of the MOSFETs

Fig 4. PWM Triggering circuit


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The output of the above circuit is as follows :

Fig 5. Output Waveform of PWM.

4.3 Ripple Inductor Current

We are given that the ripple current of the inductor to be 30% of the total inductor current.

Here under the waveform of inductor current.


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Fig 6. Ripple Inductor current waveform

4.4 Output Voltage

The simulation result shows that the output ripple voltage of our converter is from 290 up to

300V ( ΔV o = 10 V) which means 3.33% of the required output voltage. Even though, the

desired output voltage is 300 V due to switching losses and conducting losses we found

average voltage of 295 V.

The SIMULINK simulation waveform of ripple voltage is as shown below.


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Fig 7. ripple voltage output


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5. Conclusion In this project we have learnt a lot of things, we have got hands on practice on how to

manipulate MATLAB SIMULINK software with simpower system interfacing . In Our

project we use PI control Voltage Source Control (VSI) method. However Hysteresis control

also can be used to convert DC to DC by comparing the output current with two boundaries.

The output ripple voltage of this converter is very good but some effort need to exert to

decrease the ripple inductor current further.

6. Reference Pulse Width Modulation DC-DC power converter, Marian K.

Kazimierczuk

Power Electronics 3rd edition , Mohan, Undeland, Robbins

K. Liu and F. C. Lee, Zero-voltage switching technique in dc/dc

converters. IEEE Power Electronics Specialists Conference Record,

1986, pp. 58–70.

K. D. T. Ngo, Generalization of resonant switches and quasi-resonant

dc–dc converters.

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