A CMOS Multiparameter Biochemical Microsensor With

2030 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 36, NO. 12, DECEMBER 2001

A CMOS Multiparameter Biochemical MicrosensorWith Temperature Control and Signal InterfacingErik Lauwers, Jan Suls, Walter Gumbrecht, David Maes, Georges Gielen, and Willy Sansen, Fellow, IEEE

Abstract—A fully integrated multiparameter microsensor chipis presented for continuous monitoring of concentrations of dif-ferent blood gases (e.g., pH, pO2, pCO2), ions, and biomolecules,and for conductometric measurements. The chip can monitor upto seven different chemical substances depending on the mem-branes deposited on the sensor units (on-chip ion-sensitive field-effect transistors (ISFETs), amperometric and conductometriccell). The sensors, which are positioned in a flow channel, aresurrounded by on-chip interfacing and processing electronics sothat external readout goes via a simple data acquisition card. Inaddition, temperature control of the measured fluids and a one-time-use security check have been provided for proper operation.Fabrication was done in a standard 1.2- m CMOS process towhich extra postprocessing steps have been added for the chemicalsensors and membranes. The chip operates at 5 V and the totaldie area is 25.7 mm2. Full integration is obtained including theISFETs and ISFET buffers, as well as a reference electrodestructure, all integrated on the same chip in the same technology.

Index Terms—Amperometric and conductometric cell, biochem-ical, CMOS microsensor, EPROM, ISFET, signal interfacing, tem-perature control.

I. INTRODUCTION

M ASS production, high yields, and low manufacturingcosts are concepts readily associated with electronics

integrated on a single piece of silicon. Other considerationssuch as low power, high speed, etc., are also motivations forsilicon integration. Apart from these factors, small dimensionsand portability are of great value in the field of medicalhealthcare.

Analysis of blood gases in intensive-care units is commonpractice nowadays. For this analysis, many samples can be re-quired every day, consuming a lot of blood, time, and resources.Continuous monitoring of blood gases is therefore a majorimprovement for critical-care patients and reduces the amountof blood samples needed. For many other applications, as forexample, in bioreactors, the possibility to perform continuousmeasurements is also advantageous.

This paper presents the implementation of a complete mi-crosensor system for the continuous monitoring of ions, dis-solved gases, and biomolecules. Even more functionality hasbeen integrated on chip, such as a conductometric sensor, anon-chip absolute temperature control, and a one-bit EPROM

Manuscript received March 22, 2001; revised July 20, 2001. This work wassupported by the Brite-Euram COMMONSENS project.

E. Lauwers, J. Suls, G. Gielen, and W. Sansen are with the KatholiekeUniversiteit Leuven, ESAT-MICAS, 3001 Leuven-Heverlee, Belgium (e-mail:[email protected]).

W. Gumbrecht is with Siemens AG, Erlangen, Germany.D. Maes is with IMEC, 3001 Heverlee, Belgium.Publisher Item Identifier S 0018-9200(01)09334-9.

for medical security reasons. The full system is processed ina 1.2- m single-metal single-poly CMOS technology and op-erates at 5 V. Extra postprocessing steps have been added tomanufacture the sensor (interface) structures and a calibrationsystem. The total chip area is 4.11 mm6.25 mm. The processtechnology used was explained in detail in [1]. In this paper,the overall system and the electronic subblocks are explainedin detail and measurements are presented. Minor additional de-tails can be found in the visuals supplement of [2]. The paperis organized as follows. Section II presents an application forthe microsensor chip and explains the global system setup. InSection III, the sensors are described, and in Section IV, thesensor interfacing electronics. Section V covers the heat regula-tion system, and Section VI explains the control electronics andthe EPROM. In Section VII, chemical measurements are pre-sented, and finally, in Section VIII, conclusions are drawn.

II. TOTAL MICROSENSORSYSTEM

A typical use of the microsensor is measuring gas concentra-tions in blood. The chip is packaged such that the central part,where the sensors are aligned, is exposed to a flow channelwhere the blood and calibration solutions run through. Thissensor alignment (centerline a–h along the long side) and theflow-channel perimeter around it (oval shape for O-ring to sealoff the flow channel) can clearly be seen in Fig. 1. For a smallmoment during measurement, the flow is stopped such that asmall blood sample is trapped at the sensor interface. Aftera short period needed to make sure that the sample acquiresthe correct temperature, measurement results are read in bya monitor. This monitor can be as simple as a laptop with adata acquisition card. Keeping the temperature of the measuredsamples constant is important for reproducibility of the resultsand for comparison with other measurements (eventuallyperformed elsewhere).

The main goal of the sensor development is twofold. First, thecontinuous monitoring of blood gasses must be possible with in-tegrated sensors. Second, the system may only have a minimalamount of external connections in order to keep the wire con-nection to the monitor small. The minimal amount of externalconnections is five: the power supply connections (and ),the clock signal, and the I/O connections. Three extra wires havebeen added so that a standard eight-wire connection is obtained(Fig. 2). The three extra wires are an external reference voltage

for reasons of controllability, an external reference biasingcurrent, and an extra ground connection to have a good analogreference even with currents up to 100 mA. Various versions ofthe system with different sensor configurations have been man-ufactured. Because from an electronic point of view, these ver-

0018–9200/01$10.00 © 2001 IEEE

LAUWERS et al.: CMOS MULTIPARAMETER BIOCHEMICAL MICROSENSOR 2031

Fig. 1. Layout of the total system.

Fig. 2. Block diagram of the sensor chip.

sions do not differ much, only one is highlighted here and willbe fully explained.

Currently, commercial biosensors exist that provide the samerange of possible measurements. A prominent commerciallyavailable biosensor is the I-STAT from the I-STAT Corporation.This biosensor works mainly according to the same principles,however, there are two important differences. The sensors aremicrofabricated thin-film electrodes, whereas in this work, fullyintegrated sensors are used. The I-STAT does not allow for con-tinuous measurements, but uses blood samples to be placed incartridges allowing nearly instantaneous measurements.

In Fig. 2, a block diagram of the sensor chip is given. It canroughly be divided in three parts. The first part contains the sen-sors and their interfacing electronics. The second part containsthe extra functionalities: temperature control and an EPROM.

The third part contains all the electronics necessary to controlthe chip and the external communication as well as the biasing.These different subsystems will be explained in Sections III–VI.

III. SENSORS

The chip has eight integrated sensors, indicated by letters inFig. 1. Seven sensors are located in the centerline and one isabove the centerline (e). The number of eight is determined bythe available space in the flow channel. They can all be oper-ated in parallel if required by the user. However, the idea is tomake one standard chip to reduce manufacturing costs and thento program the chip according to the needs of the user. For ex-ample, if the oxygen sensor output is made available only atthe chip’s output by use of an EPROM similar to the security


Fig. 3. ISFET calibration setup.

EPROM, then the user only has to pay for this reduced function-ality. Dimensions of the individual sensors are determined bychemical considerations from previous prototypes. The size ofthe micropool in polyimide around the sensor’s active area is de-termined by the used selective-membrane dispensing technique.More information concerning the membrane dispensing is givenin [3]. It was decided to integrate six ion-sensitive field-effecttransistors (ISFETs) (b, c, d, e, f, and g in Fig. 1), one oxygensensor (h), and one conductometric sensor (a). An ISFET is ba-sically a field-effect transistor of which the gate metalizationhas been omitted. The operation is based on direct contact ofthe electrolyte with the gate oxide. The adsorption of chargedspecies at the solution–oxide interface is measured. The con-ductometric sensor, which measures the electric conductivityof an electrolyte solution, is built with two parallel sensors ofwhich one electrode is shared (the middle one). A sinusoidalinput signal is applied at the electrodes and the current throughthe electrodes which is dependent on the composition (numberof charged elements, ions and their properties) of the solution, ismeasured. The ISFETs can be used to measure the voltages be-tween the conductometric electrodes, bypassing the (interface)impedances of the current providing electrodes, hence allowinga four-point conductometric measurement. Other versions man-ufactured contain different sensor units such as an enzymaticor pCO sensor. The sensor chip continuously monitors ions,dissolved gases, and biomolecules. Traditionally, external refer-ence electrodes are used for stable measurements [4]. However,stable potentials can only be obtained with a fixed chloride con-centration in the sample solution. To tackle this problem, a refer-ence electrode with an Ag–AgCl interface has been integrated ina bypass structure (Fig. 1, oval structure around a and e) next tothe flow channel. The layout of this bypass structure is based ona previous design to monitor blood gasses [5]. The ISFET cali-bration setup is schematically shown in Fig. 3. During a calibra-tion cycle (pump two is on, pump one is off), the bypass struc-ture traps the calibration fluid on top of the reference ISFET.During measurement (pump two is off, pump one is on), thiscalibration solution then remains in place while all the other IS-FETs see the fluid being measured. Different solutions for thisreference problem have been published, such as, for example, asolution based on different pH-sensitive electrodes [5].

Fig. 4. Transistor schematic of the ISFET buffer: a modified symmetrical OTAwith pMOS source follower.

IV. I NTERFACEELECTRONICS

A. ISFET Buffer Amplifier

The response of an ISFET can be measured in two ways. Theapplied voltage (Fig. 4) can be held constant and the changein the drain current is measured as a function of the ion activity.Alternatively, the applied voltage can be changed in sucha way that the drain current remains constant. Here, the cur-rent is kept constant and the gate voltage of the ISFET is mea-sured through a buffer and then sent to the monitor for analysis.Also, for correct operation the drain-to-source voltage of theISFET has to be kept constant. Both operations are combinedin an ISFET buffer with low offset, which is a modified sym-metrical OTA with pMOS source follower (Fig. 4). The ISFETand its buffer are integrated in one process and on the samedevice to enable a more stable operation. By introducing thebootstrap transistors T2a,b using the same technology as T1a,b,i.e., a normal CMOS FET but with an oxide–nitride gate di-electric and a Pt gate, the drain–source voltages over transistorsT1a,b are kept constant. The bulk effect of transistors T1a,b iscancelled out by making this effect equal for transistors T2a,band so the drain–source voltage of T1a,b is kept constant. ThepMOS source follower T5 is added to match the input and outputvoltage swing and also to lower the output impedance. Measure-ments on previous prototypes indicated that this design is veryrobust toward technology variations and can handle an input(output) voltage from about 2 V up to 5 V with a supply of 5 V.

The ISFETs are biased in the correct linear operating pointby an external connection . It was chosen to directly applythis voltage to make sure that it is a well-controlled electricalsignal. The actual measured voltage at the solution–oxide inter-face is indicated by . In addition, this allowed having a1:4 demultiplexer instead of a 1:5 demultiplexer, which in turnallowed an easier design for the controller, as will be explainedin Section VI.

B. Potentiostatic Setup

In Fig. 5, a schematic of the potentiostatic setup is given foran amperometric measurement. The counter and working andreference electrodes are indicated by the letters, , and ,respectively. The actual open sensor window is sketched by thecircle around them. To prevent polarization of the reference due


Fig. 5. Schematic view of the potentiostatic setup.

to current drawn through it, an inert counter electrode is intro-duced to provide the current for the reduction (or oxidation) atthe working electrode. The voltage at the counter electrode isregulated such that the reference electrode is kept in the sameoperating point as for the ISFETs. By changing the inputvoltage , the potential at the working electrode can be set toinduce an electrochemical, faradic current. This current is thenconverted and amplified to a voltage through a feedback resistor

. This voltage is then sent to the output. The currents are verysmall (typically nanoamperes) and must be amplified for furthersignal handling. For a given input potential and chemical con-centration, the size of sets the maximal output range andthus the sensitivity of the current-to-voltage converter (con-verter). If

or (1)

For fixed , is proportional to .To allow large variations in and to allow a large range

of different chemical solutions with varying gas concentrations,two possible output ranges were foreseen. This was done byreplacing with a -resistor network, as shown in Fig. 6.If the connection is left open and , thenthe feedback resistor is . If, however, the connectionis applied, then a new feedback resistance is obtained which islarger than . The sensitivity is then increased:

or

(2)

In this design, a difference in sensitivity factor of 4.33 was taken.With the highest sensitivity selected, an input current of 100 nAgives a of 1 V. This corresponds to an effective resistance of10 M . The bondpad in the lower right corner of Fig. 1 is con-nected to the extra resistor branch and can easily be connectedto the bondpad on its left during packaging, which is thebondpad. The complete setup of Fig. 5 is integrated on chip.

C. Conductometric Sensor

As explained above, it is also possible to perform a completefour-point conductometric measurement on chip. The driving

electrode [Fig. 1(a)] is connected to an opamp and resistor struc-ture (see left-hand side of Fig. 6) to convert the current into anoutput voltage. Limited by the individual blocks concerned (theISFETs and the converter), signal frequencies up to 5 kHzcan be applied. This relatively low frequency originates fromlow offset specifications for the different building blocks, whichnecessitate large transistors and therefore limit the gain band-width (GBW).

V. HEAT REGULATION

For reproducible blood-gas measurements, the on-chip tem-perature needs to be fixed (for example, at 37C) throughoutdifferent measurements. Also, different fluids can only be com-pared validly if the temperature during different measurementsis the same. To keep the on-chip temperature constant, a heat-regulating loop including a temperature-sensing device and aheating device have been integrated. The temperature-sensingdevice is a parasitic vertical p-n-p bipolar transistor. If the cur-rent through the bipolar device is kept constant, then the voltagedrop over the bipolar transistor, which is connected as a diode(inset in Fig. 7), is proportional to the absolute temperature, asgiven by

(3)

is the leakage current at zero biasing, is the biasingcurrent of the transistor, and the absolute temperature. FromFig. 7, it can be seen that for every increase in temperature of1 , drops about 2.2 mV. This voltage (also referred toas ) can now be used to control the chip temperature, asindicated in the inset in Fig. 7. The heating device is a normalnMOS transistor and is designed to be able to generate 0.25 Wor 50 mA with a 5-V power supply. Through the feedback loop,the heating device is turned on to generate enough heat to keepthe on-chip temperature constant. The temperature is set through

, which is an input control voltage to the system (through the“Sensor in” pad in Fig. 2).

The thermal time constant of the regulating loop was mea-sured by externally applying a 5-V pulse to the gate of the heatertransistor and then reading out (Fig. 8). This means goingfrom no heating power to full heating power and back.varies according to a combination of two exponential functions:one exponential that characterizes the heat resistance and heatcapacitance of the silicon, and one that characterizes the heatloss to the environment [7].

(4)

It is assumed that the thermal resistance of the silicon is smallcompared to the thermal resistance to the environment [7], sothat the heating feedback system has to be designed with thefastest of both time constants in mind. The measurements inFig. 8 yield time constants of 0.0095 s and 0.15 s supporting thisstatement. This means that wherever both are put on the samechip, no measurable effect is caused on the feedback system.The second exponential is very dependent on the chip packageused. The feedback loop is designed with the first pole being the


Fig. 6. Modified feedback impedance for two possible ranges of operation.

Fig. 7. Temperature regulation: measured base–emitter voltage dependence of a bipolar transistor on the environment temperature and schematic of how this isused in the heat-regulating loop.

fastest thermal pole. Also, the attenuation of the thermal pathcan be estimated by extrapolating the curves of Fig. 8 and is inthe range of . This information is sufficient for designingthe temperature-control loop.

During measurements of the temperature-control loop, anelectronically controllable thermochuck was used to set theambient or environment temperature of the chip. In Fig. 9, ameasurement result of the temperature-control loop is given.

is the voltage drop over the bipolar transistor that is inthe feedback loop. is the voltage over a second integratedbipolar transistor. is the current for the total systemthrough the power supply. A fully automated measurementwas performed by automatically changing the thermochucktemperature. The result given is an average of ten measuredvalues. The temperature value was derived with an accuracy ofabout 1 mV. The difference between both values is dueto the resistance of the routing. With the given heating power,between 33C and 47 C ambient temperature, the voltage overthe bipolar transistors changes only with 3.4 mV. This is equal to0.24 mV/ C, which is almost a factor of ten smaller than withoutregulation. In a fully mounted and packaged chip, this valuewill be even better, because the heat loss to the environmentwill then be smaller. The chemical measurements presented inSection VII are performed on a fully packaged device.

Fig. 8. T as a function of time when the heater gate voltage switches from0 to 5 V and back.

VI. CONTROL ELECTRONICS

The control electronics consist of an EPROM (in Fig. 2),a controller, and a line driver ( in Fig. 2).

A. EPROM

The sensor chip was primarily developed for (but not limitedto) medical use. For health care reasons, the chip may only beused on one patient and this for a limited time. After use, thechip must be discarded. To make sure that two patients neveruse the same chip, a single-bit EPROM is integrated on the mi-crosensor chip. The first time the sensor is used, the EPROM


Fig. 9. Fully automated chip temperature measurement.

TABLE IOVERVIEW OF THE EPROM FUNCTION

is read. If it has not been used, then the chip can be used andthe EPROM is programmed to avoid future use. This is a nonre-versible operation and therefore provides a valid security check.Use of an EPROM can even be further exploited in future ver-sions. If more than one bit is integrated and some logic is added,the EPROM could be programmed to indicate the kind of sensorconfiguration that is available on this particular chip. Then themonitor can read this and automatically set the correct parame-ters.

The core of the EPROM is a floating-gate nMOS transistor(FGT) to which a control gate is capacitively coupled. It is nota standard EPROM circuit solution. This control gate can havetwo different voltages: 1.5 and 8 V. In order to generate this 8 V,a charge pump was also integrated. Before programming, thethreshold voltage of the FGT is small, so that with a control-gatevoltage (CGV) of 1.5 V, a current of the order of 10A flowsthrough it. If the CGV is now increased to 8 V during writemode, then the current increases drastically, creating hot elec-trons that are trapped in the gate oxide of the FGT, which in-duce an upward threshold voltage shift. When the CGV is nowbrought back to 1.5 V, only a small current of the order of 10 pAflows through the FGT. This current is measured and an outputbit is sent to the monitor. The functioning of the EPROM is sum-marized in Table I.

A schematic of the total EPROM is given in Fig. 10. Thecharge pump is a three-stage bootstrap structure. The outputvoltage (8 V in this design) can be varied easily by resizing theoutput capacitor. In order not to load this output capacitance toomuch, transistors M3 and M4 need to have a high drain–sourceresistance. Hence, M3 has a channel length of 20m and M4 of40 m, and both have a width of 2m. The sizing of transistorsM1 and M2 can be explained as follows. Assuming that a lowR/W signal means “read,” then in read mode, transistor M1 ison and M2 is off. In case the FGT has not been programmed, alarge current flows through M1, while only a small current flowsif the FGT has been programmed. This means that if M1 has a

Fig. 10. Schematic view of the complete EPROM.

large drain–source on-resistance, the output is high (low) whenthe FGT has (not) been programmed. The W/L of M1 has beentaken as . If the R/W signal is in write mode, then M2is on and has to conduct a lot of current without lowering thedrain–source voltage of the FGT too much. Hence, M2 needs alarger W/L, and a value of has been taken.

B. Controller

The minimal amount of external connections is five: thepower supply connections and , the clock signal, andthe I/O connections. Three wires have been added so thata standard eight-wire connection is obtained: an externalreference voltage for reasons of controllability, an externalreference biasing current, and an extra ground connection tohave a good analog reference even with currents up to 100 mA.To this end, the controller (part 3 of Fig. 2) has to redirect 16outputs to one output line and redirect one input line to fourdifferent inputs. This necessitates the use of a 16:1 multiplexerand a 1:4 demultiplexer. The schematics of the multiplexerand demultiplexer are identical. Only one decoder is neededif, every fourth clock cycle, the same demultiplexer input isselected, and every 16th clock cycle, the same multiplexerinput is selected. This can easily be achieved by taking a 1:16demultiplexer instead of a 1:4 and connecting every fourthoutput together. This increases the used area a little, but muchless than a second decoder would. The synchronization is doneby including three analog biasing voltages in the chain of 16multiplexer inputs. Analog voltages eliminate the possibilityof still having hard faults in the hardware without noticing,and hence provide an extra check to see if the electronics workproperly. At each demultiplexer output, a hold capacitance of40 pF has been placed to temporarily store the input voltageswhile the other ports are being refreshed. This value wascalculated from the chosen specifications for minimal clockfrequency of the system, maximal voltage droop, and a securitymargin of 25%. If a minimal frequency of 5 kHz and a maximaldroop of 1 mV are taken, then

leakage current

F (5)

The decoder and gray counter are well-known digital buildingblocks and will not be further explained. The controller can in-trinsically be clocked at 1 MHz according to simulations, butthe chip’s speed is limited by the analog electronics which have


Fig. 11. Rail-to-rail line driver (biasing circuit not drawn).

to drive large loads with a reasonable power consumption. Thetotal system speed is limited by the DAC card used in the mon-itor and the software. This limits the speed for the prototypesystem to maximally about 20 kHz.

C. Line Driver

To bring the analog signals off chip, a line driver is needed.This line driver must be able to drive loads up to 300 pF. Thebuffer must have a unity gain and must be able to drive lowvoltages from about 0.5 V as well as high voltages up to 5 V.Therefore, a rail-to-rail opamp was designed. Normally, whendesigning a rail-to-rail opamp, it is important to keep the totaltransconductance variation (when the input varies over the fullscale) small. In this application, however, the opamp has unityfeedback, so that the open-loop variation is divided by the loopgain. This means that if the gain is made large enough, it isnot necessary to add a special structure to keep thevari-ation small. For this purpose, the amplification was chosen tobe higher than 60 dB. Large transistors are used where appro-priate to provide low offsets. This limits the frequency rangeof the opamp but, as mentioned earlier, this is not a problemhere, because the read-out software and not the hardware is themain speed-limiting factor. The transistor schematic of the linedriver design is given in Fig. 11. The current of the two comple-mentary input-differential pairs is summed and amplified. Theoutput stage is a modified Miller-compensated opamp stage [8].The two diodes, located next to the compensation capacitance

, shift the positive zero, that is common in two-stage Milleramplifiers, to higher frequencies. In simulation, the driver hasa gain of 61.6 dB and a GBW of 300 kHz for a load of 300 pFand a power consumption of 5.46 mW. The driver was also in-tegrated separately as a test structure, and in Fig. 12 the resultof an input voltage sweep is given. The total dynamic linearityerror or offset variation is less than 15 mV. Only at the two ex-tremities is a little increase seen. In these regions of operation,the open-loop gain reduces because the output transistors workin their linear region, reducing the gain. On a 5-V scale, thismeans that the absolute accuracy equals

dB bits (6)

Fig. 12. Offset (or total nonlinearity) of the rail-to-rail line driver.

Fig. 13. Sensor chip die microphotograph.

However, locally for one signal, the absolute accuracy can behigher. For example, if the absolute temperature information isconsidered, the voltage range is always between 0.5 and 0.6 V(throughout different batches and runs). In Fig. 12, it can be seenthat the difference between input and output stays smaller than1 mV for an input from 0.4 to 0.7 V. Thus, also for the tem-perature information range, the difference stays well within this1-mV accuracy (or 12 bits) and not within a 15-mV accuracythat is relevant for full-swing signals only. Once above 0.7-Vinput voltage, there is a shift in offset due to the nMOS differ-ential pair that starts to operate in its active region. Signals thatneed a higher accuracy than 15 mV and with an expected outputvoltage range that contains this transition region must be com-pensated.

VII. CHEMICAL MEASUREMENTS

The chip microphotograph is shown in Fig. 13. First, anamperometric oxygen measurement performed on two differentpackaged sensors is presented. For the measurement, the flowchannel is connected to an artificial patient, enabling a solutionwith regulated oxygen concentration to be pumped over thesensor array. The amperometric measurement is done usingthe potentiostatic setup (IV-OTA and Pot-OTA). Throughthe and O inputs, the working electrode is set at avoltage of 0.6 V versus the reference electrode. Then theoxygen concentration in the fluid is varied. Measurementresults are plotted in Fig. 14. Clearly, a linear relation of about


Fig. 14. Amperometric measurement of oxygen.

Fig. 15. Measured output voltage of a fully packagedK sensor.

25 mV/10%[O ] full scale is observed between the oxygenconcentration and the output voltage of the IV-OTA. Thedifference in slope between both measurements is explainedby slight differences in the selective membranes deposited ontop of the sensor (Clarck-type Osensor). Those membranesare deposited manually on the prototype devices, which makesit a difficult to reproduce process. A difference in thickness ofthe membrane can easily induce such large differences in theoutput slope because of the high amplification factor of theIV-OTA.

Next, the measurement of the potassium concentration usingan ISFET with a potassium-selective membrane is described. InFig. 15, the output of the ISFET buffer connected directly to aplotter (bypassing the multiplexer) is shown. Hence, the absenceof an absolute voltage level in this figure. It can clearly be seenthat the output voltage follows the potassium concentration ac-cording to

(7)

This is close to the theoretical slope of 59 mV. Also, the responsetime is fairly rapid, typically within one minute.

Finally, a conductometric measurement result is presented inFig. 16. The measured resistance versus specific resistance of

Fig. 16. Measured resistance versus specific resistance of buffer solution.

the buffer fluid is plotted. The input is a 1-kHz signal with a100-mV amplitude and the feedback resistance of the IV-OTAis 100 k . The resistance is calculated using

(8)

In Fig. 16, the output voltage amplitude thus varies between 143and 400 mV, which is in the expected range. For example, fora fluid with a 100- cm specific resistance, the theoreticalvalue is

specresistancedistance cmarea electrodecm

k

(9)

The difference with the measured value in Fig. 16 can be readilyexpected for a chemical sensor as a result of the condition ofthe electrodes (oxidation, nonideal etching) and the nonideallyparallel layout.

VIII. C ONCLUSION

A fully integrated multiparameter sensor chip has been pre-sented. Strong points of the system chip include low pin count,on-chip integrated ISFETs and ISFET buffers, temperaturecontrol, and built-in security EPROM. Also, a conductometricsensor and oxygen sensor are integrated on-chip with their in-terfacing electronics. In total, the same microsystem allows upto seven different measurements. Experimental measurementresults have been presented that prove the functionality of thesystem and the feasibility of the integration of multiple chem-ical sensors on one chip, including the interfacing electronics.

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Erik Lauwers was born in Leuven, Belgium, in1973. He received the M.Sc. degree in electricalengineering in 1997 from the Katholieke UniversiteitLeuven, Belgium. Since 1997, he has been workingtoward the Ph.D. degree as a Research Assistant atthe ESAT-MICAS Laboratories of the KatholiekeUniversiteit Leuven.

His research interests are mainly in mixed-signaland analog base-band integrated circuits.

Jan Sulsreceived the Master’s degree in chemistryin 1979 from the Katholieke Universiteit Leuven(KULeuven), Belgium. The work involved the mod-eling of the ground state of Co(II)N4 Schiffs Basecomplexes. Until 1983, he studied the photochemicalreaction mechanism of transition metal complexesafter pulsed laser excitation under an I.W.O.N.L.fellowship.

Since 1984, he has been a Delegate Scientistfrom IMEC at the KULeuven, ElectrotechnicalDepartment, ESAT. His interests are in the integra-

tion of biomembranes and biorecognition systems in planar chemical sensorapplications.

Walter Gumbrecht received the Ph.D. degree inphysical chemistry in 1983 from the University ofErlangen, Nurnberg, Germany.

He joined the Corporate Technology Departmentof the Siemens AG, Erlangen, Germany, in 1984. Heis working on the development of semiconductor-based chemical and biochemical sensors and micro-systems for medical applications.

David Maes received the degree in electrical engi-neering from the University of Leuven, Belgium, in1996.

In 1996, he joined IMEC as a Process Engineer towork on the CMOS integration of chemical sensorsand of ferroelectric memories.

Georges Gielenwas born in Heist-op-den-Berg, Bel-gium. He received the M.Sc. and Ph.D. degrees inelectrical engineering from the Katholieke Univer-siteit Leuven, Belgium, in 1986 and 1990, respec-tively.

After being a Postdoctoral Research Assistantand Visiting Lecturer at the University of California,Berkeley, he returned to the Department of ElectricalEngineering of the Katholieke Universiteit Leuven,where he is currently a Professor. His researchinterests include design and computer-aided design

(simulation, modeling, synthesis, layout, and test) of analog and mixed-signalintegrated circuits. He serves regularly on the program committees of interna-tional conferences and is an associate editor for several journals.

Willy Sansen (M’72–SM’86–F’95) received theM.Sc. degree in electrical engineering from theKatholieke Universiteit Leuven (K. U. Leuven) in1967 and the Ph.D. degree in electronics from theUniversity of California, Berkeley, in 1972.

In 1969, he received a BAEF fellowship. In 1972he was appointed by the National Fund of ScientificResearch (Belgium) at the ESAT laboratory of theK. U. Leuven, where he has been a full Professorsince 1980. During 1984–1990, he was the head ofthe Electrical Engineering Department. Since 1984,

he has headed the ESAT-MICAS Laboratory on analog design, which includesabout fifty members and which is active in research projects with industry. Heis a member of several boards of directors. In 1978, he was a Visiting Professorat Stanford University, Stanford, CA, in 1981 at the EPFL Lausanne, in 1985at the University of Pennsylvania, Philadelphia, and in 1994 at the T. H. Ulm.He is cofounder and organizer of the workshops on Advances in Analog CircuitDesign in Europe. He has been involved in design automation and in numerousanalogue integrated circuit designs for telecom, consumer electronics, medicalapplications, and sensors, and has supervised over forty Ph.D. theses in thesefields. He is a member of several editorial and program committees of journalsand conferences. He has authored and coauthored eleven books and more than550 papers in international journals and conference proceedings. He is a memberof the executive and program committees of the IEEE ISSCC conference, andis the program chair of the ISSCC 2002 conference.

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