2012_60_02_P2

FEBRUARY 2012 VOLUME 60 NUMBER 2 IETPAK (ISSN 0018-926X)

PART II OF TWO PARTS

PAPERS

Antennas and ResonatorsA Broadband VHF/UHF Double-Whip Antenna . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . X. Ding, B.-Z. Wang, G.-D. Ge, and D. Wang 719Wideband Dielectrically Guided Horn Antenna with Microstrip Line to H-Guide Feed . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Wong, A. R. Sebak, and T. A. Denidni 725

CPW-Fed Cavity-Backed Slot Radiator Loaded With an AMC Reflector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Joubert, J. C. Vardaxoglou, W. G. Whittow, and J. W. Odendaal 735

The Use of Simple Thin Partially Reflective Surfaces With Positive Reflection Phase Gradients to Design Wideband,Low-Profile EBG Resonator Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. Ge, K. P. Esselle, and T. S. Bird 743

Omnidirectional Linearly and Circularly Polarized Rectangular Dielectric Resonator Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. M. Pan, K. W. Leung, and K. Lu 751

Substrate Integrated Composite Right-/Left-Handed Leaky-Wave Structure for Polarization-Flexible AntennaApplication . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. Dong and T. Itoh 760

Design and Characterization of Miniaturized Patch Antennas Loaded With Complementary Split-Ring Resonators . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. Dong, H. Toyao, and T. Itoh 772

Dual-Band Circularly Polarized Microstrip RFID Reader Antenna Using Metamaterial Branch-Line Coupler . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y.-K. Jung and B. Lee 786

Small-Size Shielded Metallic Stacked FabryPerot Cavity Antennas With Large Bandwidth for Space Applications . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. A. Muhammad, R. Sauleau, and H. Legay 792

A Simple Technique for the Dispersion Analysis of Fabry-Perot Cavity Leaky-Wave Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . C. Mateo-Segura, M. Garca-Vigueras, G. Goussetis, A. P. Feresidis, and J. L. Gmez-Tornero 803

Analyzing the Complexity and Reliability of Switch-Frequency-Reconfigurable Antennas Using Graph Models . . . . . . . . .. . . . . . . . . . . . . . . . J. Costantine, Y. Tawk, C. G. Christodoulou, J. C. Lyke, F. De Flaviis, A. Grau Besoli, and S. E. Barbin 811

Free Space Radiation Pattern Reconstruction from Non-Anechoic Measurements Using an Impulse Response of theEnvironment . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . J. Koh, A. De, T. K. Sarkar, H. Moon, W. Zhao, and M. Salazar-Palma 821

Electric Field Amplification inside a Porous Spherical Cavity Resonator Excited by an External Plane Wave . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P. A. Bernhardt and R. F. Fernsler 832

(Contents Continued on p. 717)

(Contents Continued from Front Cover)

ArraysA 76 GHz Multi-Layered Phased Array Antenna Using a Non-Metal Contact Metamaterial Waveguide . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . H. Kirino and K. Ogawa 840

Beam Switching Reflectarray Monolithically Integrated With RF MEMS Switches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . O. Bayraktar, O. A. Civi, and T. Akin 854

Design and Implementation of a Closed Cylindrical BFN-Fed Circular Array Antenna for Multiple-Beam Coverage inAzimuth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . N. J. G. Fonseca 863

Rapidly Convergent Representations for Periodic Greens Functions of a Linear Array in Layered Media . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. Van Orden and V. Lomakin 870

A Novel Strategy for the Diagnosis of Arbitrary Geometries Large Arrays . . . . . . . . .. . . . . . . . . A. Buonanno and M. DUrso 880Predicting Sparse Array Performance From Two-Element Interferometer Data . . . . . . . . . . . . . . J. A. Nessel and R. J. Acosta 886Linear Aperiodic Array Synthesis Using an Improved Genetic Algorithm . . . . . . . . . . L. Cen, Z. L. Yu, W. Ser, and W. Cen 895Beamformer Design Methods for Radio Astronomical Phased Array Feeds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Elmer, B. D. Jeffs, K. F. Warnick, J. R. Fisher, and R. D. Norrod 903

Experimental Results for the Sensitivity of a Low Noise Aperture Array Tile for the SKA .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . E. E. M. Woestenburg, L. Bakker, and M. V. Ivashina 915

Direction Finding With Partly Calibrated Uniform Linear Arrays . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . B. Liao and S. C. Chan 922Numerical and Inverse TechniquesCalculation of MoM Interaction Integrals in Highly Conductive Media . . . . .. . . . . J. Peeters, I. Bogaert, and D. De Zutter 930Electromagnetic Scattering From General Bi-Isotropic Objects Using Time-Domain Integral Equations Combined WithPMCHWT Formulations . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . Z.-H. Wu, E. K.-N. Yung, D.-X. Wang, and J. Bao 941

Efficient Surface Integral Equation Using Hierarchical Vector Bases for Complex EM Scattering Problems . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . L. P. Zha, Y. Q. Hu, and T. Su 952

Accelerated FDTD Analysis of Antennas Loaded by Electric Circuits . . . . . . . . . . . . . . . . . . . . . . . Y. Watanabe and H. Igarashi 958An Angle-Dependent Impedance Boundary Condition for the Split-Step Parabolic Equation Method . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . C. R. Sprouse and R. S. Awadallah 964

A Nested Multi-Scaling Inexact-Newton Iterative Approach for Microwave Imaging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. Oliveri, L. Lizzi, M. Pastorino, and A. Massa 971

Fast and Shadow Region 3-Dimensional Imaging Algorithm With Range Derivative of Doubly Scattered Signals forUWB Radars . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. Kidera and T. Kirimoto 984

High-Resolution ISAR Imaging by Exploiting Sparse Apertures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . L. Zhang, Z.-J. Qiao, M.-D. Xing, J.-L. Sheng, R. Guo, and Z. Bao 997

NondestructiveMaterial Characterization of a Free-Space-BackedMagnetic Material Using a Dual-Waveguide Probe . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. W. Hyde, M. J. Havrilla, A. E. Bogle, and E. J. Rothwell 1009

EvaporationDuctHeight Estimation and SourceLocalization FromFieldMeasurements at anArray ofRadioReceivers . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . X. Zhao 1020

Extrapolation of Wideband Electromagnetic Response Using Sparse Representation . . . . . . . . . . . . . . H. Zhao and Y. Zhang 1026WirelessA Wearable Two-Antenna System on a Life Jacket for Cospas-Sarsat Personal Locator Beacons . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. A. Serra, P. Nepa, and G. Manara 1035

Analysis of Cellular Antennas for Hearing-Aid Compatible Mobile Phones . . . . . . .. . . . . . . P. M. T. Ikonen and K. R. Boyle 1043A Mobile Communication Base Station Antenna Using a Genetic Algorithm Based Fabry-Prot ResonanceOptimization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. Kim, J. Ju, and J. Choi 1053

Development of Novel 3-D Cube Antennas for Compact Wireless Sensor Nodes . . . . . . . . . . I. T. Nassar and T. M. Weller 1059Influence of the Hand on the Specific Absorption Rate in the Head . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . C.-H. Li, M. Douglas, E. Ofli, B. Derat, S. Gabriel, N. Chavannes, and N. Kuster 1066

Demonstration of a Cognitive Radio Front End Using an Optically Pumped Reconfigurable Antenna System (OPRAS) . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. Tawk, J. Costantine, S. Hemmady, G. Balakrishnan, K. Avery, and C. G. Christodoulou 1075

Evaluation of a Statistical Model for the Characterization of Multipath Affecting Mobile Terminal GPS Antennas inSub-Urban Areas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Ur Rehman, X. Chen, C. G. Parini, and Z. Ying 1084

A Mixed RaysModes Approach to the Propagation in Real Road and Railway Tunnels . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . F. Fuschini and G. Falciasecca 1095

Optimum Wireless Powering of Sensors Embedded in Concrete . . . . . . . . . . . .. . . . . . . . . . . . S. Jiang and S. V. Georgakopoulos 1106Portable Real-Time Microwave Camera at 24 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. T. Ghasr, M. A. Abou-Khousa, S. Kharkovsky, R. Zoughi, and D. Pommerenke 1114

Is Orbital Angular Momentum (OAM) Based Radio Communication an Unexploited Area? . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . O. Edfors and A. J. Johansson 1126

(Contents Continued on p. 718)

(Contents Continued from p. 717)

COMMUNICATIONS

A Circularly Polarized Ring-Antenna Fed by a Serially Coupled Square Slot-Ring . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . T.-N. Chang, J.-M. Lin, and Y. G. Chen 1132

A Pseudo-Normal-Mode Helical Antenna for Use With Deeply Implanted Wireless Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . O. H. Murphy, C. N. McLeod, M. Navaratnarajah, M. Yacoub, and C. Toumazou 1135

A Novel Folded UWB Antenna for Wireless Body Area Network . . . . . . . . . . . . . . . . . C.-H. Kang, S.-J. Wu, and J.-H. Tarng 1139Hybrid Mode Wideband Patch Antenna Loaded With a Planar Metamaterial Unit Cell . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Ha, K. Kwon, Y. Lee, and J. Choi 1143

Explicit Relation Between Volume and Lower Bound for Q for Small Dipole Topologies . . . . . . . . . G. A. E. Vandenbosch 1147On the Generalization of Taylor and Bayliss n-bar Array Distributions . . . . . . . . . . . . . .. . . . . . . . . . . . . . S. R. Zinka and J. P. Kim 1152Amplitude-Only Low Sidelobe Synthesis for Large Thinned Circular Array Antennas . . . . . . . . . . . . . . . . W. P. M. N. Keizer 1157Power Synthesis for Reconfigurable Arrays by Phase-Only Control With Simultaneous Dynamic Range Ratio andNear-Field Reduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. Buttazzoni and R. Vescovo 1161

Design and Experiment of a Single-Feed Quad-Beam Reflectarray Antenna . . . . P. Nayeri, F. Yang, and A. Z. Elsherbeni 1166ObliqueDiffraction ofArbitrarily PolarizedWaves by anArray of Coplanar Slots Loaded byDielectric Semi-Cylinders . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. L. Tsalamengas and I. O. Vardiambasis 1171

Analysis of Radiation Characteristics of Conformal Microstrip Arrays Using Adaptive Integral Method . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . W.-J. Zhao, L.-W. Li, E.-P. Li, and K. Xiao 1176

Generalized Multilevel Physical Optics (MLPO) for Comprehensive Analysis of Reflector Antennas . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . C. Letrou and A. Boag 1182

Fast Dipole Method for Electromagnetic Scattering From Perfect Electric Conducting Targets . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . X. Chen, C. Gu, Z. Niu, and Z. Li 1186

An Efficient Hybrid GO-PWS Algorithm to Analyze Conformal Serrated-Edge Reflectors for Millimeter-Wave CompactRange . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. Muoz-Acevedo and M. Sierra-Castaer 1192

Time-Domain Microwave Imaging of Inhomogeneous Debye Dispersive Scatterers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . T. G. Papadopoulos and I. T. Rekanos 1197

CALL FOR PAPERS

Call for Papers: Special Issue on Antennas and Propagation at Millimeter and Sub-millimeter Waves . . . . . . . . . . . . . . . . . . . . 1203

IEEE ANTENNAS AND PROPAGATION SOCIETYAll members of the IEEE are eligible for membership in the Antennas and Propagation Society and will receive on-line access to this TRANSACTIONS through IEEE Xplore upon payment of the annual Societymembership fee of $24.00. Print subscriptions to this TRANSACTIONS are available to Society members for an additional fee of $36.00. For information on joining, write to the IEEE at the address below.Member copies of Transactions/Journals are for personal use only.

ADMINISTRATIVE COMMITTEES. R. BEST, President T. S. BIRD, President Elect J. S. TYO, Secretary-Treasurer

2012 2013 2014 2015Y. M. M. ANTARJ. T. BERNHARD*S. MACIM. MARTINEZ-VZQUEZD. F. SIEVENPIPER

M. ANDO*D. B. DAVIDSONM. EL-SHENAWEES. K. RAOM. W. SHIELDS

R. D. NEVELS*D. ERRICOLOL. JOFREC. PICHOTN. RILEY

M. SALAZAR PALMA*

Honorary Life Members: R. C. HANSEN, W. R. STONE*Past President

Committee Chairs and RepresentativesAntenna Measurements (AMTA): S. SCHNEIDERAntennas & Wireless Propagation Letters Editor-in-Chief:

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Digital Object Identifier 10.1109/TAP.2012.2186330

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 2, FEBRUARY 2012 719

A Broadband VHF/UHF Double-Whip AntennaXiao Ding, Bing-Zhong Wang, Member, IEEE, Guang-Ding Ge, and Duo Wang

AbstractThis paper presents a broadband VHF/UHFdouble-whip antenna with one lossless matching scheme com-bining two methods, embedded transmission line matchingmethod and lumped-distributed hybrid matching method. Byadjusting the length of the embedded transmission line, the com-bination of double-whip antenna and the transmission line canachieve resonance, thus realize a coarse matching. By adding alumped-distributed hybrid matching network at the feeding pointof the double-whip antenna, we can further improve the matchingfor the double-whip antenna. Moreover, based on the two-stepmatching scheme, a double-whip antenna has been designedand fabricated. Measured results show that, the VSWRs of thedouble-whip antenna, with the electrical lengths of and

at the minimum operation frequency respectively, are lessthan 2 over a 17:1 octave bandwidth, and the horizontal gains ofthe antenna are between 4.2 dB and 6.8 dB. Thanks to its highgain, broadband and low reflection, the proposed double-whipantenna in this paper is ideal for application in vehicle wirelesscommunication.

Index TermsBroadband matching network, double-whip an-tenna, VHF/UHF antenna.

I. INTRODUCTION

F EATURING of its characteristics like small size, simplestructure and omni-direction, whip antenna have beenwidely used in ultra-short wave, shortwave, and VHF/UHFwireless communication. Actually, one distinctive character-istic of whip antenna lies in its very small radiation resistanceand very large negative reactance in low frequency. Con-sequently, it gains a large Q factor and a narrow workingbandwidth, and causes most of the energy difficult to radiateand only oscillating around the antenna instead. Under thiscircumstance, direct feeding to the antenna from feed linewould make the receiver or transmitter in system front-endfail or even breakdown because of large reflection. To solvethis practical problem of over-reflection, or mismatch indeed,many methods have been discussed by scholars. One proposaluses lumped or distributed load on antenna to improve thecurrent distribution on the surface of the antenna. In [1], a LRloaded wire monopole with a matching network achieves a 20:1bandwidth, VSWR less than 3.0 and system gain greater than

Manuscript received March 02, 2011; revised July 07, 2011; accepted Au-gust 26, 2011. Date of publication October 21, 2011; date of current versionFebruary 03, 2012. This work was supported in part by the High-Tech Re-search and Development Program of China (No. 2008AA01Z206), in part bythe Research Fund for the Doctoral Program of Higher Education of China (No.20100185110021), and in part by the National Natural Science Foundation ofChina (No. 61071031), and Project 9140A01020110DZ0211.The authors are with the Institute of Applied Physics, University of Elec-

tronic and Science Technology of China, Chengdu 610054, China (e-mail:[email protected]; [email protected]).Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TAP.2011.2173141

dBi. Another idea takes embedded broadband matchingnetwork consisting of some lumped components to eliminatethe imaginary part of the antenna impedance. In [2], a 2-meterbroadband whip antenna with electronically switching threedifferent matching networks could operate over the frequencyrange of 2360 MHz with VSWR less than 3.5 and systemgain greater than dB. Reference [3] simultaneously usesload technology and on-body matching network to realize aVHF/UHF whip antenna with VSWR less than 2 and systemgain greater than 0 dBi. A third one utilizes fractal technique forthe improvement of the impedance characteristics of a varietyof VHF/UHF antennas [4]. And in recent years, the emergencyof the study on metamaterial has inspired some scholars todesign VHF/UHF metamaterials antennas [5]. Moreover, [6]matches a conical antenna with the aid of transmission line.By adding a section of the transmission line to form a resonantstructure with the conical antenna, more efficient operationat low frequencies is obtained. However, all above methodshave more or less specific deficiencies in implementation. Forexample, the load will inevitably reduce the antenna radiationefficiency and its structural strength; the broadband matchingnetworks of lumped components would sacrifice system gainfor bandwidth matching; and the cost of the metamaterials isusually far too high.In order to get higher gain, wider bandwidth and lower reflec-

tion, this paper researches a VHF/UHF double-whip antennaand two lossless matching methods. Two individual whips,which are connected by two sections of transmission lines asa double-whip antenna, work in the upper and lower bands of30520 MHz, respectively. By adjusting the length of the em-bedded transmission lines, the whip and the transmission linecould get resonant, and the antenna can approximately meet theengineering requirement for impedance matching. This methodprovides a coarse impedance match for the double-whip an-tenna. In order to further match the double-whip antenna, twomatching networks are designed respectively. A traditionallylumped matching networks and a new lumped-distributedhybrid matching network are added at the feeding point of theantenna separately. These two matching networks working overdifferent frequency bands can get a higher gain and better im-pendence matching for the double-whip antenna. The designeddouble-whip antenna with those two lossless matching methodshas a good performance in VHF/UHF frequency bands.

II. COARSE IMPEDANCE MATCH OF THE DOUBLE-WHIPANTENNA

A. Double-Whip AntennaThe structure of the double-whip antenna is shown in

Fig. 1(a). It consists of two whip antennas with heightsm and m respectively, and their diameters

0018-926X/$26.00 2011 IEEE

720 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 2, FEBRUARY 2012

Fig. 1. (a) Double-whip antenna. (b) Equivalent circuit of (a).

are both 2.5 cm. Two embedded transmission lines with lengthsand and diameters and are used to connect these

whips and the distances between the embedded lines and theground plane are and , respectively. The joint point of thetwo embedded lines is connected to two matching networksthrough an electrical switch. At the other ports of the matchingnetworks is the feeding point. Fig. 1(b) shows the equivalentcircuits of the double-whip antenna in Fig. 1(a). is theinput impedance of the whip antenna of height and thewhip of is the input impedance at the joint point, and

and are the characteristic impedances of the embeddedtransmission lines of lengths and , which can be calcu-lated by the following formula of parallel-wire conductor line

(1)

The double-whip antenna can be viewed as two parallel ele-ments connected at the joint point in Fig. 1(a), where Element1 consists of a whip antenna with the height and one em-bedded transmission line with the length ; Element 2 consistsof a whip antenna with the height and one embedded trans-mission line with the length .

Fig. 2. (a) Element 2. (b) Equivalent circuits of (a).

B. Elements DesignIn the following theoretic analysis, we take Element 2 as an

example, and Fig. 2 depicts its configuration.From Fig. 2(b), the input impedance of Element 2 at the joint

port is

(2)

where isthe reflection coefficient at the port of the whip with length .For the purpose of matching, must be real in (2) and

the exponent must satisfy the condition ,( .). So the solutions to is given by

(3)

Because the electrical length of the embedded transmission linecan not be negative, and the periodicity of the with ,

so one solution to is

(4)

Because the input impendence varies with fre-quency, from (4) we can notice that: for any whip antenna, inorder to match it with the aid of the embedded transmissionline, the electrical length of the embedded line shouldbe variable as the working frequency changes. However, as amatter of fact, the length of the embedded line must be fixedin engineering design. In other word, for a given length andcharacteristic impedance of an embedded transmission line, itcan only match the whip antenna in a narrow band. In order togive attention to broadband matching, we can take an averagevalue of over the whole matching bandwidth as the lengthof the embedded transmission line, so that we can obtain anapproximate matching for the element.Fig. 3 gives the input impedance of the whip antennam, whose first resonance frequency is at 250 MHz. Fig. 3

can give more apparent illustration about the impedance varia-tion in broadband range: from the first resonance frequency, asfrequency increases, the fluctuation of the real part of the inputimpedance almost becomes moderate in a certain range of 29

while its imaginary part approaches zero. Accordingly,we set the characteristic impendence of the transmissionline to be 29 , and take the result of into (3). Thenwe could calculate the theoretic solution to the transmission line

DING et al.: A BROADBAND VHF/UHF DOUBLE-WHIP ANTENNA 721

Fig. 3. 0.3 m whip antenna impedance .

Fig. 4. Comparison of reflection coefficients before and after embedding thetransmission line for element 2.

electrical length after taking the average value processingover the bandwidth, which is . In the practically en-gineering design, we determine the physical length at thecenter frequency MHz and get cm from

. Further, if the radius of embedded line is set ascm, we can work out the distance between the con-

nection embedded line and the ground from (1) ascm. Fig. 4 shows the reflection coefficients before and after em-bedding this embedded transmission line between the whip anda 50 feeding line. As shown in the solid line with circular, inthe frequency range of 180410 MHz, VSWR is less than 3 andin the frequency range of 410520MHz, VSWR is less than 3.5.And when the frequency is below 150 MHz, Element 2 wouldreflect all the feeding current. In order to make the double-whipantenna work below 150 MHz, the design of Element 1 shouldgive main concerns to work over the lower frequency band.Fig. 5 gives the input impedance of the whip antennam, whose first resonance frequency is at 46.8 MHz. In the

frequency range of 30150 MHz, the real part of the input resis-tance shakes around ohms from 30 to 55 MHz and from100 to 140 MHz, but changes acutely between 55 MHz and 100MHz. In order to get a fixed physical length of embedded linefrom (3) and considering the whole bands coarse matching, we

Fig. 5. 1.6 m whip antenna impedance .

Fig. 6. Comparison of reflection coefficients before and after embedding thetransmission line for element 1.

take 42 ohms as the characteristic impedance of the embeddedline, i.e., ohms. Then we calculate (3) and take theaverage value of . In the practically engineeringdesign, we determine the physical length at the center fre-quency MHz and get m from .Fig. 6 shows the reflection coefficients before and after embed-ding the transmission line between the whip and a 50 feedingline. From Fig. 6, after embedding the matching transmissionline, the VSWR2:1 bandwidth increases and the first resonancefrequency decreases to about 30 MHz.

C. Double-Whip Design

From Fig. 4, we know that Element 2 would reflect all thefeeding current as the frequency below 150 MHz and partiallyreflect feeding current as the frequency above 150 MHz. FromFig. 6 the first resonance frequency of Element 1 has decreasedto about 30 MHz. Based on the analysis in the above elementdesign, we parallel connect the two elements in the arrange-ment as shown in Fig. 1(a), and the actual antenna can be ob-tained as shown in Fig. 7. In Fig. 7, two embedded transmissionlines with black and white coating respectively are shown onthe top of the ground and two ivory-white plastic fixed parts atthe bottom of the monopole are used to connect the quadratemetal pedestals which are below the ground. Finally, we put the


Fig. 7. Double-whip antenna.

Fig. 8. Comparison of measure reflection coefficients before and after thematching transmission line.

latter two matching networks into the quadrate metal pedestal.At the joint point of the two embedded transmission line, whenlower frequency signal is coming, it will be reflected by Ele-ment 2 and flow to Element 1 primarily. On the other hand,Element 1 will reflect higher frequency signal and Element 2will pass higher frequency signal mainly. Fig. 8 shows the com-parison of measured reflection coefficients, in which the dottedline stands for the whip antenna of 1.6 m-height without the em-bedded transmission line, the triangular-solid-line for the whipantenna of 0.3 m-height without the embedded transmissionline and the solid line is for the double-whip antenna with twoembedded matching transmission lines. From the comparison,we can see that by simply using lossless transmission lines forcoarse matching, more than 50% of the frequencies sampledover the band of 30520 MHz have VSWRs less than 2, andmore than 75% of the frequencies have VSWRs less than 3.At present, the double-whip antenna with two lossless

embedded transmission lines can work from 30 MHz to 520MHz. But as shown in Fig. 8, the matching results cannotyet meet practical project needs, such as , espe-cially at the lower frequencies. In order to further improve theimpedance matching performance of the double-whip antenna,two lumped-distributed hybrid matching networks are added

Fig. 9. Absolute vale of a 47 pF monolithic ceramic capacitor impedance as afunction of frequency.

below the joint port, as shown in Fig. 1(a) and analyzed in thenext section.

III. FURTHER IMPROVEMENT ON BROADBAND MATCHINGTraditional method of broadband matching often loads with

lumped components and treats them independent of frequency.So this method could match the real part of the impedance to50 and the imaginary part to zero with several LC matchingnetworks. However, this traditional method ignores the fre-quency variability of lumped components, which does exist inpractical design and would cause a lower gain and a narrowermatching bandwidth. Reference [7] announces this phenom-enon. Example 14 in [7, p. 19] shows that the capacitancefrom a real capacitor strictly obeys the rule in thefrequency range from DC to several hundred MHz. However,when frequency increases further, the capacitance would notobey . And for a real inductor in Example 15in [7, p. 25], the situation is the same. A 47 pF monolithicceramic capacitor which is made in China and used in ourdesign, is measured by a Network Analyzer E5071C to obtainits capacitor impedance. Fig. 9 gives the absolute vale of theimpedance of this capacitor as a function of frequency. InFig. 9, when the frequency increases to about 100 MHz, thetest capacitance will not obey .Considering the above characteristics of lumped and dis-

tributed electronic components at VHF/UHF band, traditionalmatching method would not be effective at the whole operationband. So, we design two matching networks in this paper whichwork at 30120 MHz and 120520 MHz respectively. For the30120 MHz matching network, the impedance characteristicof the lumped components has not changed yet, shown as inFig. 9. A traditional lumped matching network is implemented.And in the frequency range of 120520 MHz, an effectivematching method of lumped-distributed hybrid matching isimplemented. The topology structures of the two matchingnetworks are presented in Fig. 10. Fig. 10(a) is the broadbandlumped matching network for the range of 30120 MHz, whichshares the same principle as traditional matching networks.Fig. 10(b) gives the proposed lumped-distributed matching

DING et al.: A BROADBAND VHF/UHF DOUBLE-WHIP ANTENNA 723

Fig. 10. Topology structure of matching network, (a) 30120 MHz broadbandmatching network, (b) 120520 MHz broadband matching network.

PCB Layout for the range of 120520 MHz. It consists of sixparts: and are distributed capacitors; is a third-orderladder impedance converter, which, together with , forms oneparallel multi-level impedance matching circuit. This matchingcircuit, functioning as the multi-level damping network oflumped components, can assuage the fierce fluctuation of an-tenna impedance in low frequency; is one LC low-frequencyfilter circuit which can filter out part of the input signal withfrequency lower than 120 MHz; is the blocking capacitors,and is the 50 matching line for input and output ports.The lumped-distributed hybrid matching approach owns atleast following advantages: (1) lossless matching to improvethe system efficiency; (2) distributed components are used toachieve a broadband matching because their characteristicswould not vary with frequency; (3) simple matching networkstructure can avoid electromagnetic compatibility problemsfrom complicated lumped matching circuit; and (4) planarmicrostrip line structure of matching networks is easy forconformal design.Both matching networks are printed on microstrip board

with dielectric constant 2.2 and thickness 1 mm. Table I showsthe values of the lumped electronic components in 30120MHz matching network and Table II shows the dimensionsand values of the distributed electronic components in 120520MHz lumped-distributed matching network.The final measurement results for the antenna structure are

presented in Figs. 1113. The reflection coefficients at thefeeding port are depicted in Fig. 11. After further matchingwith lumped-distributed hybrid network, the reflection coef-ficients over the whole frequency bandwidth are generally lessthan dB . Finally, we give the test results ofhorizontal gain of the double-whip antenna shown in Fig. 12,and the measured radiation patterns in E-plane and H-plane atfrequencies of 45, 170, 330, 500 MHz shown in Fig. 13.

TABLE IVALUES OF THE ELEMENTS IN 30120 MHZ MATCHING NETWORK

TABLE IIVALUES OF THE ELEMENTS IN 120520 MHZ MATCHING NETWORKS

Fig. 11. Measured reflection coefficients of the double-whip antenna, (a)30120 MHz, (b) 120520 MHz.

IV. CONCLUSION

This paper proposes a VHF/UHF double-whip antenna withtwo lossless matching methods. This antenna system has meritsas follows. First, it has a very small size, which is convenientfor vehicular concealment. The electrical lengths of the twowhips are and at the minimum operation frequency,respectively. Second, it has a very wide operation band that


Fig. 12. Measured horizon gains of the double-whip antenna.

Fig. 13. Measured radiation patterns of the double-whip antenna at differentfrequencies, (a) 45 MHz, (b) 170 MHz, (c) 330 MHz, (d) 500 MHz.

covers VHF and UHF band from 30 to 520 MHz with mea-sured VSWRs less than 2. Third, compared to other ways oflossy matching, our lossless matching methods result in rela-tively higher horizontal gains, which are between 4.2 to 6.8 dBiin the operation band.This designed antenna can be widely used in vehicular, ship-

board and civil mobile communication with high gain, wideband and low reflection.

REFERENCES[1] S. D. Rogers, C.M. Butler, andA. Q.Martin, Design and realization of

GA-optimized wire monopole and matching network with 20:1 band-width, IEEE Trans. Antennas Propag., vol. 51, no. 3, pp. 493502,Mar. 2003.

[2] K. Yegin and A. Q. Martin, Very broadband loaded monopole an-tennas, in Proc. IEEE Antennas and Propag. Soc. Int. Symp., Mon-treal, QC, Canada, Jul. 1997, vol. 1, pp. 232235.

[3] X. Ding, B.-Z. Wang, G. Zheng, and X.-M. Li, Design and realiza-tion of a GA-optimized VHF/UHF antenna with on-body matchingnetwork, IEEE Antenna Wireless Propag. Lett., vol. 9, pp. 303307,2010.

[4] J. M. Gonzlez-Arbes, S. Blanch, and J. Romeu, Are space fillingcurves efficient small antennas, IEEE Antennas Wireless Propag.Lett., vol. 2, pp. 147150, 2003.

[5] H. Lizuka and P. S. Hall, Left-handed dipole antennas and theirimplementations, IEEE Trans. Antennas Propag., vol. 55, no. 5, pp.12461253, 2007.

[6] S. Sheldon and W. P. K. Ronold, Compact conical antenna for wide-band coverage, IEEE Trans. Antennas Propag., vol. 42, no. 3, pp.436439, 1994.

[7] L. Reinhold and B. Pavel, RF Circuit Design: Theory and Applica-tions. Englewood Cliffs, NJ: Prentice Hall, 2000.

Xiao Ding was born in Sichuan Province, China,May, 1982. He received the B.S. and M.S. degreesin communication engineering and electromagneticfield and microwave engineering, respectively,from Guilin University of Electronic Science andTechnology (GUET), China. He is currently workingtoward the Ph.D. degree at the University of Elec-tronic Science and Technology of China (UESTC),Chengdu, since 2009.His research interests include short-wave, ul-

trashort-wave wire antennas and its broadbandmatching technology, millimeter-wave antenna and phased array.

Bing-Zhong Wang (M06) received the Ph.D.degree in electrical engineering from the Universityof Electronic Science and Technology of China(UESTC), Chengdu, China, in 1988.He joined the UESTC in 1984 where he is cur-

rently a Professor. He has been a Visiting Scholar atthe University of Wisconsin-Milwaukee, a ResearchFellow at the City University of Hong Kong, and aVisiting Professor in the Electromagnetic Commu-nication Laboratory, Pennsylvania State University,University Park. His current research interests are

in the areas of computational electromagnetics, antenna theory and technique,electromagnetic compatibility analysis, and computer-aided design for passivemicrowave integrated circuits.

Guang-Ding Ge is currently working toward thePh.D. degree in the Institute of Applied Physics,University of Electronic Science and Technology ofChina, Chengdu, China.His main research interests include microwave cir-

cuits, antenna theory and design, time reversal tech-nique, compact and wideband antennas and arrays forwireless communications systems.

Duo Wang was born in 1986 in Chongqing, China.He received the B.S. and M.S. degrees from theUniversity of Electronic Science and Technologyof China, in 2008 and 2011, respectively. He iscurrently pursuing the Ph.D. degree in the InstitutNational des Sciences Appliques de Rennes,France.His current research interests include the technique

of Electromagnetic Time Reversal and the design ofmicrowave antenna.


Wideband Dielectrically Guided Horn Antenna withMicrostrip Line to H-Guide Feed

Michael Wong, Member, IEEE, Abdel Razik Sebak, Fellow, IEEE, and Tayeb A. Denidni, Senior Member, IEEE

AbstractThe design, simulation, and measurement of a com-plete microstrip line-fed dielectrically guided horn antenna arepresented. The proposed antenna achieves similarly high gains ascompared to traditional air-filled horn antennas, is simpler thana typical array design, and can easily be fabricated using typicaltwo dimensional substrate machining processes. An H-guide,operating in the fundamental

mode, slowly tapers into agapped H-guide, or dielectrically guided horn, where a large airgap separates the center dielectric and metallic plates. A widebandBzier shaped microstrip to H-guide transition feeding structureis fabricated using a low loss Rogers 5880 substrate and integratedwith the proposed antenna. The fabricated prototype operatesfrom 8 to 16 GHz with a peak gain of approximately 16 dBi.

Index TermsAperture antennas, dielectric waveguides, feeds,microstrip transitions, millimeter wave antennas.

I. INTRODUCTION

H ORN antennas appear in many different forms, such asdielectric-filled horn antennas [1], metamaterial-linedhorn antennas [2], corrugated circular horn antennas [3], oreven our recently proposed Substrate Integrated Waveguide(SIW) planar slot antenna [4]. Typically, metallic air-filledhorn antennas are formed using expensive machining processesto obtain the precise angles required for highly predictableantenna patterns. Gain standard horns are typically made in thisfashion using metallic walls on the top, bottom, and sides.

In the design proposed in this paper, however, waves aremostly guided by the dielectric near the mouth of the horn, oraperture, thus reducing the dependence upon the horn anglesand precise dimensions, and removing the need for metallicsidewalls. As is shown in this paper, such control can allow thedesign of a high gain wide bandwidth antenna, while reducingdiffracted fields by concentrating the fields away from metallicedges. Such reduction in diffracted fields immediately reducesspill-over radiation and consequently lowers the sidelobe levels.In addition, by using this dielectric wave guiding property, the

Manuscript received November 18, 2010; revised April 05, 2011; acceptedJuly 25, 2011. Date of publication October 21, 2011; date of current versionFebruary 03, 2012.

M. Wong is with Research in Motion (RIM), Kanata, ON, Canada (e-mail:[email protected]).

A. R. Sebak is with Concordia University in Montreal, Quebec and withPrince Sultan Advanced Technological Research Institute (PSATRI), King SaudUniversity, 11451 Riyadh, Saudi Arabia (e-mail: [email protected]).

T. A. Denidni is with INRS-EMT, Montreal, QC H5A-1K6, Canada (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2011.2173123

magnitude and the phase of the electric fields over the aperturecan be controlled. Neglecting fringing effects due to metallicedges, this control of the near field pattern allows some controlof the far field radiation pattern.

To achieve these very desirable properties, a wide bandwidth,single mode, thin H-guide is used as a feed mechanism to ex-cite the horn, which is in turn fed by a microstrip to H-guidetransition. We have briefly discussed the H-guide horn antennaconcept in a conference paper [5] for a slightly different design.While the H-guide has existed for many years [6], it is only re-cently that the miniaturization of such dielectric waveguides, forthe example, in the form of the Non-Radiative Dielectric (NRD)waveguide [7], or the Dielectric Image Guide (DIG) [8] has be-come of interest. In addition, the excitation of waves within sub-strates has recently been proposed in the form of surface wavelaunchers [9], SIW horns [4], or other methods to form compact,high gain antennas. The fence guide [10] has been used to formhorn antennas as well.

Other recent research in the area of horn antennas aims toreduce the sidelobe level through the use of periodic structures,or metamaterials along the metallic walls of the horn [2]. Thesestructures reduce the magnitude of the electric fields close to theedges, thus reducing fringing effects at the mouth of the hornand consequently reducing the sidelobe levels.

This paper begins with a description of a thin microstrip-fedH-guide design, where the metallic plate separation is close to

. Theoretical, simulated, and measured results for an 8 to18 GHz back-to-back Bzier shaped microstrip to H-guide tran-sition using Rogers 5880 substrate are then presented. We havepreviously discussed a low frequency (3 to 7 GHz) transitionusing FR4 in [11]. Finally, theoretical, simulated, and measuredresults for the H-guide horn antenna and its aperture are thenpresented.

II. H-GUIDEThe goal of this section is to describe the parameters for

the H-guide to be used in the proposed microstrip line-fed di-electrically guided horn antenna. In this case, the dimensionsare chosen to force a single mode, where all other propagatingmodes are lossy, or evanescent. The design, fabrication, andtesting of the thin H-guide are made possible through the useof a low profile microstrip to H-guide transition, which is dis-cussed in the next section.

Consider the H-guide [6] shown in Fig. 1. The electric fieldlines of the dominant mode are shown, where the elec-tric field is directed in the y-direction for z-propagation. Cut-off

0018-926X/$26.00 2011 IEEE


Fig. 1. Side view of the H-guide structure operating in the fundamental mode.The size of the arrows represents the magnitude of the electric field. Propagationis in the z direction.

Fig. 2. Two dimensional view of the fields for the a) TE00 vs the b) TE10modes.

frequencies can be found for various modes using equations inthis section. A comparison with the next higher mode isshown in Fig. 2.

The distance between the plates is made small, so that theelectric field parallel to the plates is forced to zero as shown inFig. 1. The magnetic field for this mode will then exist in the xzplane for propagation in the z direction.

The guide wavelength for even TE modes in the H-guide canbe found by solving the following equations for , thesolution to the [6] and [12]

(1)

where variable is the width of the dielectric as shown in Fig. 1,is the transverse wavenumber inside the dielectric, and

is the first transverse wavenumber outside the dielectric. is therelative dielectric constant, and is the propagation constant infree space.

is then substituted into the characteristic equation insidethe dielectric to find , the guide wavelength for the solu-tion as follows:

(2)

Fig. 3. Insertion loss over a 10.0 cm long, 62 mil (1.575 mm) thick substratefor an H-guide that is 10 mm wide for the TE00 and TE10 modes. The cutofffrequency is marked with a dotted line at 13.7 GHz.

For this design, an H-guide made of a 10 mm wide sectionof dielectric is formed out of 62 mil (1.575 mm thick) Rogers5880 . The guide wavelength, , for even TE modeswithin a single H-guide can be found by solving the equationsfor , the allowed even mode following procedures in [6],[12], using (1), where at 13.7 GHz, found to be equal to 16.4mm. Since the distance between the top and bottom metallicplates of the substrate is only 1.575 mm , higher ordermodes in the vertical direction ( , , etc.) are highlyevanescent. Consequently, TM modes, where the electric fieldsare parallel to the metallic plates, are also highly evanescent.The waveguide is therefore predominantly single-modeup to the cutoff frequency of the second mode ( , ,odd) at 13.7 GHz as estimated by the expression [12]

(3)

where is the speed of light in free space.Using Ansoft HFSS numerical simulation software [13], it is

possible to plot the attenuation curves for the first two modes,and over a length of 10 cm, as shown in Fig. 3.

In the figure, above a frequency of approximately 15 GHz, theis attenuated only slightly, which confirms that the cal-

culation of the cutoff frequency of 13.7 GHz is reasonably ac-curate. Note, however, that the cutoff frequency is not as welldefined as for a rectangular metallic waveguide, where bound-aries conditions end abruptly at metallic sidewalls.

III. BZIER-CURVE SHAPED MICROSTRIP TO H-GUIDETRANSITION

An obvious choice of common transmission lines to excitethe single mode in the thin H-guide is the microstrip line,because of the small vertical dimension of 1.575 mmproposed in the previous section. Transitions such as those pro-posed in [14] and similar transitions are intended for Non-Ra-diative Dielectric (NRD) structures, cannot excite the intendedH-guide mode, require two separate low-loss substrates,

WONG et al.: WIDEBAND DIELECTRICALLY GUIDED HORN ANTENNA WITH MICROSTRIP LINE TO H-GUIDE FEED 727

Fig. 4. (a) Top view of the Wedge Radial Waveguide with metallic side walls.Cylindrical waves are the preferred modes. (b) Top view of the Wedge RadialWaveguide with air walls. Cylindrical waves are the preferred modes.

and are inherently narrowband because of their resonant be-havior. The solution proposed here is a non-resonant, wideband,low-loss Bzier-curve shaped microstrip to H-guide transitionusing a low-loss Rogers 5880 substrate, previously discussed in[11] using a low-cost FR4 substrate.

Consider the air-filled metallic wedge radial waveguideshown in Fig. 4(a). On the left and right walls, metallicboundaries, or perfect electric conductors (PEC), enforce thetangential electric fields to be zero. The dominant mode insuch a structure is a radial mode [12]. A line of equal phaseis shown as a dotted line in the figure for propagation in the

direction. For the design of a transition to the fundamentalH-guide mode as shown in Fig. 1, this is not an efficientchoice, since both the microstrip line and the H-guide do nothave side metallic walls. The transition from the mouth of thistransition to and H-guide would therefore be very abrupt.

Now, consider the same waveguide-based transition, with aheight of 1.575 mm and filled with air, except with virtual per-fect magnetic conductors (PMC) instead of PEC for walls, asshown in Fig. 4(b). The virtual PMC boundaries enforce themagnetic fields to be zero, and hence again, radial waves arethe preferred modes of operation. The lines of equal phase willbe identical for propagation in the direction, however, no

Fig. 5. Profiled horn antenna shape.

metallic side edges are needed, thus improving the transitionfrom the microstrip to the H-guide. Only a fraction of the wavesthat are leaving the mouth of the transition, however, are prop-agating in the desired direction towards the H-guide. Mostwaves propagate at an angle to that axis, which is energy thatwill be lost at the transition.

The profiled transition shape shown in Fig. 5 that is discussedin [11] is used as a solution to encourage the propagation ofplanar, equal phase waves at the mouth of the transition, as op-posed to radial waves. Planar waves, as opposed to radial waves,are less lossy at the mouth of the transition because the rectan-gular waveguide shape in this region prefers planar waves. Thesmooth transition from radial waveguide to rectangular wave-guide requires a profiled shape, as shown in Fig. 5.

To form the required profiled shape, several different for-mulas have been presented [15], such as sine squared, exponen-tial, hyperbolic, or polynomial curves. In this design, however,a class of cubic spline curves, called the Bzier curve, is used.The Bzier curve is flexible enough to approximate various dif-ferent curves while maintaining a smooth shape and allows thespecification of the slope, or direction, at both sides of the tran-sition.

The parametric form of the cubic Bzier curve [16] using a3rd degree Bernstein polynomial over points P0 through P3 isgiven by

(4)where the parameter, , varies between 0 and 1

To examine the direction of the curve at its endpoints, con-sider the derivative of B with respect to

(5)


Fig. 6. Control points for Bezier curve.

Fig. 7. A microstrip to H-guide transition with tapered dielectric.

We see that at the endpoints, where the parameter t is 0 or1, the direction of the Bzier curve can be found by taking thedifference between the two points, P0 and P1, or P2 and P3, re-spectively. Consequently, as shown in Fig. 6, by putting P0 andP1 in line with the axis, the beginning of the curve becomesparallel to the axis. The same property is applied to pointsP2 and P3 as shown in the same figure.

To form a smooth transition from a microstrip line to anH-guide, it is proposed that the transition be considered inseveral stages as shown in Fig. 7. The microstrip line tapersslowly into a wide microstrip line in Section I. To maintain asmooth transition, the slope in this section is enforced to beparallel to the direction of the microstrip line. In Section II,the microstrip line crosses over the dielectric and the profiledradial waveguide as described in Fig. 5. Section III is theH-guide. At the beginning of this section, the slope of the tran-sition is enforced to be parallel to the direction of the H-guideto form the rectangular waveguide section of the transition.In addition, Sections I to II are smoothed by slowly taperingthe air gap in the dielectric as compared to the abrupt air gap

Fig. 8. (a) Dimension of the Bezier transition top sheet. Bzier curve controlpoints P0 (2.425, 90.0), P1 (2.425, 62.0), P2 (25.0, 62.0), and P3 (25.0, 40.0)are shown, where units are in mm. b) Dimension of the Bezier transition boardcutouts. Bzier curve control points P0 (6.6, 90.0), P1 (6.6, 65.25), P2 (30.0,55.3125), and P3 (35.0, 47.0) are shown, where units are in mm.

originally proposed in [11]. Dimensions for the inner and outercurves are shown in the caption in Fig. 8. A simulated plot ofthe magnitude of the electric fields are shown in Fig. 9.

For this prototype, the actual top sheet is cut from a thin brasssheet and has finite thickness. The brass layer touches the mi-crostrip line. For the middle layer, the board is made out ofRogers 5880 with a microstrip line for testing the connectorsbefore the final assembly. The bottom layer is formed out of acopper-plated FR4 sheet, where the metallic copper layer musttouch the middle layer. A photograph of the assembled pro-totype and measurements of the insertion and return loss areshown in Fig. 10.


Fig. 9. Magnitude of the electric fields within the back-to-back microstrip toH-guide transition.

Fig. 10. Photograph of the assembled prototype.

Fig. 11. Comparison of simulated vs. measured insertion and return loss.

The prototype shown in Fig. 10 was tested and measured. Theinsertion and return loss are plotted in Fig. 11.The match is fairly good up to 15 GHz, however, the measuredinsertion loss tends to fall away as 18 GHz approaches. The dis-crepancies between measured and simulated results are due tosome additional loss due to the coaxial to microstrip connectorsand their soldering on the board that was not taken into accountin the simulation. The same connector also causes degradationin the return loss from roughly 25 dB to 15 dB.

IV. ANTENNA DESIGNUsing the Bzier shaped microstrip to thin H-guide transi-

tion discussed in the previous section, the antenna design be-comes relatively straightforward. One possible design has beendiscussed in [5], where the usage was proposed over a smallbandwidth only. In this design, we show through simulation thatthis antenna exceeds a return loss of 10 dB from 8 to 18 GHz

Fig. 12. Gapped H-Guide. The magnitude and direction of the electric field isrepresented by arrows. The effective dielectric constant in region I is higher thanthat of region II.

while achieving a gain that varies between 12 and 18 dBi. How-ever, due to manufacturing imperfections the fabricated proto-type achieves a maximum of about 16 dBi at only 16 GHz.Some advantages of this design are that the wave exiting theaperture is nearly planar, and is mostly concentrated in the di-electric, away from metallic edges in the vertical direction, andwith no metallic side walls required. The concentration awayfrom metallic edges in the vertical direction reduces sidelobes,as does the absence of metallic side walls.

Let us first consider the groove guide as proposed in [17].If an air gap is added between the dielectric and top metallicplate, the effective dielectric constant of the dielectric is simplyreduced, so that guided propagation is still possible. We havebriefly discussed this property in [5]. Now, consider the casewhere an air gap is added above and below the dielectric so thatthe dielectric floats between the two metallic plates and propa-gation is still possible. The proposed dual air gap configurationis shown in Fig. 12. The width of the dielectric slowly increasesto compensate for the loss in effective dielectric constant as theair gap widens.

A rough approximation for the reflection coefficient seen atthe input can be formulated as shown in Fig. 13. If Section Iapproaches a propagation constant that is equal to an H-guide

, then . Each subsequent section then has an ef-fective propagation constant, , and length . The reflectioncoefficient, with reference to the start of the taper, is then givenas [18]

(6)

The contribution of each discrete reflection in (6) is not asimple task and planned for discussion in future studies. For thispaper, a numerical simulation is, therefore, shown in Fig. 14,where the angle is varied, while the taper in the dielectricremains constant.

In this parametric study, the radiation boundary is placed atthe aperture of the horn, so that the effects of the aperture on


Fig. 13. Segmentation of an H-guide taper. Each section has an effective prop-agation constant, , and length .

Fig. 14. Variation in angle of metallic plates. Inner dielectric tapers from 10mm to 30 mm for each run. The horn angle is measured from the horizontalplane to the plane of one of the metallic plates.

the reflection coefficient are removed. The variations in Fig. 14are therefore due only to the change in metallic plate angles. Wecan conclude that for an optimal angle for operation from 6 to 18GHz, where a return loss of 10 dB or better is required, is equalto 8 degrees in the vertical plane, close to the H-guide to gappedH-guide transition. This corresponds to a dielectric taper from10 mm to 30 mm, given that Rogers 5880 is used as a substrate,and the H-guide plate separation distance is equal to 1.575 mm.

A two-stage design, with a smaller angle to improve thereturn loss, and a large angle, is used to widen the apertureand thus improve the gain. Finally, to speed up the wave at theaperture and thus adjust the phase, a curved shape with radiusHW/2 is cut into the dielectric. The top and side views of theantenna are shown in Fig. 15.

Fig. 15. Improved dual air gap H-guide aperture horn antenna design using1.575 mm thick Rogers 5880. , , , , , , ,

, .

V. ANTENNA PROTOTYPE SIMULATION AND MEASUREMENT

In this section, the antenna prototype fabrication, return loss,and radiation pattern measurements are discussed. The antennawas first simulated using a waveport that directly feeds theH-guide in HFSS as shown in Fig. 16. This type of feedingforces the fundamental mode and therefore allows the fullpotential of the antenna design to be seen. The simulation wasthen performed using the microstrip to H-guide transition asdiscussed in previous sections as shown in Fig. 17. This is amore realistic case for comparison with measured data, sincethe thin H-guide must be fed in some way. At lower frequen-cies, all modes higher than the fundamental are evanescent,and therefore, any higher order modes that are introduced bythe transition are naturally attenuated through the structure.However, at higher frequencies, other modes may propagatefreely, so the that modes that may be introduced by the tran-sition or fabrication imperfections may propagate and affectthe radiation pattern. These effects are most notable in thesidelobes. Finally, the assembled prototype is shown in Fig. 18.

The return loss plots for the three cases are shown in Fig. 19.When comparing the return loss of two simulated results, it canbe seen that the transition is so effective that is does not intro-duce any significant penalty in the return loss. When comparingthe measured results with the simulated ones, it can be con-cluded that the transition has been reasonably well fabricatedsince the variations and trends in the return losses are all sim-ilar.

Radiation patterns in azimuth and elevation for simulated andmeasured data are shown in Figs. 20through 25 for 9, 12, and 15GHz. The sidelobes have been degraded once the microstrip to


Fig. 16. a) Antenna Simulation using waveport. b) Electric fields within an-tenna at 15 GHz in the horizontal plane within antenna simulation using a wave-port. c) Electric fields at 15 GHz in the vertical plane within antenna simulationusing a waveport.

Fig. 17. (a) Antenna simulation using microstrip to H-guide transition. (b)Electric field at 15 GHz within antenna simulation.

Fig. 18. Photograph of antenna prototype including microstrip to H-guide tran-sition.

H-guide transition has been introduced, however, the measureddata matches the simulated data reasonably well if the transi-tion is included. Some additional degradation in the sidelobesis visible at 15 GHz in the elevation pattern, however, this ismostly due to unexpected additional leakage from the microstripto H-guide transition.

The transition from coaxial connector to microstrip line,which is not included in the simulation, caused many smallsidelobes to appear at 15 GHz. We have shown this to be the


Fig. 19. Return loss for the measured prototype compared to the sim-ulations using a waveport, and using the microstrip to H-guide transition.

Fig. 20. Azimuth pattern at 9 GHz.

Fig. 21. Elevation pattern at 9 GHz.

cause of these ripples by covering this transition with foamand aluminum foil, which significantly reduced these sidelobelevels in subsequent measurements. Because of the crudenessof this approach, these results were not included. In futuredesigns, a shielded transition may be used.

Simulated data for 18 GHz is shown in Figs. 26 and 27 using awaveport only. A simulation that includes the transition was notpossible at 18 GHz due to memory limitations in the computerbeing used. The measured data at 18 GHz showed non-ideal




sidelobe levels as compared to simulated results and thereforehave not been shown here. The sidelobe levels at 18 GHz usingthe waveport, however, show promising results.

The peak measured gain is approximated by first subtractingthe power received from a gain stardard horn antenna fromthe power received with the H-guide horn antenna, where thetransmit antenna is a dual-ridged horn antenna. This delta,as shown in Fig. 28, is then compared to the known gain



Fig. 26. Simulated azimuth pattern with waveport at 18 GHz.

Fig. 27. Simulated elevation pattern with waveport at 18 GHz.

standards gain. The peak measured gain as shown in Table Iexceeds the simulated results partly because the higher ordermodes created by the transition change the fields at the aperture,causing a narrower main beam. Errors can also partly be due tomeasurement and calibration errors.

The simulated gain with the transition is slightly higher thanthe gain with the waveport due to the distortion of the beamas seen in Figs. 20to 25. A gain table in Table I outlines theseresults.

Fig. 28. The actual measured power received from a gain standard metallicair-filled horn antenna is compared to the power received from the H-guide hornantenna in 0.1 GHz steps as shown above. The power received is at the boresight (center) without any re-pointing where the transmit antenna is a widebanddual-ridged horn antenna.

TABLE IGAINS

Measured gain is approximated by comparing received power with a gainstandard horn antenna and applying the difference to the known gain. Gainis approximated 12 GHz because the received power of the gain standardwas not available.

VI. CONCLUSIONIn this paper, a new wideband high gain H-guide horn

antenna, and a microstrip to H-guide transition have beenpresented and discussed. The proposed new antenna has someunique advantages compared to traditional horn antennas, suchas lower dependence upon precise angles, similar gains, andsimpler designs. This antenna also successfully demonstratesthat the thin single H-guide is a useful transmission line. Mea-sured data including return loss and radiation patterns haveconfirmed that simulated results are reasonably accurate.

VII. ACKNOWLEDGEMENT

The authors would like to thank all the technicians who as-sisted in the fabrication and measurement of prototypes dis-cussed in this paper. Namely, J. Landry from Concordia Uni-versity, and T. Antonescu and M. Thibault from Ecole Poly-technique in Montreal, QC, Canada. A. R. Sebak would liketo thank the King Saud University and the National Plan forSciences and Technology (NPST) for funds through ResearchGrant 09ELE858-02.


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Michael Wong (S05M10) received the B.Sc.degree in electrical engineering from QueensUniversity, Kingston, ON, Canada, in 1997 andthe M.Sc. and Ph.D. degrees, both in electricalengineering, from Concordia University, Montreal,QC, Canada, in 2006 and 2010, respectively.

From 1998 to 2004, he was a Systems EngineeringAssociate with Mobile Satellite Ventures (now Light-Squared), Ottawa, ON, Canada, working on trafficengineering for mobile satellite telephony with theMSAT satellites. During this time, he also performed

consulting work with Telesat, a Canadian satellite operator, to study the ef-fects of interference from solar effects and terrestrial services on fixed satelliteservices. He is currently with Research in Motion (RIM) working on calibra-tion software for board level testing with the next generation Blackberry smart-phones.

Abdel Razik Sebak (F10) received the B.Sc.degree (with honors) in electrical engineering fromCairo University, Cairo, Egypt, , in 1976 and theB.Sc. degree in applied mathematics from EinShams University, Cairo, in 1978. He received theM.Eng. and Ph.D. degrees from the University ofManitoba, Winnipeg, MB, Canada, in 1982 and1984, respectively, both in electrical engineering.

From 1984 to 1986, he was with the Canadian Mar-coni Company, working on the design of microstripphased array antennas. From 1987 to 2002, he was a

Professor in the Electrical and Computer Engineering Department, University ofManitoba. He is a Professor of Electrical and Computer Engineering, ConcordiaUniversity, Montreal, QC, Canada. His current research interests include phasedarray antennas, computational electromagnetics, integrated antennas, electro-magnetic theory, interaction of EM waves with new materials and bio-electro-magnetics.

Dr. Sebak received the 2000 and 1992 University of Manitoba Merit Awardfor outstanding Teaching and Research, the 1994 Rh Award for OutstandingContributions to Scholarship and Research, and the 1996 Faculty of EngineeringSuperior Academic Performance. He is a Fellow of IEEE. Dr. Sebak has servedas Chair for the IEEE Canada Awards and Recognition Committee (20022004)and the Technical Program Chair of the 2002 IEEE-CCECE and 2006 ANTEMconferences.

Tayeb A. Denidni (M98SM04) received the B.Sc.degree in electronic engineering from the Universityof Setif, Setif, Algeria, in 1986, and the M. Sc. andPh.D. degrees in electrical engineering from LavalUniversity, Quebec City, QC, Canada, in 1990 and1994, respectively.

From 1994 to 1996, he was an Assistant Professorwith the engineering department, Universit duQuebec in Rimouski (UQAR), Quebec, Canada.From 1996 to 2000, he was also an AssociateProfessor at UQAR, where he founded the Telecom-

munications laboratory. Since August 2000, he has been with the PersonalCommunications Staff, Institut National de la Recherche Scientifique (INRS),Universit du Quebec, Montreal, QC, Canada. He founded the RF laboratory,INRS-EMT, Montreal, for graduate student research in the design, fabrication,and measurement of antennas. He possesses ten years of experience with an-tennas and microwave systems and is leading a large research group consistingof two research scientists, five Ph.D. students, and three M.S. students. Overthe past ten years, he has graduated numerous graduate students. He has servedas the Principal Investigator on numerous research projects on antennas forwireless communications. Currently he is actively involved in a major projectin wireless of PROMPT-Quebec (Partnerships for Research on Microelec-tronics, Photonics and Telecommunications). His current research interestsinclude planar microstrip filters, dielectric resonator antennas, electromag-netic-bandgap (EBG) antennas, antenna arrays, and microwave and RF designfor wireless applications. He has authored over 100 papers in refereed journals.He has also authored or coauthored over 150 papers and invited presentationsin numerous national and international conferences and symposia.

From 2006 to 2007, Dr. Denidni was an associate editor for IEEE ANTENNASAND WIRELESS PROPAGATION LETTERS. From 2008 to 2010, he served as anassociate editor for IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION.He is a member of the Order of Engineers of the Province of Quebec, Canada.He is also a member of URSI (Commission C).


CPW-Fed Cavity-Backed Slot RadiatorLoaded With an AMC Reflector

Johan Joubert, Senior Member, IEEE, J. (Yiannis) C. Vardaxoglou, Senior Member, IEEE, William G. Whittow, andJohann W. Odendaal, Senior Member, IEEE

AbstractA low profile coplanar waveguide (CPW) fed printedslot antenna is presented with uni-directional radiation properties.The slot antenna radiates above a closely spaced artificial magneticconducting (AMC) reflector consisting of an array of rectangularpatches, a substrate and an electric ground plane. The electromag-netic bandgap (EBG) performance of the cavity structure betweenthe upper conducting surface in which the slot is etched, and theground plane at the bottom of the reflector, is investigated using anequivalent waveguide feed in the place of a half-wavelength sectionof the slot antenna. From the reflection coefficient of the equivalentwaveguide feed one can determine the frequency band where min-imum energy will be lost due to unwanted radiation from the cavitysides. The dimensions of the cavity were found to be very importantfor minimum energy loss. Experimental results for the final an-tenna design (with a size of ), mountedon a back plate, exhibit a 5% impedance band-width, maximum gain in excess of 10 dBi, low cross-polarization,and a front-to-back ratio of approximately 25 dB. This low-profileantenna with relatively high gain could be a good candidate for a2.4 GHz WLAN application.

Index TermsElectromagnetic bandgap materials, periodicstructures, slot antennas.

I. INTRODUCTION

S LOT radiators are suitable candidates for portable units andunobtrusive base stations of mobile communications sys-tems, because of their compactness, flush-mounting and simplestructure [1]. Slot radiators are also attractive when an antennahas to be integrated into a metallic surface eg. as single ele-ments or arrays conformal to an airborne structure for electronicwarfare applications [2]. When a slot is printed on one sideof a substrate, the element will radiate bidirectionally, and anelectric conducting surface has to be added at a distance of aquarter-wavelength below the slot as a reflector to achieve op-timum uni-directional radiation. If the reflector distance is re-duced to achieve a lower profile, which is highly desirable forconformal antennas in most cases, parallel plate modes will beexcited that can cause significant energy leakage. A fair amountof work was published on so-called conductor-backed CPW-fed

Manuscript received May 05, 2011; revised July 15, 2011; accepted August16, 2011. Date of current version February 03, 2012.J. Joubert and J. W. Odendaal are with the Centre for Electromagnetism, Uni-

versity of Pretoria, Department of Electrical, Electronic and Computer Engi-neering, Pretoria 0002, South Africa (e-mail: [email protected]; [email protected]).J. C. Vardaxoglou and W. G. Whittow are with the Centre for Mobile Com-

munications Research, Department of Electronic and Electrical Engineering,Loughborough University, Loughborough LE11 3TU, U.K. (e-mail: [email protected], [email protected]).Digital Object Identifier 10.1109/TAP.2011.2173152

slot antennas, where single or multiple dielectric layers were in-serted between the upper conducting surface containing the slotand the reflector [3][6]. Energy leakage between the parallelplates because of unwanted modes remains a problem for con-ductor-backed slot radiators and special techniques have to beused to overcome this, eg. the use of closed cavities behind theslot [1], the placement of shorting pins around the slot [2], or theuse of twin slot configurations for phase cancellation [5], [6].This paper presents results of an investigation of a CPW-fed

slot radiator above a metamaterial-based AMC surface as a re-flector. The intention was to investigate a single radiating el-ement with a low profile and compact lateral dimensions, inorder to improve on results for a similar structure investigated in[7]. Fig. 1 shows the geometry of the proposed structure. It cana

2012_60_02_P2

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