2010_58_04.pdf

APRIL 2010 VOLUME 58 NUMBER 4 IETPAK (ISSN 0018-926X)

PAPERS

AntennasVertically Multilayer-Stacked Yagi Antenna With Single and Dual Polarizations . . . . . . . . . . . . . . . . . . . . O. Kramer, T. Djerafi, and K. Wu 1022A Compact Tri-Band Monopole Antenna With Single-Cell Metamaterial Loading . . . . J. Zhu, M. A. Antoniades, and G. V. Eleftheriades 1031Investigation Into the Polarization of Asymmetrical- Feed Triangular Microstrip Antennas and its Application to Reconfigurable

Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. Sung 1039Ultrawideband Dielectric Resonator Antenna With Broadside Patterns Mounted on a Vertical Ground Plane Edge . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . K. S. Ryu and A. A. Kishk 1047Transparent Dielectric Resonator Antennas for Optical Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . E. H. Lim and K. W. Leung 1054A Low-Profile Linearly Polarized 3D PIFA for Handheld GPS Terminals . . . . . .. . . . . . A. A. Serra, P. Nepa, G. Manara, and R. Massini 1060Analysis of Strong Coupling in Coupled Oscillator Arrays . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . V. Seetharam and L. W. Pearson 1067ArraysCompact Elongated Mushroom (EM)-EBG Structure for Enhancement of Patch Antenna Array Performances . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Coulombe, S. Farzaneh Koodiani, and C. Caloz 1076Low-Profile PIFA Array Antennas for UHF Band RFID Tags Mountable on Metallic Objects . . . . . . .. . . . . . . H.-D. Chen and Y.-H. Tsao 1087CMOS Phased Array Transceiver Technology for 60 GHz Wireless Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Fakharzadeh, M.-R. Nezhad-Ahmadi, B. Biglarbegian, J. Ahmadi-Shokouh, and S. Safavi-Naeini 1093Direction of Arrival Estimation in Time Modulated Linear Arrays With Unidirectional Phase Center Motion . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. Li, S. Yang, and Z. Nie 1105Reactive Energies, Impedance, and Factor of Radiating Structures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. A. E. Vandenbosch 1112ElectromagneticsElectromagnetic Boundary Conditions Defined in Terms of Normal Field Components . . . . . . . . .. . . . . . . . . I. V. Lindell and A. H. Sihvola 1128A Cloaking Metamaterial Based on an Inhomogeneous Linear Field Transformation . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . S. Maci 1136The Design of Broadband, Volumetric NRI Media Using Multiconductor Transmission-Line Analysis . .. . S. M. Rudolph and A. Grbic 1144Uniform Asymptotic Evaluation of Surface Integrals With Polygonal Integration Domains in Terms of UTD Transition Functions . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. Carluccio, M. Albani, and P. H. Pathak 1155Non-Orthogonal Domain Parabolic Equation and Its Tilted Gaussian Beam Solutions . . . . . . . . . . . . . . . . . . . . . . . Y. Hadad and T. Melamed 1164Emissivity Calculation for a Finite Circular Array of Pyramidal Absorbers Based on Kirchhoffs Law of Thermal Radiation . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Wang, Y. Yang, J. Miao, and Y. Chen 1173Near-Field Electromagnetic Holography in Conductive Media . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . E. G. Williams and N. P. Valdivia 1181Thin Microwave Quasi-Transparent Phase-Shifting Surface (PSS) . . . . . . . . . . . . . . . . . . . . . . . . N. Gagnon, A. Petosa, and D. A. McNamara 1193Polarization Rotating Frequency Selective Surface Based on Substrate Integrated Waveguide Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. A. Winkler, W. Hong, M. Bozzi, and K. Wu 1202Design and Analysis of a Tunable Miniaturized-Element Frequency-Selective Surface Without Bias Network . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . F. Bayatpur and K. Sarabandi 1214A Novel Band-Reject Frequency Selective Surface With Pseudo-Elliptic Response . . . . . . . . . . . . . .. . . . . . . . . . . . . . A. K. Rashid and Z. Shen 1220Broadening of Operating Frequency Band of Magnetic-Type Radio Absorbers by FSS Incorporation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. N. Kazantsev, A. V. Lopatin, N. E. Kazantseva, A. D. Shatrov, V. P. Maltsev, J. Vilckov, and P. Sha 1227

(Contents Continued on p. 1021)

(Contents Continued from Front Cover)

Numerical MethodsEmbedding Caldern Multiplicative Preconditioners in Multilevel Fast Multipole Algorithms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Peeters, K. Cools, I. Bogaert, F. Olyslager, and D. De Zutter 12363D Isotropic Dispersion (ID)-FDTD Algorithm: Update Equation and Characteristics Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . W.-T. Kim, I.-S. Koh, and J.-G. Yook 1251Aperture Antenna Modeling by a Finite Number of Elemental Dipoles From Spherical Field Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Serhir, J. M. Geffrin, A. Litman, and P. Besnier 1260Invasive Weed Optimization and its Features in Electromagnetics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. Karimkashi and A. A. Kishk 1269PropagationRadio-Wave Propagation Into Large Building StructuresPart 1: CW Signal Attenuation and Variability . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . W. F. Young, C. L. Holloway, G. Koepke, D. Camell, Y. Becquet, and K. A. Remley 1279Radio-Wave Propagation Into Large Building StructuresPart 2: Characterization of Multipath . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . K. A. Remley, G. Koepke, C. L. Holloway, C. A. Grosvenor, D. Camell, J. Ladbury, R. T. Johnk, and W. F. Young 1290Numerical Investigations of and Path Loss Predictions for Surface Wave Propagation Over Sea Paths Including Hilly Island

Transitions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . G. Apaydin and L. Sevgi 1302On the Effective Low-Grazing Reflection Coefficient of Random Terrain Roughness for Modeling Near-Earth Radiowave

Propagation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. Liao and K. Sarabandi 1315Truncated Gamma Drop Size Distribution Models for Rain Attenuation in Singapore . . . . . .. . . . . . L. S. Kumar, Y. H. Lee, and J. T. Ong 1325ScatteringComparison of TE and TM Inversions in the Framework of the Gauss-Newton Method . . . . . . . . . . . .. . . . . . . . . . . . P. Mojabi and J. LoVetri 1336Experimental Study of the Invariants of the Time-Reversal Operator for a Dielectric Cylinder Using Separate Transmit and Receive

Arrays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Davy, J.-G. Minonzio, J. de Rosny, C. Prada, and M. Fink 1349WirelessWireless Communication for Firefighters Using Dual-Polarized Textile Antennas Integrated in Their Garment . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . L. Vallozzi, P. Van Torre, C. Hertleer, H. Rogier, M. Moeneclaey, and J. Verhaevert 1357A Compact Six-Port Dielectric Resonator Antenna Array: MIMO Channel Measurements and Performance Analysis . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . R. Tian, V. Plicanic, B. K. Lau, and Z. Ying 1369

COMMUNICATIONS

Dual-Band Multiple Beam Antenna System Using Hybrid-Cell Reuse Scheme for Non-Uniform Satellite Communications Traffic . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Wang, S. K. Rao, M. Tang, and C.-C. Hsu 1380

A Low Cross-Polarization Smooth-Walled Horn With Improved Bandwidth . .. . L. Zeng, C. L. Bennett, D. T. Chuss, and E. J. Wollack 1383Design of Compact Differential Dual-Frequency Antenna With Stacked Patches . . . . L. Han, W. Zhang, X. Chen, G. Han, and R. Ma 1387A Broadband Impedance Matching Method for Proximity-Coupled Microstrip Antenna . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . D. Sun and L. You 1392Electromagnetically Coupled Band-Notched Elliptical Monopole Antenna for UWB Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . R. Eshtiaghi, J. Nourinia, and C. Ghobadi 1397Bandwidth Enhancement of Printed E-Shaped Slot Antennas Fed by CPW and Microstrip Line . . . . . . . . . . . . . A. Dastranj and H. Abiri 1402Handling Sideband Radiations in Time-Modulated Arrays Through Particle Swarm Optimization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . L. Poli, P. Rocca, L. Manica, and A. Massa 1408Experimental Validation of a Linear Array Consisting of CPW Fed, UWB, Printed, Loop Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . F. M. Tanyer-Tigrek, I. E. Lager, and L. P. Ligthart 1411Limits on the Amplitude of the Antenna Impulse Response . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . H. Foltz and J. McLean 1414AIM Analysis of 3D PEC Problems Using Higher Order Hierarchical Basis Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . B. Lai, X. An, H.-B. Yuan, N. Wang, and C.-H. Liang 1417Generalized Stability Criterion of 3-D FDTD Schemes for Doubly Lossy Media . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. Y. Heh and E. L. Tan 1421Low Observable Targets Detection by Joint Fractal Properties of Sea Clutter: An Experimental Study of IPIX OHGR Datasets . . . . . . .

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . X.-K. Xu 1425

CALL FOR PAPERS

Joint Special Issue on Multiple-Input Multiple-Output (MIMO) Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1430Joint Special Issue on Ultrawideband (UWB) Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1431

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Digital Object Identifier 10.1109/TAP.2010.2047538

1022 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 4, APRIL 2010

Vertically Multilayer-Stacked Yagi Antenna WithSingle and Dual Polarizations

Olivier Kramer, Tarek Djerafi, and Ke Wu, Fellow, IEEE

AbstractThere are many applications such as local posi-tioning systems (LPS) and wireless sensor networks that requirehigh-directivity and compact-size or small footprint antennas.The classical Yagi-Uda antenna may be useful in meeting suchdemands, which however, becomes very large in size to achievea high-gain performance due to a large number of directors aswell as space required between those elements. In this paper,high-gain yet compact stacked multilayered Yagi antennas areproposed and demonstrated at 5.8 GHz for LPS applications.This structure makes use of vertically stacked Yagi-like parasiticdirector elements that allow easily obtaining a simulated gain of12 dB. Two different antenna configurations are presented, onebased on dipole geometry for single polarization, and the other ona circular patch to achieve dual polarization. The characteristicsof these antennas with respect to various geometrical parametersare studied in order to obtain the desired performance. Measuredresults of the fabricated antenna prototypes are in good agreementwith simulated results. The measured dipole Yagi antenna yields11 dB gain over 14% bandwidth with a size of .Radiation patterns of the dual-polarized Yagi antenna are nearlyidentical to those of the single-polarized antenna, which has asize of , and also its two-port isolation isfound to be as low as 25 dB over 4% bandwidth. The proposedantennas present an excellent candidate for compact and low-costmicrowave and millimeter-wave integrated systems that requirefixed or variable polarization capabilities and small surface foot-print.

Index TermsBalun, circular patch, dipole, dual polarization,microstrip antenna, stacked antenna, Yagi-Uda.

I. INTRODUCTION

T HE Local positioning system (LPS) is a radio system usedto search for and track down in real time objects of in-terest within a limited space range. This is a typical applicationof wireless sensor networks. With the LPS, a mobile object canbe localized and can also collect information about its position ata precise time instant [1]. Applications of such a LPS system aremultiple and diverse. Those systems are used for zone securitywith great efficiency, particularly indoor applications such as atairports where security is mandatory. Also location tracking can

Manuscript received March 12, 2009; October 19, 2009; accepted October24, 2009. Date of publication January 26, 2010; date of current version April 07,2010. This work was supported in part by the Natural Sciences and EngineeringResearch Council of Canada (NSERC) and in part by Regroupement strategiqueof FQRNT.

The authors are with the Dpartement de Gnie lectrique, Poly-GramesResearch Center, cole Polytechnique de Montral, Montral, QC H3T1J4, Canada (e-mail: [email protected]; [email protected];[email protected]; [email protected]).


clearly improve maneuver of the business and logistics man-agement, for example, hospitals can improve their healthcareservices by keeping a constant track of the location of doctors,nurses, equipments, etc. LPS can also be used in many otherapplications such as construction and agriculture, leisure andsports, etc. For a full integration with GPS systems that are usu-ally used for outdoor scenarios, LPS is expected to achieve thechallenging task of being accurate, low-cost and autonomousat ISM frequencies such as 5.8 GHz. Therefore, compact-size,light-weight and low-cost antenna is required for such LPS de-sign and implementation.

The Yagi antenna, which has been very popular because of itssimplicity as well as its customizable high gain (three-elementYagi antenna can reach 9 dB when optimized) [2], can be usedfor this type of applications. The basic unit of a three-elementYagi antenna consists of a half-wavelength driver dipole backedby a longer reflector and a director on the other side.

Several microstrip-based Yagi or quasi-Yagi antenna struc-tures have been reported in the literature [2][8]. Interesting ap-proaches are related to the design of a microstrip Yagi arraybased on the microstrip patch antenna such as the array de-veloped in [3] for mobile satellite system in the L-band. Theantenna developed was based on patches, consisting of one re-flector, one driving element and two directors. An array of fourantenna elements was successfully used to achieve the requiredperformances. The design reported in [4] was made on the basisof patches instead of dipoles as the driver element. An inter-esting printed Yagi antenna configuration was presented in [5],[6] where the Yagi-like printed dipole array antenna was fed bya microstrip-to-coplanar strip transition. In this case, a truncatedmicrostrip ground plane was utilized as a reflecting element. Onthe other hand, an active quasi-Yagi version was proposed [7]for 5.8 and 60 GHz applications. In [8], the proposed antennaconsisted of a dipole as a driving element, a parasitically cou-pled reflector and six directors. The antenna was designed byutilizing the same design rules as used in the conventional Yagidipole antenna while taking into account the fact that the an-tenna was made on a planar substrate. The antenna was designedfor 5 GHz band and has achieved a gain of 10 dB.

To overcome the problem of size and footprint within theplanar structure, two novel high-gain compact structures basedon the Yagi-Uda antenna concept are presented for the first timein this work. These structures are constructed in a multilayertopology by stacking together the reflector, the driver, and thedirectors. Compared to the above-described uniplanar Yagi an-tennas, this design is able to provide a number of advantages.First of all, the usage of the third dimension (the vertical dimen-sion) that has not been widely used in the design of microstrip

0018-926X/$26.00 2010 IEEE

KRAMER et al.: VERTICALLY MULTILAYER-STACKED YAGI ANTENNA WITH SINGLE AND DUAL POLARIZATIONS 1023

antennas, allows an effective reduction in size and footprint.In fact, multilayer processing techniques have become moremature in integrated circuit design, fabrication and integration.Second, a high permittivity substrate can be used in this case,thereby reducing spacing between the directors, which is criticalfor a high-density integration between antenna and circuits. Infact, the use of special tunable substrates and/or materials suchas ferrites or ferroelectric films allows a very easy realization ofhigh-sensible frequency- or bandwidth-agile antenna systems.Third, the possibility of a dual polarization design based on theYagi antenna concept is made possible, and the coupling-basedfeed mechanism can achieve wide bandwidth characteristics. Fi-nally, the most important advantage can probably be visualizedthrough the design of such antennas over millimeter-wave andterahertz ranges where the substrate spacing/thickness betweenYagi antenna elements can naturally be made compatible withcurrent three-dimensional circuit processing techniques such asthe state-of-the-art 3-D through silicon via (TSV) techniques.

Various stacked structures were investigated in [9] and [10].The advantage of those topologies is that substrate thickness canbe adjusted to achieve an optimized bandwidth performance.Unlike the Yagi antenna, which is a traveling-wave antenna [11],the structure is a resonant mode antenna. In the Yagi structure,the director is smaller than the driver and the distance betweenthem is between and . In [9] the thickness of theused substrate is about , the bandwidth is 70% widerthan the single patch without gain improvement and the upperlayer patch is bigger. Li studied theoretically different patchshapes (square, circular, triangle, etc.) and investigated the op-timal combination to achieve a lower cross polarization or cir-cular polarization [10]. The ratio of substrate thickness to wave-length in the dielectric is close to 0.02. Stacked triangular mi-crostrip antennas were also investigated experimentally in [12]to achieve a bandwidth about 17%-5% at the centre frequencyof 3.407 GHz.

To demonstrate the proposed concepts and design features,we will present two case studies in connection with the designof respective structures. The first design (see Fig. 1) is a multi-layered printed-circuit version of the proposed Yagi antenna. Itconsists of a ground plane (as a reflector), a dipole (as a driver),and four directors. In the second design (see Fig. 3), the antennapresents a dual polarization using circular patches in the designof one driver element and four directors.

In this paper, the configuration of the proposed Yagi-Udaantennas is described in detail. Design specifications are dis-cussed, considering the effects of different dimensions on an-tenna performances. Both of the proposed antennas as describedin Fig. 1 and Fig. 3 are fabricated. Simulated and measured re-sults are then compared, and the work is finally concluded.

II. ANTENNA DESIGN CONSIDERATION

A. Stacked Dipole Yagi Antenna (Design #1)The configuration of the dipole stacked Yagi antenna shown

in Fig. 1, is based on the classical Yagi-antenna design principle.It consists of one dipole driver element, and four parasitic ele-ments. The antenna is designed on the basis of the same design

Fig. 1. Proposed structure of design 1 (dipole stacked Yagi antenna).

Fig. 2. Layer II of design 1.

Fig. 3. Proposed structure of design 2 (dual polarization circular patches).

rules that are used in the conventional Yagi dipole antenna, ex-cept that the antenna is made on a planar substrate.

There have been many different design versions for printeddipole antennas and baluns, as well as coplanar strip dipoles


fed by coplanar waveguide and stripline balun used to feed aprinted quasi-Yagi antenna proposed in [5]. The proposed dipoleallows the avoidance of a balun which becomes usually neces-sary when dipole antennas are fed by an unbalanced line. Thedipole (layer II in Fig. 1) is printed on both sides of a dielectricsubstrate [13][15]. The bottom-tapered ground transition is de-signed to provide an impedance matching tuner with balancedoutput. The designed feed network of the dipole antenna withthe tapered balun transition is tuned by the angle of the taperedground plane for impedance matching and balanced output. Thissimplified feeding structure results in the reduction of transmis-sion line length and in turn radiation loss. Moreover, it exhibitsattractive wide bandwidth capabilities. The feeding dipole band-width (without any Yagi antenna elements) is relatively large; abandwidth of 29% can be achieved at 5.8 GHz using a 0.762mm thick substrate with a relative permittivity of 2.33 (RogersRT/Durod5870).

Director elements fabricated on substrates having a relativepermittivity of 3.48 results in significantly reduced spacing be-tween them. In the Yagi-antenna design, the gap between theelements is approximately given by

(1)

where is the relative dielectric constant of the substrate (theconstant of air in the classical Yagi-Uda antenna design) and

is the free space wavelength. This equation clearly showsthat if the dielectric constant is increased, the gap is reduced.In this design, the substrates should be stacked ideally withoutany air gaps. However, the layers are still very thick at 5.8 GHz,which are not feasible and affordable for fabrication. Of course,this is no longer a problem at very high frequencies such asmillimeter-wave ranges. To some extent, this class of antennasbecomes more attractive and easier-to-implement at lower costwhen the operating frequency becomes higher.

Antenna characteristics such as gain, front-to-back ratio,beamwidth and center frequency can be altered by changingthe length of the driven element, the length of the parasiticelements, the spacing between reflector and dipole, the spacingbetween director and dipole, the spacing between directors orsubstrate thickness as well as the dielectric constant. It is shownthat an array configuration is completely determined whenany two of these constraints are specified [16]. The proposedstacked dipole Yagi antenna is simulated by using AnsoftDesigner v2.0, a commercial simulator that can solve electricand magnetic fields via a method of moment.

A reflector plane is added to the driver element, the size andthe distance of the reflector are optimized. For different spacingbetween driver dipole and reflector, Fig. 4 shows the gain vari-ation versus the reflector plane size. It is found that the op-timum reflecting spacing for the maximum directivity is be-tween and as in case of the standard Yagi struc-ture. The gain increases as the size of reflector increases, and thisvariation becomes less pronounced beyond . The dimen-sional ratio of the reflector to the driven element can be some-where between 2 and 2.6. Compared to the standard Yagi an-tenna, this ratio is doubled. The variation of the conductivity of

Fig. 4. Gain variation versus ground plane size for different driver-to-reflectorspacing.

Fig. 5. Gain variation versus director length for different director-to-directorspacing.

the reflector material has also influence, leading to the degrada-tion of bandwidth and/or directivity as observed in [17].

Three directors are added to the dipole and reflector. Fig. 5shows the variation of the gain with respect to the director lengthin dielectric for different director-to-director spacing. In thissimulated result, the optimum director-to-director spacing is inthe order of to , compared with typical 0.2 to 0.35wavelengths in the design of a standard Yagi. The wavelengthin the substrate thickness must be added to obtain the actualspacing. On the other hand, the gain increases with length, andthe optimal length of director is around the dipole dimensionwhere the coupling is maximized. A further increase of theirsize should reduce the array gain rapidly. The dimensional ratioof the director to the driven element can be between 0.8 and0.95.

As shown in Fig. 6, the gain enhancement is not significantwhen the fourth director is added, compared to the Yagi antennawith three stacked elements. The addition of an identical fourthdirector would increase the gain only by 0.25 dB. Upper planarsubstrate with a high permittivity is added in order to reduce theantenna size (footprint) and to increase the gain [19], [20]. Fig. 6shows the effects of adding this substrate without directors on


Fig. 6. Gain variation as a function of director length for: staked Yagi withthree and four identical directors; Stacked Yagi with three directors and highpermittivity plan.

the antenna gain. An improvement of 2 dB can be achieved andthe optimal length of the directors becomes shorter.

The gain enhancement is given by the following equations[20]

(2)

in which

(3)

and

(4)

with B being the thickness of the lower layer and , areits relative permittivity and permeability, respectively. and

are the relative permittivity and permeability of the upperlayer. is the refractive index. is defined as thefrequency deviation parameter.

(5)

where frequency is near by the center frequency .Composite planar dielectric structures can be used for in-

creasing the directivity of a point source when a resonance con-dition is established. Particular attention has been given to thephysical interpretation of this resonance gain effect in [18]. Thisis described in terms of leaky waves (LW) excited in the struc-ture. Under certain resonance conditions, a pair of weakly atten-uated TE/TM leaky waves becomes the dominant contributionto the antenna aperture field. Equation (2) applied to our specificconfiguration yields an approximate enhancement of 0.9 dB.

A director is added on the upper substrate. Various parame-ters are optimized starting from the initial value defined in pre-vious paragraphs. The entire structure of design#1 is optimized

TABLE IDESIGN 1 FOR DIPOLE STACKED YAGI ANTENNA

TABLE IIDESIGN 2 FOR PATCH STACKED YAGI ANTENNA

in order to achieve high gain and large bandwidth at 5.8 GHz.Director lengths are perturbed to optimize the return loss withminimum gain loss. The distances between elements along thevertical axis are: (8.58 mm) between reflectorlayer I and driver layer II; (5.6 mm) between driverlayers II, and director layers III, IV and V.

Fig. 7 shows the influence of elements width on the band-width of the design #1. This design can achieve a bandwidthbetween 6% and 27% at 5.8 GHz. It was noticed that this param-eter has the most significant effect when it is selected between

and , and becomes much less influential outside thisband. The return loss is more sensitive to the width of the ele-ments (driver and director) rather than antenna directivity.

The total height of the designed structure is 29 mm. Com-pared with (1), it can be observed that the spacing between el-ements can be reduced by a factor of two. The upper directorlayer is in direct contact with director V. The change of relativepermittivity in the interface between the air gap and the substrateresults in a higher equivalent relative permittivity, thus reducingthe air gap. The size of the stacked substrates is .

B. Circular Patch Stacked Yagi Antenna With DualPolarization (Design #2)

In this design demonstration, the classical dipole is replacedby a patch antenna. The proposed concept offers a possibilityto use a wide range of patches instead of the classical dipoleas driver. The circular patch has been used as duo, namely adual polarization feeder, and an optimization facility, since thatis only required to optimize one dimension (radius).


Fig. 7. S11 as a function of element width.

The antenna consists of two substrate layers. The circularpatch is etched on the top substrate and the feed lines and groundare etched on the bottom substrate. To achieve the dual polar-ization the patch is fed by two orthogonal lines. Such a feedingby coupling allows increasing the bandwidth. This patch has abandwidth of 4.5% at 5.8 GHz. The feed layer (layer I in Fig. 3)is grounded on its bottom side and fed on the other side by 2.26mm width, 20.27 mm long strip lines. The feed linesstart from the middle of their respective side, and the width iscalculated to match the antenna to 50 impedance. The circularpatch has a radius of (9.68 mm) and the spacing betweenthe layers I and II is 0.28 mm. The antenna is simulated usingAnsoft HFSS v10, a simulator which makes use of the finite el-ement method.

The proposed circular patch stacked Yagi antenna is shownin Fig. 3. This antenna is designed with the geometry of severallayers, and is very similar to the dipole Yagi antenna topologythat was described in the above section. In this structure, theRogers RF/Duroid 5870 substrate is used for all the layers with

and thickness of 0.762 mm.As in the case of the first design, the antenna is optimized to

achieve better gain and larger bandwidth, but also the isolationcharacteristics between the two feed lines must be taken intoaccount.

One constraint of the stacked dipole Yagi antenna is relatedto its requirement for a large ground plane. Fig. 8 shows thegain enhancement of the patch antenna in connection with theground size, suggesting that the gain increases with the size ofthe ground plane. The optimal ratio of the reflector to the drivenelement is in the order of 1.6, which is comparable to the stan-dard Yagi antenna topology. This antenna can be used to reduceoverall dimensions of patch stacked Yagi antenna; a high gainwith reduced ground plane can be obtained.

Fig. 9 plots the gain versus the number of parasitic elementsfor adjacent director-to-director spacing of . Note thatadding the sixth director increases the gain only by 0.1 dB.Similar to the first design case, the gain performances aregenerally controlled by the director spacing and director length.

To define initial values, the array is optimized by varying theradius while maintaining the spacing parameters. Fig. 10 showsgain variation versus the element length on dielectric substratefor different director-to-director spacing. The ideal spacing is in

Fig. 8. Gain versus ground plane size.

Fig. 9. Gain versus the number of parasitic elements.

Fig. 10. Gain variation with director length for different director-to-directorspacing.

the order of . The ideal radius of the directors should bedefined between and . The ratio to the driver patchis the same as in the first design case.

Optimization of the element spacing is followed by the per-turbation of different radius starting from the initial value to op-


Fig. 11. Effect of diameter variation on the gain and on the isolation betweenthe two polarizations.

Fig. 12. Photograph of the first design example for dipole-based Yagi antennastructure.

timize the bandwidth with smaller gain losses. Maintaining theoptimized spacing constant, the reflector spacing radius of thefirst director (R2) can be used to tune the return loss characteris-tics of antenna as well as the isolation between the two polariza-tions. The effect of variation of the radius on the gain is verifiedalso. As shown in Fig. 11, the isolation between the two polar-izations has the excursion about 1015 dB for diameter variationin the order of . The effect of diameter variation on thegain is less pronounced as observed in Fig. 11.

The optimized patch radius for layers III, IV, V and VI are,respectively, , ,

, and .The size of the layers and the ground plane are .

Because of the effective permittivity of substrates that is lowerthan the previous configuration, the spacing between the direc-tors has increased accordingly. Hence, the same substrate hasbeen used throughout the antenna in order to simplify the fab-rication process. The vertical spacing between the directors isclose to 14.6 mm while the total height of this designedconfiguration is 60 mm.

III. RESULTS AND DISCUSSION

The fabricated prototypes of both antennas are shown inFigs. 12 and 13. The layers are aligned by using four threaded

Fig. 13. Photograph of the second design example for patch-based Yagi an-tenna structure.

Fig. 14. S-parameter characteristics of the prototyped antenna #1.

rods with nuts. With this assembly, the gaps between the layerscan be easily and precisely controlled.

A. The First Design Example1) Transmission Characteristics: S-parameters are mea-

sured using Anritsu 37397C Vector Network Analyzer to char-acterize the transmission properties of the proposed structureover the frequency range of interest. Simulated and measuredreturn losses versus frequency are presented in Fig. 14. It canbe seen that the measured center frequency is shifted slightlyfrom the designed target but still very close to 5.8 GHz. Thesimulation results obtained by HFSS show a good agreementwith the measured results. The antenna bandwidth ( 10 dB)covers frequencies from 5.4 to 6.2 GHz or almost 14% at 5.8GHz.

2) Radiation Pattern: Radiation pattern measurement ismade in a MI Technology anechoic chamber. Fig. 15 presentscalculated and measured co-polar (E plane) radiation patterns.From these results there is good agreement between the mea-sured and simulated radiation patterns.


Fig. 15. Radiation pattern performances of antenna #1 at 5.8 GHz.

Fig. 16. S-parameter characteristics of the prototyped antenna #2.

It is also observed that the radiation pattern is very directiveand symmetric. The 3 dB beam width is approximately 70 forboth measured and simulated patterns. The side lobes are verylow. The oscillation of the right lobe is due to the mechanicalconstraints of our anechoic chamber.

3) Antenna Gain: As shown in Fig. 11, the peak antennagain is about 12 dB based on our simulations and almost 11 dBcan be obtained in practice according to our measured results.Although the gain is slightly lower than expected, both curvesmatch very well. The 1 dB drop in measured gain is caused bythe imperfections of the compact range anechoic chamber usedand also by losses such as surfaces waves, dielectric losses, andconnector losses (coaxial lines).

The total height of the designed configuration is 29 mmcompared to the designed structure described in [8],

which has a dimension of and uses two more directorsto achieve lower gain. Based on equations detailed in [2], thehorn antenna is 125 mm in length with radiating aperture of 155by 22.15 mm in order to get a gain of 12 dB, and a coaxial towaveguide transition must be added in this case. It is clear that

Fig. 17. Radiation pattern performances of antenna #2 at 5.8 GHz.

the designed multilayer-stacked Yagi antenna is much morecompact than the metallic horn antenna.

B. The Second Design Example1) Transmission Characteristics: Measured and calculated

-parameters of the second antenna are shown in Fig. 16. Themeasured center frequency for both polarizations (Tx and Rx) isat 5.8 GHz and agrees well with the simulated results. The band-width is from 5.6 GHz to 5.85 GHz or 4% at 5.8 GHz, for bothpolarizations. The isolation (S21 parameter) between the twoports is nearly 30 dB over the entire frequency band, instead ofthe 25 dB calculated isolation. The bandwidth of this antennais narrower but sufficient for most LPS or wireless sensor ap-plications at 5.8 GHz. This bandwidth can also be increased byusing a wideband patch [21], [22]. The measured performancematches well the simulation prediction. A small difference be-tween the measured and simulated results (S11 is not exactly thesame as S22), can be attributed to the fabrication accuracy.

2) Radiation Pattern: Fig. 17 presents the simulated andmeasured co-polar (E plane) radiation patterns for the seconddesign. The measured radiation pattern agrees well with the sim-ulated ones. The radiation pattern is again very directive andsymmetric. The measured beam width (3 dB definition) is ap-proximately 60 degrees, compared to 70 degrees for the simu-lated pattern.

Both polarization curves are very similar (that is normal be-cause of the symmetry of this structure).

3) Antenna Gain: Fig. 17 shows the gain performance basedon both simulated and measured results. The respective max-imum gains for the simulation and the measurement are 11.66dB and 9.76 dB. There is a loss of 1.9 dB between the two gains.This loss is due to a frequency shift of the fabricated antenna (amaximum gain of 10.28 dB can be reached at the frequency of5.75 GHz), the fabrication tolerance and losses in the measure-ment circuits, in addition to the imperfections of the compactrange anechoic chamber.

To achieve a comparable gain, DeJean in [25] proposed abi-Yagi and quad-Yagi which consist of seven planar patch ele-ments. The size of the ground plane is compared to

the size of our demonstrated antenna.


IV. CONCLUSION

In this paper, two classes of novel antenna based on theclassic Yagi-Uda antenna concept, have been proposed anddemonstrated for the first time theoretically and experimentally.By using multilayer-stacked substrates, these designs allowcompact size realization and achieve good performance at thedemonstrated frequency of 5.8 GHz. Two different antennaconfigurations are presented and showcased, one based ondipole for single polarization, and the other on circular patchfor dual polarization. The characteristics of the two proposedantenna types with respect to various parameters such as re-flector dimensions, director dimensions, and spacing betweenthese elements have been studied. The measurement results ofS-parameters show a reasonably good bandwidth (a bandwidthof 15% can be reached with the novel structures) and themeasured radiation pattern performance is very similar to thesimulated results. Both designs have a peak gain of about 10dB. Compared to the microstrip array Yagi antenna, whichmakes use of planar patches, this novel design can yield highgain and also the entire structure is very compact in size byusing the vertical dimension. In this case, a large number ofdirectors can be implemented in such multilayered geome-tries. Based on such new design concepts, many innovativestructures with interesting properties, including electronicallytunable substrates, can be realized by using different planarpatches in the driver layer, or stacked substrate with zeroair gap (especially for high frequency applications such asmillimeter-wave or even terahertz frequency ranges). Thiswork suggests that the proposed concept provides light-weight,low-cost, high-performance and full integration solutions forlocal positioning platforms, wireless sensor systems and radarsensor applications. It can be expected that those new antennaswill provide a very attractive design alternative for a wide rangeof microwave and millimeter-wave system applications.

ACKNOWLEDGMENT

The authors wish to thank J. Gauthier and S. Dube of Poly-Grames Research Center, cole Polytechnique de Montreal, inthe fabrication and mounting of prototypes.

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[18] D. R. Jackson and A. A. Oliner, A leaky-wave analysis of high-gainprinted antenna configuration, Trans. Antennas Propag., vol. 36, no.7, pp. 905910, Jul. 1988.

[19] A. Hoorfar, Analysis of a Yagi-Like printed stacked dipole array forhigh-gain application, Microw. Opt. Technol. Lett., vol. 17, no. 5, pp.317381, Apr. 1998.

[20] D. R. Jackson and N. G. Alexopoulos, Gain enhancement methodsfor printed circuit antennas, Trans. Antennas Propag., vol. 33, pp.976987, Sep. 1985.

[21] S. K. Padhi, N. C. Karmakar, and C. L. Law, An EM coupled dual-polarized microstrip patch antenna for RFID applications, Microw.Opt. Technol. Lett., vol. 39, no. 5, pp. 354360, Dec. 2003.

[22] Y. M. M. Antar, D. Cheng, and G. Jiang, Wide-band microstrip patchantenna for personal communication, in Proc. 6th Int. Conf. on Elec-tronics, Circuits and Systems, Pafos, Cyprus, Sep. 1999, vol. 3, pp.13051308.

[23] P. R. Grajek, B. Schoenlinner, and G. M. Rebeiz, A 24 GHz high-gainYagi-Uda antenna array, Trans. Antennas Propag., vol. 52, no. 5, pp.12571261, May 2004.

[24] D. Gray, J. Lu, and D. V. Thiel, Electronically steerable Yagi-Udamicrostrip patch antenna array, Trans. Antennas Propag., vol. 46, pp.605608, May 1998.

[25] G. R. DeJean, T. T. Thai, S. Nikolaou, and M. M. Tentzeris, De-sign and analysis of microstrip Bi-Yagi and Quad-Yagi antenna arraysfor WLAN applications, Antennas Wireless Propag. Lett., vol. 6, pp.244248, Jun. 2007.

Olivier Kramer was born in Lyon, France, in 1984.He received the B.Sc. degree (with distinction) fromthe Ecole nationale suprieure en Systmes Avancset Rseaux of the Institut National Polytechniquede Grenoble (ESISAR-INPG), France, in 2007. Heis currently working toward the M.Sc.A. degree atEcole Polytechnique, Montreal, QC, Canada.

His current interests involve millimeter-wave mul-tilayer structures.


Tarek Djerafi was born in Constantine, Algeria,in 1975. He received the Dipl. Ing. degree fromthe Institut dAeronautique de Blida (IAB), Blida,Algeria, in 1998 and the M.A.Sc. degree in electricalengineering from Ecole polytechnique de Montreal,Montreal, QC, Canada, in 2005, where he is cur-rently working toward the Ph.D. degree.

His research deals with design of millimeter-waveantennas and smart antenna systems, microwaves,and RF components design.

Ke Wu (M87-SM92-F01) received B.Sc. degree(with distinction) in radio engineering from NanjingInstitute of Technology (now Southeast University),China, in 1982 and the D.E.A. and Ph.D. degrees inoptics, optoelectronics, and microwave engineering(with distinction) from the Institut National Poly-technique de Grenoble (INPG) and the University ofGrenoble, France, in 1984 and 1987, respectively.

He is a Professor of electrical engineering, andTier-I Canada Research Chair in RF and mil-limeter-wave engineering at Ecole Polytechnique

(University of Montreal). He also holds a number of Visiting (Guest) and Hon-orary Professorships at various universities including the first Cheung KongEndowed Chair Professorship at Southeast University, the First Sir Yue-KongPao Chair Professorship at Ningbo University, and honorary professorship atNanjing University of Science and Technology and City University of HongKong. He has been Director of the Poly-Grames Research Center and theFounding Director of Centre de recherche en lectronique radiofrquence(CREER) of Quebec. He has (co)-authored over 700 referred papers, a numberof books/book chapters and patents. His current research interests involvesubstrate integrated circuits (SICs), antenna arrays, advanced CAD andmodeling techniques, and development of low-cost RF and millimeter-wavetransceivers. He is also interested in the modeling and design of microwavephotonic circuits and systems. He serves on the Editorial Board of MicrowaveJournal, Microwave and Optical Technology Letters, and Wileys Encyclopediaof RF and Microwave Engineering. He is an Associate Editor of InternationalJournal of RF and Microwave Computer-Aided Engineering (RFMiCAE).

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 4, APRIL 2010 1031

A Compact Tri-Band Monopole AntennaWith Single-Cell Metamaterial Loading

Jiang Zhu, Student Member, IEEE, Marco A. Antoniades, Member, IEEE, and George V. Eleftheriades, Fellow, IEEE

AbstractA compact tri-band planar monopole antenna is pro-posed that employs reactive loading and a defected ground-planestructure. The reactive loading of the monopole is inspired bytransmission-line based metamaterials (TL-MTM), which enablesthe loaded antenna to operate in two modes. The first resonanceexhibits a dipolar mode over the lower WiFi band of 2.40 GHz 2.48 GHz, and the second resonance has a monopolar mode overthe 5.155.80 GHz upper WiFi band. Full-wave analysis showsthat the currents of the two modes are orthogonal to each other, re-sulting in orthogonal radiation patterns in the far field. The featureof a defected ground-plane, formed by appropriately cutting an-shaped slot out of one of the CPW ground-planes, leads to thethird resonance that covers the WiMAX band at 3.303.80 GHz.Air bridges at the intersection between the antenna and the CPWfeedline ensure a balanced current. A fabricated prototype hascompact dimensions of 20.0 mm 23.5 mm 1.59 mm, andexhibits good agreement between the measured and simulated

parameters and radiation patterns. The measured radiationefficiencies are 67.4% at 2.45 GHz, 86.3% at 3.50 GHz and 85.3%at 5.50 GHz.

Index TermsDefected ground plane, folded monopole antenna,metamaterials, multiband antenna.

I. INTRODUCTION

T WO commonly used protocols for Wireless Local AreaNetworks (WLANs) based on access points to relay data,are WiFi and WiMAX, which promise higher data rates and in-creased reliability. A challenge in designing such multiple wire-less communication protocol systems is to design compact, lowcost, multiband and broadband antennas.

The planar monopole is attractive for WLAN antenna designbecause it has a low profile, it can be etched on a single substrateand can provide the feature of broadband or multiband oper-ation. The traditional approach is to use multibranched stripsin order to achieve multiband operation [1], which generallyleads to a large volume or requires a large ground-plane. Alter-natively, the concept of the frequency-reconfigurable multibandantenna [2] has been proposed to develop multiband monopoleantennas for WiFi and WiMAX applications [3]. Such recon-figurable antennas are reported to have the advantages of beingable to switch to a desired service and to achieve good out-of

Manuscript received June 18, 2009; revised September 25, 2009; acceptedOctober 27, 2009. Date of publication January 26, 2010; date of current versionApril 07, 2010. The work of J. Zhu was supported by an IEEE Antenna andPropagation Society Graduate Fellowship.

The authors are with the Edward S. Rogers Sr. Department of Electrical andComputer Engineering, University of Toronto, Toronto, ON M5S 3G4, Canada(e-mail: [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.


Fig. 1. Tri-band monopole antenna with single-cell MTM loading and a de-fected ground-plane. All dimensions are in mm: , , , , , , , ,

, , , , , ,

and slot width . (a) Top view. (b) 3D schematic.

band noise rejection performance. However, this is traded offwith an increased design complexity and an increased fabrica-tion cost associated with switches and bias circuits.

Transmission-line metamaterials (TL-MTM) [4][6] providea conceptual route for implementing small resonant antennas[7][17]. TL-MTM structures operating at resonance were firstproposed in order to implement small printed antennas in [8]and [9]. Furthermore, a dual-band MTM-inspired small antennafor WiFi applications was shown in [18] and multiband MTMresonant antennas were shown to exhibit several left-handedmodes in [19]. However, typically TL-MTM antennas sufferfrom narrow bandwidths. Recently, [12] addressed the band-width problem by proposing a two-arm TL-MTM antenna res-onating at two closely spaced frequencies. Another method toenhance the bandwidth consists of merging two resonances to-gether in a TL-MTM printed monopole antenna [20].

In this work, a compact tri-band monopole antenna is pro-posed using reactive loading, that was inspired by previousTL-MTM work, and a defected ground-plane [21], in orderto meet the specifications of the WiFi bands (lower WiFi bandof 2.40 GHz 2.48 GHz and upper WiFi band of 5.15 GHz 5.80 GHz) and the WiMAX (3.30 GHz 3.80 GHz) band whilemaintaining a small form factor. Herein, we thoroughly explainthe operation of the proposed tri-band antenna and we fullycharacterize its performance both numerically and experimen-tally. As shown in Fig. 1, the proposed co-planar waveguide(CPW)-fed monopole antenna is loaded in a left-handedmanner, inspired by the NRI-TL metamaterial unit cell [11].

0018-926X/$26.00 2010 IEEE


Fig. 2. (a) Case 1: unloaded monopole antenna, (b) Case 2: dual-bandmonopole antenna with single-cell MTM loading and (c) Final: tri-bandmonopole antenna with single-cell MTM loading and a defected ground.The dimensions of these antennas are given in the caption of Fig. 1. (a) Case 1.(b) Case 2. (c) Final.

This loading consists of a single MTM cell which allows themonopole antenna to operate in two modes [20], covering boththe WiFi bands. The first is a folded monopole mode, where themonopole together with the single-cell MTM loading forms afolded monopole around the frequency of 5.5 GHz [11]; Thesecond mode is a dipole mode where the single-cell MTMloading forces horizontal currents to flow along the top edges ofthe ground-plane, thus rendering the entire ground a radiator ataround 2.5 GHz. It will be shown in Section II, that the currentscorresponding to the two modes are orthogonal to each other,as shown in Fig. 5. The third resonance covering the WiMAXband from 3.3 GHz to 3.8 GHz, is achieved by defecting theground-plane by cutting an -shaped slot from one side of theCPW ground. Air bridges are added at the antenna terminals toensure that only balanced currents flow on the CPW feedline.The resulting antenna is compact (including the ground-plane),completely uniplanar, low profile and via-free. Therefore, theproposed antenna is easy to fabricate using simple photolithog-raphy. A prototype antenna has been fabricated and tested. Themeasurements show good impedance matching at the WiFiand WiMAX bands, orthogonal pattern diversity in each of theWiFi bands and a reasonable radiation efficiency, all of whichmake it well suited for wireless LAN applications.

II. ANTENNA DESIGNThe antenna was designed on a low-cost FR4 substrate with

height mm, and . A rect-angular patch was chosen as the monopole radiation element.The length of the patch was adjusted according to the generaldesign guideline that the lowest resonance is determined whenthe length of the monopole, , is approximately . There-fore, an monopole results in the lowest res-onance occurring at 6.0 GHz, as can be seen in Fig. 4. Thisrefers to the initial design where the metamaterial-inspired reac-tive loading and the defected ground-plane are not employed,which is shown as Case 1 in Fig. 2. In order to compare the per-formances with the proposed tri-band antenna shown in Fig. 1,the size of the monopole element and the width of the groundfor Case 1 were kept the same as the proposed design. How-ever, the length of the ground in Case 1 was adjusted to 20 mmin order to achieve good impedance matching. The antenna wasfed by a CPW transmission-line, which can be easily integrated

Fig. 3. The equivalent circuit when the proposed tri-band antenna operates atthe monopole mode. The dimension refers to the size of the NRI-TL unit celland it corresponds to roughly the length of the monopole with reference toFig. 1(a).

Fig. 4. Simulated for Case 1: unloaded monopole antenna, Case 2:dual-band monopole antenna with single-cell MTM loading and Final: tri-bandmonopole antenna with single-cell MTM loading and a defected ground, asshown in Fig. 2.

with other CPW-based microwave circuits printed on the samesubstrate. The CPW feed was connected to the coaxial cablethrough a standard 50 SMA connector. All the structures weresimulated in the finite-element method (FEM) based full-wavesolver, Ansoft HFSS. The connector and coaxial cable were in-cluded in all simulations to characterize their effects on the an-tenna performance. Since the operating frequency of the initialdesign (unloaded monopole) was above the range of interest forexisting wireless LAN applications, different approaches usingsingle-cell MTM loading and a defected ground were pursuedto create the corresponding second and third resonances, at alower frequency range in order to meet the wireless LAN spec-ifications.

A. Single-Cell Metamaterial Reactive LoadingIn order to maintain the antennas small form-factor while

decreasing the operating frequency, the CPW monopole wasloaded with a single asymmetric negative-refractive-indextransmission-line (NRI-TL) metamaterial-based unit cell.The equivalent circuit for the antenna of Fig. 1 is shown in Fig. 3(at the folded monopole mode). The series capacitance, ,was formed between the monopole on the top of the substrateand the rectangular patch on the bottom of the substrate (seeFig. 1(b)). The MTM cell was asymmetrically loaded with twoshunt inductances, and . was formed by the inductivestrip at the base of the monopole, while was formed bythe thin inductive strip that joins the rectangular patch beneath

ZHU et al.: A COMPACT TRI-BAND MONOPOLE ANTENNA WITH SINGLE-CELL METAMATERIAL LOADING 1033

Fig. 5. HFSS-simulated surface current distribution on the conductors of thetri-band monopole antenna with single-cell MTM loading and a defectedground-plane at the resonant frequencies of (a) 5.80 GHz and (b) 2.44 GHz.(a) Folded monopole mode (5.80 GHz). (b) Dipole mode (2.44 GHz).

the monopole to the rectangular patch beneath the right-handground plane.

In order to simplify the fabrication, a via-free approach toimplement the asymmetric unit cell was used, which can berealized using low-cost thin-film technology. In order to achievea shunt RF short to ground at high frequencies, the capacitorwas used, which connects the shunt inductor to ground (seeFig. 1(b)). The capacitor was formed between the right-handground plane and the rectangular patch beneath it.

Case 2 in Fig. 2 refers to the dual-band monopole antennawith single-cell MTM loading. In Case 2, the feature of the

Fig. 6. Simulated of the equivalent dipole antenna representing the tri-band monopole antenna with single-cell MTM loading and a defected ground-plane when operating under the dipole mode, as shown in Fig. 5(b).

defected ground is temporarily removed, while the other ge-ometrical parameters remain the same as the tri-band antennashown in Fig. 1(a). From Fig. 4, it can be observed that themonopole antenna with this single unit-cell MTM loading ex-hibits dual-resonance characteristics [20]. The geometrical pa-rameters, namely the size of the square patches underneath themonopole patch and the length of the thin strip determine ,

and shown in Fig. 3, respectively. They were adjusted inorder to obtain in-phase currents along the top monopole andalong the thin bottom strip at the resonant frequency around 6.0GHz. When operating in the folded monopole mode, the an-tenna acts as a two-arm folded monopole along the -axis, sim-ilar to the four-arm folded monopole of [11]. As discussed in[11], it is possible to eliminate the odd-mode currents along theCPW feedline by adjusting the printed lumped elements, thusenabling the -directed even-mode currents on the antenna toradiate. This can be seen from the HFSS-simulated current dis-tribution of Fig. 5(a).

In addition to the monopole resonance at around 5.0 GHz 6.0 GHz, the metamaterial loading introduces a second reso-nance around 2.4 GHz 2.5 GHz, which is desired for WiFiapplications. At this frequency, the antenna no longer acts as afolded monopole along the -axis, but rather as a dipole orientedalong the -axis, as shown in Fig. 5(b), where the current pathwas sketched from HFSS. Since the currents along the right edgeof the left ground-plane section are flowing against the currentsalong the left edge of the right ground-plane section, only thein-phase currents along the top edges of both the ground-planesections contribute to the radiation, which renders the ground-plane as the main radiating element. The length of the currentpath , which is related to the size of the ground-plane,determines the resonant frequency. This is verified by the sim-ulation for a central-fed dipole shown in Fig. 6. In Fig. 6, theequivalent dipole was simulated with the same substrate and thesame length of . The resulting resonant frequencyis 2.42 GHz, which agrees with the dipole-mode resonance forthe simulated of the tri-band antenna shown in Fig. 4. Be-sides, since the dipole-mode currents for the proposed designare flowing in a meandered path, an even larger miniaturizationfactor is achieved, compared to the loaded monopole antenna re-ported in [20] where the dipole-mode currents only flow alongthe top edges of the ground-plane.


Fig. 7. HFSS-simulated current distribution at the intersection between the an-tenna terminal and the CPW feedline under the dipole mode of operation.

Ideally, the dipole mode would not be excited for the reg-ular monopole antenna due to the symmetric current distributionalong the central line of the CPW feedline. In order to excite thismode, a single unit cell of MTM reactive loading was utilized.At low frequencies, the host transmission-line sections are veryshort and can be considered negligible. Assuming that the feedis placed at the base of the shunt inductor , the entire circuit issimply transformed into a series resonator formed between theloading capacitors and and the loading inductors and

, as shown in Fig. 1(b). If the resonant frequency of the se-ries resonator is designed to overlap with the dipole mode reso-nance of the antenna, it forms a short circuit and therefore forcesthe in-phase currents along the -axis flowing from one side ofthe ground to the other, through the path shown in Fig. 5(b).This enables the ground-plane to radiate in a dipolar fashion,which is orthogonal to the radiation at the higher frequencies.Bearing in mind the design considerations discussed above, theoptimized loading patches have dimensions of 3.0 3.0and 2.5 2.5 , respectively, and the two sections of thinstrips have the same length of 5.5 mm and width of 0.25 mm. Inaddition, the position of the central line of the CPW inner con-ductor is tuned and placed 1.7 mm off from the central symmetryline of the antenna, in order to obtain a good impedance matchto 50 , which in turn results in asymmetric ground-planes, asshown in Fig. 1(a).

Since the currents need a path from one ground to the other,an air bridge made of copper wires was placed at the intersec-tion between the antenna terminal and the extended CPW feed-line, in order to provide a shorting path. Additional air bridgeswere placed at the CPW feed side and parallel to the first airbridge, which ensure balanced currents and preserve the CPWmode. Fig. 7 shows the transition of the currents from unbal-anced to balanced at the terminal, where one can observe thatthe majority of the unbalanced currents pass through the firstair bridge but after the third air bridge, they are effectively sup-pressed. It is worth mentioning that in practice, the CPW feedwith air bridges can be replaced with a conductor backed CPWwith a ground-via fence which offers the additional advantageof lower EMI radiation.

Lastly, it can been seen that there is a small dip around 3.6GHz for Case 2 shown in Fig. 4. This corresponds to another res-onant current path along the ground plane, which is nevertheless

weakly excited. The length of this current path is approximately, which can be verified using an equivalent dipole sim-

ulation, similar to that shown in Fig. 6.The dual-mode operation due to the metamaterial loading is,

indeed, verified by the HFSS simulation for Case 2 shown inFig. 2(b). As seen in Fig. 4, the simulated magnitude of forthe monopole antenna with single-cell MTM reactive loading(the dashed line) has a desired higher resonance at around 5.48GHz which is lower than 6.0 GHz for Case 1 and covers thebandwidth ( ) starting from 4.95 GHz to up to 7 GHz,while the lower resonance is centered at 2.46 GHz and has abandwidth of 90 MHz.

B. The Defected Ground-Plane Antenna

In order to create the third resonance to meet the require-ment for the WiMAX application (3.30 GHz 3.80 GHz) inthe responses of the monopole antenna with the proposed re-active loading (Case 2 in Fig. 2(b)), a slot was cut out of theantenna ground-plane, thus forming a defected ground-plane.Similar to [22], an -shaped slot was chosen to achieve a longereffective slot length, without having to increase the size of theground-plane. However, unlike having the slot cut at the topedge of the ground [22], the proposed design has the slot cutat the bottom edge, to avoid the discontinuity of the currentalong the top edge of the ground-plane, which contributes to thedipole-mode radiation. The width of the slot is whilethe vertical and horizontal length of the slot, and , wereadjusted in order to achieve a good impedance match throughoutthe WiMAX band. This leads to the final design topology asshown in Fig. 2(c). It can be seen from Fig. 4 (solid line) thatinserting the -shaped slot provides the third resonance around3.5 GHz for the WiMAX band, while the dual-mode operationfor the WiFi bands at around 2.5 GHz and 6.0 GHz is preserved.

The resonance due to the slot can be explained by observingthe surface current distribution on the conductors of the an-tenna, as shown in Fig. 8. As can be seen, there is a strongconcentration of the currents along the -shaped slot on theleft ground-plane. The slot forces the current to wrap aroundit and thus creates an alternate path for the current on the leftground-plane, whose length is approximately at its reso-nance. It is also noted from Fig. 8 that the -shaped slot does notsignificantly affect the balanced CPW mode, since it is placedfar enough away from the CPW. Even if there were a minimalamount of unbalanced current, it would be eliminated by the airbridges applied at the intersection between the antenna and theextended CPW feedline, as shown in Fig. 7.

Fig. 9 shows the simulated magnitude of for a parametricstudy of the length of the horizontal slot . It can be observedthat the horizontal cut has a large influence on exciting the slotmode. When , which refers to the case that there isonly a vertical slot cut from the bottom, the slot mode is barelyexcited, compared with the simulated parameter characteris-tics in the case without a defected ground (Case 2) in Fig. 4.Moreover, it can be seen that the vertical slot cut from the bottomdoesnt affect the folded monopole and dipole modes. As isgradually increased, a better impedance match is achieved over


Fig. 8. HFSS-simulated surface current distribution on the conductors of thetri-band monopole antenna with single-cell MTM loading and a defectedground-plane at the resonant frequency of 3.76 GHz.

Fig. 9. Simulated of the tri-band monopole antenna with single-cellMTM loading and a defected ground-plane for the different lengths of thehorizontal slot .

the WiMAX band, while the performances of the monopole anddipole modes are both preserved.

The final design of the tri-band monopole antenna withsingle-cell MTM reactive loading and a defected groundis shown in Fig. 1. The geometrical parameters were deter-mined based on the previous discussion and were fine-tunedin order to meet the WiFi and WiMAX bands require-ments. As shown in Fig. 1, the full size of the tri-bandmonopole antenna (including the ground-plane with the sizeof ) is 20.0 mm 23.5mm 1.59 mm, or , with respect tothe lowest resonant frequency of 2.45 GHz. The compact sizeand the tri-band performance of the antenna make it a goodcandidate for emerging WLAN applications.

III. SIMULATION AND EXPERIMENTAL RESULTSThe tri-band monopole antenna was fabricated and tested.

The fabricated prototype is shown in Fig. 10, and the measuredversus the simulated magnitude of from HFSS are shown inFig. 11. The antenna exhibits a simulated bandwidth of80 MHz for the lower WiFi band from 2.40 GHz to 2.48 GHz

Fig. 10. The fabricated prototype of the tri-band monopole antenna with single-cell MTM loading and a defected ground-plane. (a) Front side. (b) Back side.

Fig. 11. Measured and HFSS simulated for the proposed tri-bandmonopole antenna with single-cell MTM loading and a defected ground.

and a bandwidth from 5.13 GHz to beyond 7 GHz for the higherWiFi band. It also exhibits a bandwidth of 590 MHzfor the WiMAX band from 3.30 GHz to 3.89 GHz. The perfor-mances beyond 7 GHz are out of the scope of our interest forWiFi and WiMAX applications. The measured band-width is 90 MHz from 2.42 GHz to 2.51 GHz for the lower WiFiband, and from 5.20 GHz to beyond 7 GHz for the second WiFiband. The measured bandwidth for the WiMAX band is 620MHz from 3.35 GHz to 3.97 GHz. The simulated results andthe measured results show good agreement.

The simulated and measured radiation patterns for theproposed tri-band monopole antenna with single-cell MTMreactive loading and a defected ground-plane are plotted inFigs. 1214 for the three principle planes at the frequenciesof 5.50 GHz, 2.45 GHz and 3.50 GHz, respectively, wheregood agreement between the simulations and measurementscan be observed. Fig. 12 shows the radiation patterns at 5.50GHz for the -plane ( -plane and -plane) and the -plane( -plane). The fact that the antenna exhibits radiation patternswith a horizontal -directed linear E-field polarization, verifiesthat the antenna operates in a folded monopole mode around5.50 GHz, due to the -directed in-phase currents along themonopole and the thin vertical inductive strip, as shown inFig. 5(a). The -directed currents along the thin horizontalinductive strip have a contribution to the cross polarizationin the -plane. It can also be seen that at this frequency, theslot on the left ground has a minimum contribution to theradiation since the currents are dominated by the -directed


Fig. 12. Measured and simulated radiation patterns for the tri-band monopole antenna with single-cell MTM loading and a defected ground-plane at 5.50 GHz.Solid line: measured co-polarization, dashed black line: simulated co-polarization, solid blue line: measured cross-polarization, and dash-dot black line: simulatedcross-polarization. (a) -plane. (b) -plane. (c) -plane.



ones along the monopole. At this frequency, the simulatedradiation efficiency is 85.9%, which is in good agreementwith the measured efficiency of 85.3%, using the Wheeler Capmethod [23]. The Wheeler cap measurements were conductedaccording to the method described in [23], where the measured

data in free-space and within the Wheeler cap were rotated onthe Smith chart in order to obtain purely real values for the inputresistance at resonance. The sphere used in the measurementsis shown in Fig. 8 of [11], together with the pertinent details ofits size. For this size Wheeler cap, its resonant frequency was


TABLE ISIMULATED AND MEASURED GAIN AND RADIATION EFFICIENCY FOR THE TRI-BAND MONOPOLE ANTENNA

WITH SINGLE-CELL MTM LOADING AND A DEFECTED GROUND-PLANE

calculated to be 0.69 GHz, which is well below the operatingrange of the tri-band antenna. Therefore, during the efficiencymeasurements at the three distinct resonant frequencies, specialattention was paid in order to avoid any of the cavity resonancesby slightly re-adjusting the position of the antenna within theWheeler cap. Since the antenna was completely enclosed by theWheeler-cap sphere during the measurements, this eliminatedany potential radiation from the feed cable.

At 2.45 GHz, the -directed currents contribute to the radi-ation, as shown in Fig. 5(b). This is consistent with the radia-tion patterns measured at the same frequency shown in Fig. 13.The radiation patterns in the -plane and the -plane, whichcorrespond to the two -planes of the ground-plane radiatingmode, indicate that the structure radiates in a dipolar fashionat this frequency. Similar to [20], there is, however, a partialfilling of the null around 90 in the -plane, that can be at-tributed to constructive interference from the -directed currentsalong the CPW and the vertical thin inductive strip, and also the

-directed current around the slot on the left ground-plane. Thisadditional current also manifests itself in the cross-polarizationdata of the -plane in the direction, as can be seen inFig. 13(b). In the -plane, which corresponds to the -plane ofthe radiating ground-plane, the radiation pattern is as expectedomnidirectional. Therefore, at 2.45 GHz, the antenna exhibitsa -directed linear -field polarized radiation pattern. It is or-thogonal to the one observed at 5.50 GHz, which verifies thein-phase -directed currents across the ground. The simulatedand measured efficiencies are 69.8% and 67.4%, respectively,which show good agreement.

Fig. 14 shows the radiation patterns at 3.50 GHz. Since the-shaped slot which is cut out of the left ground-plane results

in meandered currents along both the -direction and the -di-rection, which have independent contributions to the radiation,it is observed from Fig. 14 that the antenna exhibits two linearelectric fields that are orthogonally polarized in both the and

directions. Additionally, the axial ratio attains values close to,but greater than, one around the broadside direction, in-dicating circular polarization behavior. The measured efficiencyat this frequency is 86.3%, compared to a simulated efficiencyof 88.2% at the same frequency.

The measured and simulated gain and radiation efficiencyvalues, at the frequencies of 2.45, 3.50, and 5.50 GHz, are sum-marized in Table I.

IV. CONCLUSIONA tri-band and compact monopole antenna is proposed, that

can be used for WiFi and WiMAX applications. The antennaconsists of a regular CPW-fed printed monopole antenna withthe embedded features of metamaterial-based single-cell reac-tive loading and a defected ground-plane, which introduce an-

other two resonances at the lower frequencies, in addition tothe monopole resonance. The theoretical performance is verifiedby full-wave simulations and experimental data. The fabricatedprototype with a size of provides a90 MHz ( ) bandwidth from 2.42 GHz to 2.51 GHz forthe IEEE 802.11b/g/n standard (lower WiFi band) and a broadband from 5.20 GHz to beyond 7 GHz for the IEEE 802.11a/nstandard (upper WiFi band), and also a bandwidth of 620 MHzfrom 3.35 GHz to 3.97 GHz for the WiMAX band. The an-tenna exhibits dipole-like and monopole-like radiation patternswithin the lower and upper WiFi bands, respectively, which areorthogonal to each other. The radiation patterns at the WiMAXband exhibit two orthogonal linear E-field polarizations as ex-pected. Reasonable radiation efficiencies, in the range of 70% 90%, are obtained for all three bands.

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