1872 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 40, NO. 9, SEPTEMBER 2005

A CMOS Gyrator Low-IF Filter for a Dual-ModeBluetooth/ZigBee Transceiver

Brian Guthrie, John Hughes, Tony Sayers, and Adrian Spencer

AbstractA low-IF polyphase channel filter for a dual-modeBluetooth/Zigbee transceiver is described. Implemented in astandard 0.18- m CMOS process, the filter has a fifth-order0.5-dB equiripple bandpass response and employs novel transcon-ductor and preamplifier designs. It consumes 1 mW andachieves image band rejection 44 dB, input referred noise of52 2 Vrms and input referred third-order intermodulation

intercept of 20 dBVp, which gives a spurious-free dynamic rangeof 68.4 dB. Chip area including its tuning circuit is 0.23 mm2.

Index TermsChannel filter, CMOS, - , transceiver, gy-rator.

I. INTRODUCTION

THIS PAPER describes a dual-mode filter intended to meetspecifications for the Bluetooth [1] and ZigBee [2] sys-tems. The circuit should enable ZigBee capability to be addedto mobile phones, where Bluetooth is already becoming an es-tablished feature. Bluetooth applications are well known andinclude wireless headsets, file sharing, and printing. ZigBee isbased on the 802.15.4 specification [2] recently ratified by theIEEE and is used for very simple wireless connectivity. The ad-dition of ZigBee capability to a phone could enable the controlof devices such as lights, electronic devices and central heatingusing the mobile handset.

A dual-mode solution is being developed because, in theprice-sensitive market for mobile devices, it is essential that thiscapability be added for minimal extra cost. It is also unlikelythat handset manufacturers would welcome a third radio trans-ceiver in the handset (in addition to the mobile and Bluetoothtransceivers). Fig. 1 shows the context of the dual-mode filter.The signal from the antenna is amplified and split into andpaths, followed by downconversion and filtering. The andsignals are fed into two demodulators, only one of which willbe operating at a time. As ZigBee operates in the same radioband as Bluetooth, it is envisaged that the addition of this ca-pability would require very little change to LNAs, mixers, andsynthesizers designed for single-mode Bluetooth. A ZigBeedemodulator can be added at little extra cost. However, thesignal bandwidth for ZigBee is twice that of Bluetooth and thusa dual-mode channel filter is required which, for consistencywith single-mode Bluetooth, should be a low-IF filter. The filter

Manuscript received December 3, 2004; revised February 24, 2005.B. Guthrie is with Lifespan Scotland Ltd., Inverness, Scotland, IV2 3ED U.K.J. Hughes and T. Sayers are with Philips Research Laboratories, Redhill RH1

5HA, U.K.A. Spencer is with TRL Technology, Tewkesbury, Gloucestershire, GL20

8ND U.K.Digital Object Identifier 10.1109/JSSC.2005.848146

Fig. 1. Dual-mode receiver architecture.

to be described is switchable between a center frequency of1 MHz with a bandwidth of 1.2 MHz in the Bluetooth modeand a center frequency of 2 MHz with a bandwidth of 2.4 MHzin the ZigBee mode.

In Section II, we review the principles of complex filters anddescribe the filter synthesis procedure. This is followed in Sec-tion III by a description of the practical filter and the develop-ment of the circuit blocks that it employs. Measured results arepresented in Section IV, and the conclusion is given in Section V.

II. FILTER SYNTHESIS

The function of the channel filter in a low-IF receiver is topass the selected channel while rejecting neighboring channelinterferers. As these interferers occur at frequencies on eitherside of the passband, image responses must be suppressed,and this requires using a bandpass filter with an asymmetricamplitude response i.e., . Filters of thistype require complex coefficients, and these can be created bypolyphase networks employing two paths driven by signalswhich are identical but in exact phase quadrature as supplied inthe down-converted - and -channels of a low-IF transceiver.

A. Complex FiltersThe principle of the complex filter is illustrated in Fig. 2.

Starting with a real low-pass filter (i.e., with real coefficients),the transformation is applied. This shifts the polesup the imaginary axis by and transforms the low-pass re-sponse (actually a bandpass response centered at ) intoan identical bandpass response centered at [3]. Thetransformation preserves both amplitude and phase characteris-tics and produces the required feature of having no image re-sponse at negative frequency.

Synthesis of complex filters follows similar procedures tothose for real filters except that it makes use of complex inte-grators such as those shown in Fig. 3. The real integrator has an

0018-9200/$20.00 2005 IEEE

GUTHRIE et al.: CMOS GYRATOR LOW-IF FILTER FOR A DUAL-MODE BLUETOOTH/ZIGBEE TRANSCEIVER 1873

Fig. 2. Complex filter basics.

Fig. 3. Current-mode integrators. (a) Real. (b) Complex.

input , an output , and a transfer characteristic describedby

(1)

where is the integrator time constant.The complex integrator has two inputs and and two

outputs and . The cross-branch transconductorsand produce feedback currents which are proportional tothe outputs and which sum with the input currents. It is readilyshown that the transfer characteristic is described by

(2)

where the frequency shift is given by

(3)

This demonstrates that the transformation isbeing performed as required by the complex integrator.

B. Complex Bandpass SynthesisThe synthesis starts with the low-pass LCR prototype shown

in Fig. 4(a). Using traditional state-variable methods, theleapfrog filter structure shown in Fig. 4(b) is constructed that

Fig. 4. Channel filter architecture. (a) Fifth-order low-pass prototype.(b) Gyrator low-pass filter. (c) Gyrator complex bandpass filter.

simulates the nodal equations of the prototype using state-vari-ables , , , , and . The specific real current integratorsare shown boxed. It can be seen that the forward and backwardconnection of the integrators have formed a series oftransconductor loops which are in fact gyrators.

Translating this low-pass design into its complex bandpasscounterpart involves providing two paths, each containing thelow-pass filter, and then replacing the real integrator pairs withcomplex integrators. This results in the architecture shown inFig. 4(c). It can be seen that, as well as the transcon-ductor loops, the cross-branch transconductor loops form a fur-ther set of gyrators.

The design values for a fifth-order 0.5-dB equiripple Cheby-chev dual-mode filter with Bluetooth ( MHz,

MHz) and ZigBee ( MHz, MHz) re-sponses are given in Table I. The design uses a common set oftransconductors, and the dual response is implemented simplyby switching the values of the capacitors.

III. PRACTICAL FILTER DESIGN

The practical implementation of this filter uses balancedCMOS transconductors and floating antiparallel pairs ofNwell-polysilicon capacitor arrays with the extra capacitors for

1874 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 40, NO. 9, SEPTEMBER 2005

TABLE ISINGLE-ENDED DESIGN VALUES FOR THE DUAL-MODE CHANNEL FILTER

Fig. 5. Practical channel filter.

TABLE IICHANNEL FILTER REQUIREMENTS

Bluetooth switched in using simple nMOS switches. Note thatthe design values of its balanced transconductors and floatingcapacitors are half of those of the single-ended design given inTable I.

The practical filter is shown in Fig. 5. The required passbandvoltage gain of 20 dB is achieved by adding CMOS ampli-fiers before and after the filter block. Splitting the gain in thisway enables the intermodulation and noise performances to betraded optimally. The automatic tuning control reduces the fre-quency response spreads which result from variations in pro-cessing, temperature, and supply voltage to the required valueof 6%. The requirements for the channel filter are summa-rized in Table II.

In the rest of this section, we look in detail at the principalcircuit blocks used by the practical filter; the balanced transcon-ductor, the pre- and post-amplifiers, and the frequency tuningcontrol.

Fig. 6. Class-AB CMOS transconductors [6]. (a) Single-ended. (b)Differential. (c) Balanced.

A. Balanced TransconductorOne promising approach which uses the PMOS/NMOS tran-

sistor pair proposed by Nauta [6] for mainly high-frequencyfilters and employed recently by Andreani [7] at lower fre-quencies is shown in Fig. 6(a). If the pMOS and nMOStransistors and have identical parameters (this is an un-necessary constraint but one which simplifies the description),then and the overall transconductance ofthis single-ended cell is where . Biasing theinput at the midrail voltage produces equal draincurrents in both and and the output current is zero.When the input voltage changes by , the drain currents of

and are unbalanced as given by

(4)

(5)

and a linearly related current, , flows atthe output. Thus, the transconductor is inherently linear despitethe square law relationships determining the individual currents.Furthermore, the transconductor is exceptionally efficient be-cause it operates in class AB for peak output currents as high as

.

GUTHRIE et al.: CMOS GYRATOR LOW-IF FILTER FOR A DUAL-MODE BLUETOOTH/ZIGBEE TRANSCEIVER 1875

One feature of this transconductor is that is influencedby the bias current which is strongly determined by thesupply voltage , and this allows a very simple means fortuning the filter. It also carries a penalty for power supply noisefeedthrough. Assuming square-law saturated MOS behavior,the transconductance is given by

(6)

where . If now the value of is modu-lated by a signal , then is also modulated

(7)

and the output current is given by

(8)(9)

So, in addition to the wanted output signal , there isan extraneous signal which results in intermodulationof signals with supply noise. Consequently, great care must betaken when designing the tuning control circuit so that bothand noise are effectively suppressed.

Our balanced filter requires balanced transconductors sothat inversion of signals can be achieved by simply crossingover signal pairs. If one attempts to form a balanced arrange-ment from two single-ended transconductor cells, as shownin Fig. 6(b), it is found that, when used in feedback networkssuch as the transconductor loops occuring in a gyrator filter,the circuits become unstable. This can be resolved with thearrangement shown in Fig. 6(c) [6]. It comprises two mainsingle-ended transconductors and a common-modefeedback network between the input ports. This employs fourhalf-size single-ended transconductors ( , each using halfwidth transistors and half the bias current) coupled between theinputs of the main transconductors. It sets the common-modeinput voltage to correctly bias the transconductor transistorsand presents a high impedance to differential input signals.

One problem with using this transconductor in gyrator cir-cuits arises because of feedthrough from input to output via thegatedrain capacitances. Had these capacitances been reciprocal(i.e., ), then balanced gyrators (see Fig. 7) wouldhave equal feedforward and feedback capacitances producingno resultant feedthrough. Unfortunately, the intrinsic capaci-tances of a MOS transistor in saturation are nonreciprocal [8], asshown in Fig. 8. As a saturated MOS transistor is pinched-offat its drain, voltage disturbances at the drain do not influencethe channel charge (apart from the negligible contribution fromthe extrinsic capacitance) and so . On the other hand,voltage disturbances at the gate directly influence the channelcharge (to produce a change in the drain current) and so

. In a gyrator loop, the strong feedfor-ward via is not balanced by an equal feedback via , andthis can produce filter responses which peak at high frequency.

The solution to this is shown in Fig. 9. The pMOS capaci-tors are connected between each input and a source follower

Fig. 7. Balanced gyrator loop of two transconductors with feedthroughcapacitance (common-mode networks not shown for clarity).

Fig. 8. Intrinsic capacitances of a MOS transistor in saturation [8].

Fig. 9. Modified transconductor with capacitive feedthrough equalization.

connected to each output. Voltage disturbances at the transcon-ductor input produce feedforward via but the capacitive cur-rent is routed harmlessly via the source follower to , andthe transconductor experiences no extra feedforward. However,voltage disturbances at the transconductor outputs do producefeedback via . Clearly, creates only capacitive feedbackwhile the internal transconductors produce only feedforward. Ifwe make , thenthe fully differential transconductor has reciprocal feedthroughcapacitance. Now, when the transconductors are connected asgyrators, the feedforward cancels the feedback and feedthroughis eliminated. In practice, the values of and for thetransconductor are readily obtained from the transistors oper-ating point information and the resulting design of can betrimmed to optimize the filters amplitude response.

B. Post- and Pre-AmplifierA CMOS voltage gain stage can be made simply from a

pair of single-ended transconductors driving a load made from

1876 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 40, NO. 9, SEPTEMBER 2005

Fig. 10. Amplifier circuit. (a) Post-amplifier. (b) Pre-amplifier.

a pair of diode-connected single-ended transconductors. Thepost-amplifier [Fig. 10(a)] is implemented this way, and nocommon-mode network is required because it is not used ina feedback loop and its inputs receive a common-mode biasfrom the filter output. However, this is not a good solutionfor the pre-amplifier because it is required to block dc inputsfrom the mixer, and adding a common-mode circuit to bias thepre-amplifier would be a poor choice for two reasons. First,the common-mode arrangement produces unacceptable noise.Second, because random mismatches between common-modetransconductors would produce randomly varying / pathdelay mismatch, which would result in poor image rejection. Theproblem can be avoided by using the pre-amplifier arrangementshown in Fig. 10(b). The common-mode feedback is producedby the resistors and (in practice, nMOS transistorsoperating in the triode region) which produce an effectivefeedback resistance for differential signals of .For M and M , the effective feedbackresistor is 11 M . The arrangement produces minimal noiseand no random path delay mismatch, and furthermore theincreased effective resistance of the feedback network permitsthe use of smaller input blocking capacitors.

C. Frequency Response ControlThe chosen center frequency and bandwidth for ZigBee are

exactly twice those for Bluetooth. This makes switching be-tween the modes a very simple matter. Table I shows that wehave chosen to do this by switching the filters nodal capac-itances rather than the transconductance values, which wouldhave changed the filters power consumption. We have chosen touse floating capacitors, even though this adds an inevitable con-tribution of parasitic capacitance, because they occupy a quarterof the area of the grounded capacitor alternative. Each of thesefloating capacitors is made from two arrays of poly-Nwell unitcapacitor pairs connected in anti-parallel. One set of these twoarrays is connected directly to the filters nodes and produces the

Fig. 11. Tuning control circuit.

ZigBee response; the other set is connected in parallel via simplenMOS switches to produce the Bluetooth response. During filterdesign, the arrays must be trimmed to the filters design values,making allowance for the N-well-substrate capacitance and thetransconductor input capacitance. In practice, these parasiticscan be kept to a few percent of the required nodal capacitances.

The filters transconductor and capacitor values, and hencethe frequency reponse, are subject to spreads due to processvariation, temperature change, and aging. These spreads maybe tuned out by making a global change to the transconductorset by an adjustment to the regulated supply voltage , andthis is the basis of the automatic tuning control used in this de-sign. The usual approach using a phase-locked loop [9] was notused here because the required tuning accuracy was only 6%,and so a simpler arrangement was adopted to save power con-sumption and chip area.

The tuning control loop used in this design is amasterslave arrangement in which the master tuning loopadjusts the value of a reference transconductor, by adjusting itssupply voltage, to achieve a pre-defined time constant equal to

with a reference switched capacitor operated at a clockfrequency . The filter, i.e., the slave, has transconductorsand capacitors with similar design and supply voltage to thoseof the control loop and the close component tracking producesa filter with well-defined pole frequencies.

In more detail, the control loop [Fig. 11(a)] comprises askewed diode-connected transconductor that generatesa quiescent voltage that is offset from that of the refer-ence transconductor, a parallel arrangement of the referencetransconductor and switched capacitor , an integratorcomprising and , and an inverter and a chargepump [Fig. 11(b)] for regulating the supply. Inoperation, the reference transconductor and switched capacitorsupply the integrator with opposite polarity currents and ,

GUTHRIE et al.: CMOS GYRATOR LOW-IF FILTER FOR A DUAL-MODE BLUETOOTH/ZIGBEE TRANSCEIVER 1877

Fig. 12. Measurement setup.

Fig. 13. Measured amplitude response of filter. (a) Bluetooth. (b) ZigBee).

which are integrated and amplified and then used to control thecharge pump [10]. With the switched capacitor set to the value

, the loop stabilizes when the integrated currents sum tozero and is at the value required to achieve the desiredtime constant. When the transconductor or capacitor have non-typical values, adjusts to change the transconductancevalue to restore the time constant to its nominal value.

Necessarily, the loop regulates with a sawtooth ripple onand so the supply voltage used by the filter is

low-pass filtered by and . With the source followerregulating transistors and scaled to produce the same

, the value of tracks closely the mean value of ,and the filters transconductors will closely track the controlloops reference transconductor. The signal handled by thefilter produces almost no ripple current in the supply,but the mean level of the supply current increases with signalamplitude, and this causes a mismatch with and somedetuning of the filter.

The decoupling capacitors , , , and were allnMOS gate-oxide capacitors connected to minimize their gate

Fig. 14. Measured passband response (Bluetooth) with and withoutfeedthrough equalization.

Fig. 15. Measured compression characteristic (Bluetooth).

voltages. They also serve the purpose of transmitting any sub-strate interference onto the rail. With near equal interfer-ence on both and rails, very little signal is coupled intothe filter.

IV. MEASUREMENT RESULTS

The circuits were designed in a 1.8-V, 0.18- m digital CMOSprocess and measured under the control of LabVIEW software(Fig. 12).

With V and V, the typical measuredamplitude responses for Bluetooth and ZigBee are shown inFig. 13(a) and (b). Clearly, the passband responses are nearlyideal and the image-band rejection at is dB.The zero at dc is produced by the pre-amplifier blockingcapacitors. With the feedthrough equalization disabled, thepassband response (for Bluetooth) is as shown in Fig. 14 withexcessive high-frequency peaking with original transcon-ductors. This demonstrates that the modified transconductorusing feedthrough equalization is performing very effectively.The Bluetooth signal compression characteristic (similar forboth modes) is shown in Fig. 15 and indicates linear gainup to an input amplitude of 25 dBVp differential and aninput-referred 1-dB compression point of 23 dBVp (71 mVp)differential. The Bluetooth output differential noise responseis shown in Fig. 16 and, when integrated over the passband,gives an input-referred rms noise voltage of 50.4 V and asignal-to-noise ratio (SNR) of 60 dB. The ZigBee input-referredrms noise voltage is 52.2 V and its SNR is 59.6 dB. Fig. 17shows the Bluetooth third-order intermodulation characteristicfor input tones of 4 and 7 MHz, indicating an input-referredthird-order intercept (IIP3) of 24 dBVp differential. The ZigBeeIIP3 was 20 dBVp. The spurious-free dynamic range (definedhere as for Bluetooth

1878 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 40, NO. 9, SEPTEMBER 2005

Fig. 16. Measured noise response of filter (Bluetooth).

Fig. 17. Measured third-order intermodulation characteristic (Bluetooth).

TABLE IIIMEASURED PERFORMANCE OF A TYPICAL CHANNEL FILTER

is 71.2 dB and for ZigBee is 68.4 dB. The common-moderejection at the center of the passband for Bluetooth was 29 dBand for ZigBee was 35 dB. The power supply rejection forboth Bluetooth and ZigBee was greater than 50 dB. The typicalperformance for both modes is summarized in Table III. Mea-surement of ten samples gave a tuning error of less than 3%with a supply voltage as low as 1.5 V and an output offsetvoltage of less than 20 mV.

V. CONCLUSION

This paper has described a CMOS continuous-time filtertechnique giving both high performance and low power con-sumption. It was used to implement a complex bandpasschannel filter meeting all of the requirements for a dual-modeBluetooth/ZigBee receiver. Its high performance was due inparticular to the use of CMOS class AB transconductors, suit-ably modified to prevent capacitive feedthrough, which resultedin well-controlled passband responses, low intermodulation,and low power consumption. It was also due to the use of anovel pre-amplifier which gave high image-band rejection andlow noise. The channel filters power consumption of 1 mWis so low as to be insignificant when compared with that ofits companion RF circuits. The technique may be applied to awide range of radio systems, either with lower performance forultralow-power radio or with higher performance [11] wherelower power consumption might be of benefit.

REFERENCES[1] L. Harte, Introduction to Bluetooth: Technology, Market, Operation,

Profiles, and Services. Fuquay Varina, NC: Althos, Jan. 2004.[2] J. A. Gutierrez, E. Callaway, and R. Barnett, Low Rate Wireless Personal

NetworksEnabling Wireless Sensors With IEEE 802.15.4. NewYork: IEEE Press, 2003.

[3] A. S. Sedra, W. M. Snelgrove, and R. Allen, Complex analog bandpassfilters designed by linearly shifting real lowpass prototypes, in Proc.IEEE Int. Symp. Circuits and Systems, 1985, pp. 12231226.

[4] J. B. Hughes, A. Spencer, A. Worapishet, and R. Sitdhikorn, 1 mWCMOS polyphase channel filter for Bluetooth, Proc. Inst. Elect. Eng.-Circuits Devices Syst., vol. 149, no. 516, pp. 348354, Oct./Dec. 2002.

[5] K. W. Moulding, J. R. Quartley, P. J. Rankin, R. S. Thomson, and G. A.Wilson, Gyrator video IC with automatic tuning, IEEE J. Solid-StateCircuits, vol. 15, no. 6, pp. 963968, Dec. 1980.

[6] B. Nauta and E. Seevinck, Linear CMOS transconductance element forVHF filters, Electron. Lett., vol. 25, no. 7, pp. 448450, Mar. 1989.

[7] P. Andreani and S. Mattisson, On the use of Nautas transconductorin low-frequency CMOS Gm-C bandpass filters, IEEE J. Solid-StateCircuits, vol. 37, no. 2, pp. 114124, Feb. 2002.

[8] Y. P. Tsividis, Operation and Modeling of the MOS Transistor. NewYork: McGraw-Hill, pp. 370372.

[9] B. Shi, W. Shan, and P. Andreani, A 57-dB image band rejection CMOSGm-C polyphase filter with automatic frequency tuning for Bluetooth,in Proc. IEEE Int. Symp. Circuits and Systems, 2002, pp. V-169V-172.

[10] J.-T. Wu and K.-L. Chang, MOS charge pumps for low voltage oper-ation, IEEE J. Solid-State Circuits, vol. 33, no. 4, pp. 592597, Apr.1998.

[11] Y. Palaskas, Y. Tsividis, V. Prodanov, and V. Boccuzzi, A divide andconquer technique for implementing wide dynamic range continuous-time filters, IEEE J. Solid-State Circuits, vol. 39, no. 2, pp. 297307,Feb. 2004.

Brian Guthrie received the M.Eng. (Hons) degreein electronic engineering from Aberdeen University,Aberdeen, Scotland, U.K.

He joined the Wireless Group of Philips ResearchLaboratories, Redhill, U.K., in 1999, where he wasinvolved in defining the ZigBee standard and workedon early ZigBee prototypes. His work includesbuilt-in self-testing of single-chip Bluetooth trans-ceivers and, more recently, ultralow-power radiocircuit design and protocols. He is currently withLifeScan Scotland Ltd., Inverness, Scotland, U.K., a

subsidiary of Johnson & Johnson, who make blood glucose meters for diabetespatients.

GUTHRIE et al.: CMOS GYRATOR LOW-IF FILTER FOR A DUAL-MODE BLUETOOTH/ZIGBEE TRANSCEIVER 1879

John Hughes received the B.Sc. (Eng) degreefrom Bristol University, Bristol, U.K., in 1960 andthe Ph.D. degree from Southampton University,Southampton, U.K., in 1992.

He was a Professorial Research Fellow with Im-perial College, London, U.K., from 1996 to 2000.He is currently a consultant to the Wireless Groupat Philips Research Laboratories, Redhill, U.K., andis a Visiting Professor with Mahanakorn Universityof Technology, Bangkok, Thailand. He has workedat Philips Research Laboratories in the U.K. and on

secondment at Philips Research Laboratories, Eindhoven, The Netherlands, andSignetics Inc., Sunnyvale, CA, on a variety of topics, including television, mag-netic thin-film memories, semiconductor memories, gigabit logic for PCM tele-phone transmission, microprocessors, analog circuits for subscriber loop sys-tems, circuit design and CAD for mixed-signal ICs, switched-current circuitsfor filtering and data conversion, and analog circuit design for low-voltage,low-power radio receivers. He has published about 100 papers and a similarnumber of patents. He has coauthored six books and was coeditor of the IEEElectronic Circuits and Systems series book, Switched-Currents: An AnalogueTechnique for Digital Technology (London, U.K., IEE Press).

Dr. Hughes was the recipient of the 1991 IEE Institution Premium and the1992 Eurel Prize.

Tony Sayers graduated from Cambridge University,Cambridge, U.K., in 1982.

He has since been with Philips Research Labora-tories, Redhill, U.K. Projects he has worked on haveincluded mixed analog-digital circuits for DECT, aCartesian loop amplifier IC for TETRA, and RF cir-cuits for paging receivers including mixers and fre-quency synthesizers. Most recently, he has worked onRF CMOS radios for the ZigBee standard, concen-trating in particular on dual-mode Bluetooth/ZigBeesolutions.

Adrian Spencer received the B.Eng. (Hons.) degreein electronic engineering from Staffordshire Univer-sity, Staffordshire, U.K.

He was with Philips Research Laboratories, Red-hill, U.K., from 1995 to 2004, where he was involvedin early circuit design work in the novel Philips Sil-icon-On-Anything process, moving into the designof low-power architectures for paging receivers andtransmitters for GSM and 3G mobile telephony. Morerecent work has included single-chip Bluetooth trans-ceiver integration, built-in self-testing of transceivers,

and integration of GPS into GSM transceivers. He holds several patents in thearea of mobile radio. He is currently with TRL Technology, Tewkesbury, U.K., aradio systems manufacturer which specializes in defense and government appli-cations, working on high-performance ultrawideband frequency synthesizers.

tocA CMOS Gyrator Low-IF Filter for a Dual-Mode Bluetooth/ZigBee TrBrian Guthrie, John Hughes, Tony Sayers, and Adrian SpencerI. I NTRODUCTION

Fig.1. Dual-mode receiver architecture.II. F ILTER S YNTHESISA. Complex Filters

Fig.2. Complex filter basics.Fig.3. Current-mode integrators. (a) Real. (b) Complex.B. Complex Bandpass Synthesis

Fig.4. Channel filter architecture. (a) Fifth-order low-pass prIII. P RACTICAL F ILTER D ESIGN

TABLEI S INGLE -E NDED D ESIGN V ALUES FOR THE D UAL -M ODE C HFig.5. Practical channel filter.TABLEII C HANNEL F ILTER R EQUIREMENTSFig.6. Class-AB CMOS transconductors [ 6 ] . (a) Single-ended. A. Balanced Transconductor

Fig.7. Balanced gyrator loop of two transconductors with feedthFig.8. Intrinsic capacitances of a MOS transistor in saturationFig.9. Modified transconductor with capacitive feedthrough equaB. Post- and Pre-Amplifier

Fig.10. Amplifier circuit. (a) Post-amplifier. (b) Pre-amplifieC. Frequency Response Control

Fig.11. Tuning control circuit.Fig.12. Measurement setup.Fig.13. Measured amplitude response of filter. (a) Bluetooth. (Fig.14. Measured passband response (Bluetooth) with and withoutFig.15. Measured compression characteristic (Bluetooth).IV. M EASUREMENT R ESULTS

Fig.16. Measured noise response of filter (Bluetooth).Fig.17. Measured third-order intermodulation characteristic (BlTABLEIII M EASURED P ERFORMANCE OF A T YPICAL C HANNEL F ILTERV. C ONCLUSIONL. Harte, Introduction to Bluetooth: Technology, Market, OperatiJ. A. Gutierrez, E. Callaway, and R. Barnett, Low Rate Wireless A. S. Sedra, W. M. Snelgrove, and R. Allen, Complex analog bandpJ. B. Hughes, A. Spencer, A. Worapishet, and R. Sitdhikorn, 1 mWK. W. Moulding, J. R. Quartley, P. J. Rankin, R. S. Thomson, andB. Nauta and E. Seevinck, Linear CMOS transconductance element fP. Andreani and S. Mattisson, On the use of Nauta's transconductY. P. Tsividis, Operation and Modeling of the MOS Transistor . NB. Shi, W. Shan, and P. Andreani, A 57-dB image band rejection CJ.-T. Wu and K.-L. Chang, MOS charge pumps for low voltage operaY. Palaskas, Y. Tsividis, V. Prodanov, and V. Boccuzzi, A divide