(19) (11) EP 1 548 934 B1

(12) EUROPEAN PATENT SPECIFICATION

(45) Reference to the grant of patent:

03.09.2008  Patentblatt  2008/36

(21) Application number: 04030390.1

(22) Date of filing:  22.12.2004

(51) Int. Class:H03F 5/00    (2006.01)

H03F 1/32    (2006.01)H03F 3/26    (2006.01)

(54) Vollsymmetrischer Leistungsverstärker

Fully differential push-pull amplifier

Amplificateur push-pull totalement différentiel

(84) Named Parties:AT BE BG CH CY CZ DE DK EE ES FI FR GB GR HU IE IS IT LI LT LU MC NL PL PT RO SE SI SK TR

(30) Priority: 22.12.2003 DE 10360347

(43) Of publication of application:

29.06.2005  Patentblatt  2005/26

(73) Proprietor: Blöhbaum, Frank79112 Freiburg i.Br. (DE)

(72) Inventor: Blöhbaum, Frank

79112 Freiburg i.Br. (DE)

(56) References cited: :WO-A-00/11779US-A- 4 229 706US-B1- 6 242 977

FR-A- 2 547 470US-A- 4 531 100

   

   Note: Within nine months from the publication of the mention of the grant of the notice to the European Patent Office of opposition to the European patent granted an appeal. The appeal must be submitted in writing and justified. It shall not be inserted as if the opposition fee has been paid. (Art. 99 (1) European Patent Convention).

Description

[0001]  The main quality criteria for power amplifiers, particularly for applications in audio, are the lowest possible distortion, high bandwidth, the lowest possible internal resistance as low transient intermodulation distortion and high stability of the operating point of the power components.

[0002]  Power amplifier can be constructed with very different active components whose properties generally determine the structure of the amplifier: tubes of different construction, forAs triodes, tetrodes, pentodes, or semiconductor devices, suchAs bipolar transistors, field effect transistors, MOSFETs and IGBTs. Tubes have to be used as power amplifiers usually have a relatively high internal resistance and therefore must be adjusted with the

aid of specially constructed transformers, the transformers, the low impedance of the speaker. These transformers, however, limit the achievable bandwidth and are themselves major source of nonlinear distortion. Which therefore desirable direct coupling of the low-impedance load, as a rule of the loudspeaker requires the use of active components with the lowest possible internal resistance or as large as possible steepness. For this reason, as the art enforced the use of semiconductor devices such as bipolar transistors, MOSFETs, IGBTs, or in rarer cases.

[0003]  Particularly frequently be realized amplifier with higher power requirements push-pull amplifier as complementary, wherein complementary bipolar transistors (npn and pnp transistor) or complementary MOSFETs (n-channel and p-channel transistor) or complementary IGBTs (n-channel and p-channel type) with a ground potential related to bipolar or unipolar voltage source are used.

[0004]  The common feature of in Figures 1 and 2 is illustrated the principle of a typical semiconductor-tipped push-pull amplifier or a corresponding push-pull output stage is the use of a complementary transistor pair. For closed-circuit setting via the complementary transistor pair serves an adjustable bias voltage which is between the two base terminals of the complementary transistor pair. The other circuit components are then different from each other.

[0005]  In the basic circuit of Figure 1 are used in the power unit at least two series-connected voltage sources whose junction point is connected to ground, so that, based on the ground potential is a positive and a negative supply voltage. The drive signal is related to ground potential balanced signal. The load, suchExample of the loudspeakers, is located between the connection point of the complementary transistor pair and the ground terminal.

[0006]  In the basic circuit of Figure 2 in the power supply only a single voltage source is used with one-sided ground connection. The drive signal is a unipolar signal related to ground potential. If the load comprises a DC path, it must capacitively from the lying on different levels output terminals, which are formed by the junction of the complementary transistor pair and the ground terminal, are separated.

[0007]  At the latter feature is also no significant changes by the insertion of a very low resistance measurement between the load resistor and the speaker and the mass, as suggested in some publications for current measurement.

[0008]  Are complementary push-pull amplifiers with tubes, it is not, as control valves according to the flow of electrons between the cathode and anode, only a negative power, so that a lack of complementary electron tube amplifier element.

[0009]  For the realization of highly linear power amplifier, the use of tubes as a power amplifier would be very desirable because they can be used in this application because of its square tube current-voltage characteristic curve very low distortion. The ideal would be the combination of tubes as a voltage amplifier with high share semiconductor devices, such as bipolar transistors, MOSFETs or IGBTs, as output-side power amplifier. Of great advantage here would be a dc coupling between the gain stages, so that would otherwise be required to provide isolation for coupling or bypass capacitors or transformers have no detrimental effect on the frequency response and harmonic distortion.Developments to date have focused on the optimization of pure solid state amps on the

one hand and pure tube amplifiers on the other. In addition, are also occasionally known amplifier with hybrid assembly components, separated by pipes or Transistor optimized using pre-and power amplifier included, which are usually capacitively coupled to each other.

[0010]  When the complementary push-pull output stages of FIG 1 with the two opposite at ground potential related polarity voltage sources, the use of complementary active components in the power unit necessarily inherent to a non-symmetric structure. Thus, both branches of the push-pull amplifier are not working up to the same devices, and thus in principle only partially symmetric. The full electronic symmetry would require that up to the line type all the electronic properties of npn and pnp transistors are equal. This is only approximately attainable. The differences disturb the symmetry behavior of the gain in the two signal paths and others complicate the realization of further distortion reducing circuit structures.

[0011]  A problem in search of a possible symmetrical complementary transistor pair is the very limited selection of pnp bipolar transistors, p-channel MOSFETs and p-channel IGBTs for high power amplifiers. This issue is not resolved by the creation of a quasi-complementary output stage, but moved at the expense of symmetry only in the driver stage. The complementary pairs of transistors used in the real world are not real couples on closer analysis. Thus, differing complementary transistors of a complementary pair, for example, substantially in terms of the amount of current amplification factor in bipolar transistors or the steepness with MOSFETs and IGBTs. This is a major cause of non-linearities in the amplifier behavior.

[0012]  For example, in the same drain-source withstand voltage, the same maximum allowable drain current and the same maximum permissible power loss of the chip size of p-channel MOSFETs approximately three times larger than the chip area of the complementary n-channel MOSFETs. This inevitably leads to considerably different capacities, particularly the gate-source capacitance and the drain-source capacitance. Due to the different capacitive loads result in problems of control and no further circuit measures result in different slopes (= Slew-rate) for positive and negative edges at the amplifier output. If such a negative feedback amplifier, the frequency compensation is determined by the much larger capacity of the negative branch (p-channel MOSFETs and IGBTs). This may reduce the achievable bandwidth performance unacceptable degree or cause instability. In bipolar transistors, these Anpassprobleme (= matching problems) which are necessary for the complementary symmetry, similar.

[0013]  In summary, it should be noted that inherently to the prior art push-pull amplifier in complementary technology can not achieve full symmetry. Several methods are known which attempt to rectify the resulting non-linearities by means of a negative feedback and / or a forward error correction (= Forward Error Correction). The required amount of circuitry for high quality power amplifier is very high. Furthermore, the correct operation of the error compensation of the typical applications for audio control with transient pulses critical. For example, in the US Patentanmeldung US 5,892,398a power amplifier described by the output signal controlled operating voltages used in accordance with the bootstrap principle for the compensation module for which highly linear production is again difficult. The typical audio power amplifier case, the control complex, and capacitive loads such as speakers and crossovers of different designs, requires a high degree of stability in the negative feedback condition, the conventional complementary push-pull output stages due to the different high-frequency properties of the two signal paths is hardly available.

[0014]  Transformerless push-pull power amplifier, use the active components only one conductivity type and the strictly symmetrical with respect to the electrical properties can be realized have so far been particularly associated with tubes on the market. They are commonly referred to as OTL amplifier (Output Transformer-Less) and sold commercially under that name. Tube amplifiers that are built on this principle, have the following fundamental disadvantages: tubes have too little slope to the low impedance of the load resistor and the speaker directly control. Without further circuit measures will result from a small deliverable power and a very high total harmonic distortion.

[0015]  To increase the steepness, many tube systems are connected in parallel. This can be excessive in parallel tubes or Tube systems, the slope of the resulting power component to a value just sufficient to increase. The distortion factor is still high. There are, however significant problems with the stability of the operating points of the many tube systems. The thermal drift of the anode current at a given bias and the variation of the tube characteristics due to aging, such as eg declining Katodenemissionsfähigkeit lead, at best, to an increased distortion, in the worst case, the total failure of the device.

[0016]  The operating point of the power stage is from these tube circuits using a fixed grid bias such as in US-Patent 4,719,431or set each by a cathode resistor. The associated pre-amplifier and the voltage is capacitively coupled. Examples of this are shown in U.S. Patents US 4,719,431in Fig.4 and US 6,242,977 B1in Fig.2. A DC coupling is in accordance with the prior art only indirectly by means of a high-resistance voltage divider, which in turn are bridged capacitively must. Without this bypass capacitor of the AC voltage drop would be much too large. On the other hand a direct DC coupling the drift was already highly vulnerable operating point of the output tube or the parallel-connected output tubes would destabilize further, as would then affect the drift of the Vorröhre multiplied with the DC gain of the output tube (s) in addition.

[0017]  According to the state of the art in tube circuits inevitable use of coupling and impair / or bypass capacitors, the achievable linearity significantly. As H. Lemme in the "Electronics" issue 10/2003, pp. 90-94with the article "capacitors as troublemakers" increase, particularly capacitors demonstrate a high DC voltage stress on high quality amp distortion to the unacceptably high levels.

[0018]  In the US-Patent US 4,229,706is a dc-coupled push-pull amplifier in semiconductor technology, the output stages consist of two identical npn power transistors, which are fed by two "floating" voltage sources. The two bipolar transistors of the output stage are cross-coupled with respect to their load output. The pre-amplifiers for each signal path via a respective driver circuit having a differential input. To the two non-inverting inputs the signal to be amplified is applied as a differential voltage. The two inverting inputs are connected via a resistor network with the two load terminals and with a negative pole of a mass-based bias. The outputs of the two driver circuits each feeding directly the base of the associated output transistor.

[0019]  In US 4,229,706Although widely described arrangement is symmetrical and DC-coupled, but this is paid for by the following disadvantages:

[0020]  The kind of realized quiescent current setting works only with bipolar transistors, since the height of the base current (current flow of the base-emitter junction) on the driver and the resistor network is controlled.

[0021]  The quiescent current setting of the exclusive control of the base current works in cross-coupled amplifiers only using the same ideal of bipolar power transistors. In reality, the inevitable spread of the current amplification factor absolute and, in dependence on the temperature in significant differences between the quiescent currents of the two power transistors. For an unacceptable high false differential current caused by the load resistance of the speaker.

[0022]  Moreover, the exclusive use of semiconductor components in the circuit of the US-Patent 4,229,706therefore also a disadvantage, since then the desirable flexibility of the design is very limited. Thus, for example, extremely difficult to realize with a low-distortion power amplifier tubes, which can drive the high base currents of the bipolar power transistors which can be used exclusively. It would be desirable in this case the use of MOSFETs or IGBTs as power device, because these devices are voltage-controlled and therefore a combination with low distortion power amplifiers on tube base would be possible.

[0023]  In US 6,242,977 B1is in Fig.4 a push-pull amplifier is shown with only one conductivity type MOSFETs voltage amplifier without a preceding, wherein the two MOSFETs via two "pseudo" floating voltage sources are fed. The two spurious floating voltage sources are formed here by means of a capacitive decoupling from a single voltage source. This circuit type has practically realizable high quality power amplifier following disadvantages:

[0024]  The operating point of the MOSFETs by means of a fixed gate voltage via a voltage divider on the one hand (cf. FIG 4, node 276, resistors 308 and 310) and a control voltage (from the Regleinrichtung 268) on the other set. The temperature-dependent change of the drain current at constant gate-source voltage results in the worst case, a significant change in the absolute value of the drain current, and can even lead to thermal destruction of the MOSFET. The set DC gate voltage must always be the outcome of the potential upstream but in none of the US 6,242,977 B1embodiments shown by way of capacitor are electrically isolated. This is the desirable realization of a DC-coupled amplifier without Koppe Lund / or bypass capacitors is not possible.

[0025]  In 4 of US 6,242,977 B1shown generation of floating voltages from a single voltage source for power amplifiers problematic because there occur, may require highly volatile, high current small decoupling resistors 286, 288, 290 and 292 and then in succession extremely large capacitors 294 and 296. These capacitors omitted if the two power amplifiers in each case a real floating voltage source as in the mentioned prior art in FIG 1 ofUS 6,242,977 B1is provided.

[0026]  The object of the invention, the indication of an improved push-pull amplifier having a fully symmetrical structure with functional units of the same conductivity type, is as large as possible freedom for the use of tubes and / or semiconductors and / or monolithic integrated circuits permits a transformerless connection enables the load and on coupling waived and bypass capacitors and transformers.

[0027]  The problem is solved by the features of claim 1 The advantages of the invention are that the improvements proposed a power amplifier is realized with very high linearity, preferably as audio amplifier is of the highest quality category. The operating points are adjustable in a simple manner, wherein residual asymmetries of the functional units via

additional control devices are securely stabilized. The control devices can be formed so that they can compensate for changes due to drift and temperature. By the required flexibility can be used for the power amplifiers and amplifier blocks in each case the most appropriate components and combine them. Whether the power amplifier is basically driven by a balanced differential signal to the ground reference level or whether the difference signal is a DC level is superimposed on, or whether a unipolar drive signal is present, at most minor changes required in the two signal paths or in the control equipment.

[0028]  The invention and advantageous embodiments and developments are now described with reference to the accompanying drawings:

Figures 1 and 2 schematically show two well-known complementary amplifier,3 shows schematically a known push-pull amplifier with identical and DC-coupled functional units and with cross coupling in the power stage,4 shows schematically a first embodiment of the invention,5 shows schematically an embodiment of an offset voltage control,6 shows an embodiment with an extended offset voltage control,7 shows an embodiment with tubes in the power stage,8 shows an embodiment with a different bias current setting,9 shows an embodiment with modified input terminals,10 illustrates an embodiment with optocouplers for quiescent current setting and11 shows an embodiment with an asymmetrical signal input.

[0029]  The well-known complementary amplifier of Figure 1 contains in series circuit comprises an NPN transistor Q1 and a pnp transistor Q2 between two voltage terminals + Ub-Ub and by two series-connected voltage sources V2, V3 are supplied. The connection node of these voltage sources is connected to the ground potential GND and to a terminal of the load RL. The other terminal of the load RL is located at the connection node of the two transistors Q1, Q2. For the quiescent current setting of the two transistors Q1, Q2 provides an adjustable bias voltage V bias between the two bases of these transistors. By suitably adjusting the bias voltage Vbias is the bias current can be set between the two transistors so that they, forIn each instance, AB amplification operation work. The single ended input voltage is supplied via an input + in the base of the npn transistor. The amplifier is complementary with respect to the mass away from load terminal non-inverting. Capacitors for electrical isolation are not shown in Figure 1. They are in any case necessary if the input signal is a DC voltage is superimposed and the Eingangsspannungshub is not very large Q1 relative to the base-emitter voltage of transistor.

[0030]  The well-known complementary amplifier of Figure 2 requires in contrast to FIG 1, only one voltage source V4 for the two Q3 in series connected complementary transistors, Q4 from an NPN and a PNP transistor, which lie between a positive supply terminal + Ub and the ground terminal GND . The voltage source V4 is connected at one terminal connected to ground GND and the other fed with the supply terminal + Ub. The quiescent current adjustment for the two transistors is carried out as in Figure 1 with an adjustable bias voltage Vbias, which lies between the two base terminals. Since the common emitter terminal is attached to this complementary circuit also approximately not more at the ground potential GND, there is in the idle state, a voltage difference between the common emitter terminal and the ground GND, which corresponds approximately to half the supply voltage V4. Be the to be connected between these two nodes must load RL, if it has a

direct current path, therefore separated from this DC voltage through a coupling capacitor C2. In addition to this coupling capacitor C2, the signal input + are also through a capacitor separated in voltage from the signal source, since the signal input + has approximately likewise half the supply voltage V4 and thus generally very different from the rest potential of the signal source is different.

[0031]  3 shows the already mentioned in US-Patent US 4,229,706Audio-described push-pull amplifier having a symmetrical structure, which is in the two signal paths DC-coupled and whose two npn output stage transistors Q1, Q2 respectively from a floating voltage supply V1, V2 are fed. The isolation of the ground reference potential GND is achieved with the floating power supply usually uses its own power transformer and a bridge rectifier circuit. The load RL is connected between the two emitters, which also form the bases for the connected crosswise floating supply voltages V1, V2. The balanced input signal is supplied via the inputs in + and-in, wherein the positive input + in non-inverting the input of a is executed in bipolar operational amplifier U1. The negative input-in is connected to the noninverting input of a first operational amplifier U1 identical second operational amplifier U2. The output of the first and second operational amplifier is directly, ie without coupling capacitor, connected to the base of the first and second output transistors Q1, Q2 connected. The power supply of the two V3 is operational amplifier via two series-connected voltage sources, V4, whose point of attachment to the ground potential GND is connected and operational amplifier whose mass facing away from poles, the positive and negative supply voltage + V,-V for both U1, form U2.

[0032]  The quiescent current adjustment of the two output transistors Q1, Q2 via a respective resistor R4 and R5 between the emitter and the negative supply voltage-V. The associated base current, the output of the associated operational amplifier U1 and U2. Here now is a problem, because by the resistance R4 and R5 certain current must be identical to the operational amplifier U1 and U2 be supplied base current, since this current V3 only with the voltage sources, V4 is linked and no other current path is available. It is therefore not regulated by the emitter bias current, but the base current. The resulting emitter bias current, the substantially from the floating voltage supply V1 or V2 is fed and flows back through the load RL is, directly from the current amplification factor of the respective output transistor Q1 and Q2-dependent and sprinkled with what is actually undesirable. The current regulation, strictly speaking, the control of the base current is performed by tapping the respective emitter potential and return on a relatively high value resistor R2 and R3 to the inverting input of the first and second operational amplifier U1, U2. Thus both emitters that give rise to the termination points of the load RL, as have the same potential, these two circuit nodes also has a relatively high value resistor R6 is connected and thus is the difference between the two potentials as a kind of offset voltage at the inverting inputs of the operational amplifier U1, U2 returned. Thus in the control case, the relatively small quiescent current does not hinder the modulation of the output transistors Q1, Q2, the respective emitter current increased by a resistor-diode path R7, D1 and R8, D2, which represents the respective emitter niederohmigeren a current path to ground GND available when the diode is conducting. It is mentioned that by the shown in Figure 3 base current return to the operational amplifier U1, U2 only current-controlled amplifiers, bipolar transistors therefore be able to be controlled, but not voltage-controlled components.

[0033]  4 shows schematically the circuit diagram of a first embodiment of a symmetrical push-pull amplifier according to the invention. The two signal paths are symmetrical and each contain a Vorverstärkerblock N1B or N2B and an output amplifier and M1 M2. The

power amplifier M1, M2 are connected to a floating voltage supply V1, V2, wherein the load circuits through a load RL, which between the base points of the two amplifiers is M1, M2 are connected crosswise closed. The two base points of the amplifier is set at the same time, the connection points or terminals RL1, RL2 for the externally connected load RL and form the output signal.

[0034]  The power supply device of the amplifier blocks N1B, N2B contains at least two series-connected voltage sources V3, V4, whose junction point is connected to ground GND. The resulting supply voltages + Ub-Ub and feed the supply terminals of the amplifier blocks that exist in the embodiment of FIG 4 each from an input-side amplifier N1, N2 with high open-loop gain. The positive pole of the output stage of the amplifier N1, N2 is fed to the positive supply voltage + Ub or with a different positive and also referenced to ground voltage source. The negative pole of the output stage of the amplifier N1, N2 is connected via a potential difference producer 4, 5 RL1 associated with the load connection, RL2 connected. 5 About this potential difference generator 4, an impressed current flows. Due to the high open loop gain in conjunction with a high impedance non-inverting and inverting input high impedance, with negligible offset voltage and negligible offset current and with a low impedance output, the amplifier N1, N2 conduct such as a more or less the ideal operational amplifier.

[0035]  As a power amplifier M1, M2 are shown in FIG 4, two three-pole, whose respective input electrode to the output of the preceding amplifier N1, N2 is connected. The first is M1 power amplifier is of a floating first voltage source V1 and the second amplifier M2 fed from a floating second voltage source V2. The first is M1 amplifier via its first supply terminal connected to a pole V1 of the first voltage source and via its second supply terminal, which also serves as Endverstärkerausgang, RL1 to the first output and the load RL to the other pole of the first voltage source V1. The second amplifier M2 is via its first supply terminal with a pole of the second voltage source V2 and via its second supply terminal, which also serves as Endverstärkerausgang, to the second output RL2 and via the load RL to the other pole of the second voltage source V2,

[0036]  The required operating-point determining control voltage is in the corresponding potential difference generators 4, 5 by a determined by a Ruhestromeinstelleinrichtung Ix with an adjustable current source I1 predetermined current that generates the potential difference producer 4, 5 the desired potential offset. The negative to the supply terminal-Ub connected Ruhestromeinstelleinrichtung Ix contains an adjustable current source I1, the current means of the two to the output terminals RL1, RL2 connected resistors R2, R3 is split equally between the two potential difference generator 4, 5, so that the operating point for the two amplifier M1, M2 is symmetric.

[0037]  This adjustment can of course be done in different ways, for example, purely manually as in the embodiment of FIG 4 on an adjustable current source I1, an adjustable voltage source V bias (see Figure 8) or via a potentiometer, which controls the current source I1. Expedient is also a temperature-influenced control, via a temperature sensor that detects the temperature of at least one power component, the adjustable voltage or current source I1 and Vbias so affected that the quiescent current of the amplifier power components M1, M2 remains constant or has a predetermined temperature profile. Even more innovative a full automatic control is such that the closed-circuit current value of the power components of the power amplifier M1, M2 is measured and compared with a predetermined reference value, and any deviations from the desired value by automatically

adjusting the Ruhestromeinstelleinrichtung be controlled to Ix. Of course, combinations of these control measures are described under protection.

[0038]  The two amplifier N1 and N2 have an inverting and a noninverting input and owing to their high open loop gain of a respective operational amplifier. When the resultant output operational amplifier, unseen by the most appropriate load port RL1 or RL2 of which is regulated by a feedback network to the ground potential GND. The feedback network is determined by the series connected resistors R6 and R7 or R8 and R9 are formed, which lie between the load terminal RL1 and ground GND or the load connection RL2 and ground GND. The common junction of resistors R6, R7 and R8, R9 is connected to the inverting input of the amplifier N1 or N2. N1 is the non-inverting input of the first amplifier to the positive input signal and + in the non-inverting input of the second amplifier N2 is connected to the negative signal input-connected in the push-pull amplifier. By the feedback network R6, R7 and R8, R9, the output of the amplifier N1 or N2 regulated so that the potential difference generator 4, 5 and the voltage divider R6, R7 and R8, R9, the potential at the noninverting input of the amplifier N1 or N2 is identical to the potential at the positive and negative input signal to + or-in. This provision of the feedback network is independent of the size of the bias current setting for the performance of the amplifier components M1, M2, which is determined by the variable current source I1. The resulting level offset at the potential difference producer 4 or 5 is by the amplifier N1 or N2 corrected.

[0039]  The resistors R10 and R11 from the positive or negative signal input to ground GND have no significance for the feedback. They serve only to determine the GND-potential for the respective non-inverting inputs of N1 and N2, and the line matching, since the signal inputs are high impedance to the noninverting inputs of the amplifier N1 and N2 in the rule. Later, an embodiment is shown in which the signal inputs are connected to the inverting inputs of the amplifier N1 and N2. With the device connected to the feedback network inputs, these inputs are then much lower impedance, it can account for the line adjustment if necessary.

[0040]  In order to realize a fully symmetrical amplifier according to the invention for a balanced input signal, the corresponding resistors in the two signal branches of course, be equal. It is R6 = R7 = R8 and R9. The ratio of resistors R6/R7 and R8/R9 determines the overall gain of the balanced amplifier.

[0041]  The upper half amplifier, essentially of the assemblies consisting N1B and M1, increases the half-wave of the symmetrical input signal and the lower half amplifier, consisting of the modules N2B and M2, reinforces the other half-wave.

[0042]  The amplifier M1, M2 can be practically implemented in various ways: as individual power devices suchAs power MOSFETs, IGBTs or bipolar power transistors, but also advantageous from combinations of like or different amplifier components with or without local negative feedback.

[0043]  In the diagram of Figure 5 is the output stage of the amplifier blocks and N2B N1B example as a triode or XTR1 XTR2 executed. In the embodiment shown in FIG 5, the amplifier blocks N1B and N2B so that each of the combination of an input amplifier N1, N2 with high open-loop gain and the amplifier N1, N2 is connected tube XTR1, XTR2, the control grid to the corresponding amplifier output is connected. The anode of the tube XTR1, XTR2 is connected to the positive supply voltage + Ub or with a different positive

and also referenced to ground voltage source. The cathode of the tube XTR1, XTR2 is connected via a potential difference producer 4, 5, which contains the simplest case, a resistor R4, R5, and over which a load-independent current flows, with the associated load connection RL1, RL2 connected.

[0044]  For the embodiment 5 is shown as a triode tubes XTR1, XTR2 can of course also tetrodes, pentodes, and Hexodes Heptoden with the prior art circuit corresponding to the additional gratings are used.

[0045]  As a power amplifier M1, M2 are shown in FIG 5, two n-channel MOSFETs having their respective gate electrode connected to the cathode of the preceding tube XTR1, XTR2 is connected. The respective source electrode is connected to the associated load connection RL1, RL2 and the respective drain electrode to the positive potential of the floating voltage source V1, V2. The operating-point determining gate-source control voltage is in the corresponding potential difference generators 4, 5 by a determined by a Ruhestromeinstelleinrichtung Ix with an adjustable current source I1 predetermined current, which generates via the resistors R4, R5 the desired potential offset. The negative to the supply terminal-Ub connected Ruhestromeinstelleinrichtung Ix contains an adjustable current source I1, the current means of the two to the output terminals RL1, RL2 connected resistors R2, R3, R4 equally between the two resistors, R5 divided so that the operating point for the two MOSFETs M1, M2 is symmetric.

[0046]  To simplify the functional description was according to the embodiment of FIG 4 is initially assumed that the amplifier N1 and N2 have a behavior like an ideal operational amplifier. The embodiments from Figure 5 assume that the active components differ in the two signal branches from the ideal properties and also with respect to the symmetry condition more or less to each other are different. By means of additional control loops according to embodiments of the invention can eliminate such shortcomings but for the overall operation of the push-pull amplifier.

[0047]  In the diagram of Figure 5 shows schematically a control arrangement such as the "bias control" B1 shown that the added push-pull amplifier of Figure 4. Since the same circuit and functional units in all figures of the drawing provided with the same reference numerals are unnecessary, repetitive functional descriptions. The circuit block B1 contains two control circuits B2, B3, whose inputs are connected to the amplifier outputs RL1, RL2 and their outputs to the feedback networks of the two amplifier blocks N1B or N2B are linked.

[0048]  The input of the first control loop B2 is represented by two series-connected identical resistors R16, formed R17, to the outputs RL1, RL2 are connected and their common connection point avg via a resistor R15 to the inverting input of operational amplifier U3 is connected. On the non-inverting input of U3, a DC voltage V5 is applied. The output of amplifier U3 is by means of a capacitor C2 due to the non-inverting input. The power of the amplifier U3 is conveniently carried out via the existing supply potential + and Ub-Ub or another power supply. A at the output of U3 connected inverter N3 inverts the output signal and supplies it via a resistor R18 back to the common connection point of the resistors R8, R9 and thus inverting input of the amplifier N2. For further consideration, this circuit node, which is an important interface for control and in Fig 5 is shown as a line labeled "bias 1". A second feedback of the output signal of amplifier U3 is N1 via a resistor R19 to the common junction of resistors R6, R7 and thus inverting input of the amplifier.

For further consideration, this circuit node, which is also represented an important interface for control and in Figure 5 as a line labeled "Bias 2".

[0049]  The input of the second control loop is formed by a subtracter B3 D1, whose minuend input + to the load terminal RL2 and whose subtrahend - is connected to the load terminal RL1. The output of the subtractor D1 determines the potential of the node diff, and is connected via a resistor R14 to the inverting input of an amplifier U4 is connected, which is connected via a capacitor C1 connected to its amplifier output. This output is also connected via a series circuit comprising an inverter N4 and a resistor R12 connected to the circuit node bias 1 and via a resistor R13 connected to the other circuit node bias the second U4 at the noninverting input of the amplifier is connected to the ground potential GND.

[0050]  The inventive function of the control block bias scheme B1 is that avg means of the first control loop B2 the mean of the potentials detected at the outputs RL1, RL2 and compared with a predetermined reference voltage V5. About the integrator formed by amplifier U3, the deviation of the potential is increased from the nominal value avg V5. Means acts of inverter N3 and the resistors R18 and R19 this increased deviation from the nominal value of V5 in the same way, so the two directly coupled control interface bias 1 and bias 2 that the average of the potentials at the output ports RL1, RL2 to the predetermined target value V5 is regulated. Thus, the amount of the absolute potential of the outputs RL1, RL2 defined set.

[0051]  Means of the second control loop is B3 diff the time average of the signed difference detected the potentials at the outputs RL1, RL2, this value is compared with a predetermined target voltage and adjusted deviations by means of the amplifier U4 formed integrator through the two control interface BIAS 1 and BIAS 2 . Since the difference voltage may be negligibly small as a rule is to say in the range of 0 V, is the reference input (= non-inverting input) supplied to the amplifier U4 as a reference voltage, the ground potential GND. Because possible deviations are usually different direction on the two control interfaces bias 1, bias effect 2 must be inserted into the return port on the bias of an inverter N4.

[0052]  The time constant τ2 = R14 * C1 of the second control loop is B3 selected so that no interference of signals to be amplified alternating voltage (= AC signals) takes place in the intended useful frequency range of the balanced amplifier. For use as an audio amplifier, for example, greater than or equal τ2 1s useful.

[0053]  The time constant τ1 = R15 * C2 of the first control loop B2 is set so that also a value is greater than or equal achieved 1s in order to avoid the event of any deviations from the ideal symmetrical behavior to influence the to be amplified AC signals in the intended useful frequency range.

[0054]  If the amplifier N1 and N2 in the Vorverstärkerblöcken implemented so that in the idle state may be considerable, even drifting loaded, the potential difference between the non-inverting and the inverting input occurs this amplifier N1, N2, then by a suitable design of the bias control of the disruptive influence to minimize the operating point stability.

[0055]  Such a case arises, for example, if the amplifier N1, N2 are not executed in semiconductor technology, but in tube technology. It could, for example, the first stage of a tube amplifier can be realized as a cathode base level. In this case corresponds, for

example, the grating the non-inverting input of the amplifier N1 or N2 and the cathode to the inverting input of N1 or N2. By the cathode current results in a rise in potential across the resistor R7 and R9. Would like in the example of Figure 5, the reference voltage V5 = 0V amount, it would by then idle at 0V potential is governed by absolute and RL1 RL2 a potential difference across the feedback resistors R6 and R8 arise. This in turn leads to a proportional bias current distribution as a function of the resistors R2, R4, R6 and R3, R5, R8 with: 

[0056]  The operating points of tubes are subjected to thermal and aging-related changes. Thus changes over the temperature and the aging of the input tubes, which have substantially the voltage gain in the amplifiers N1 or N2 cause the current component I (R6) or I (R8), and thus after the first relationship M1 by the I (R4) certain potential of the gate-source path of the output amplifier. Accordingly, a negative does this by the current I (R5) certain potential on the gate-source path of the amplifier M2. To eliminate this disturbing influence, or at least minimize, the reference voltage V5 is not fixed, but as the embodiment of FIG 6, the control interface bias 1 and bias 2 on the series-connected equal resistors R24 and R25 linked, so that at the common resistance tap a new node voltage avgln for a third control loop B4 is available. This node voltage is avgln via a resistor out R26 to the noninverting input of the control amplifier U3, which is different than not in Figure 5 fed with a reference voltage V5, but of its reference voltage at this terminal itself generates, connected by a capacitor C3 is there. The average of the potentials at the control interface BIAS 1 and BIAS 2 is thus about the RC-element low-pass filtered R26 and C3 and then serves as a reference voltage U3 at the noninverting input of the control amplifier. The corner frequency of the RC element R26, C3 is preferably at least one decade lower than the lowest selected to be transmitted useful signal. Through this circuit shown in Figure 6 is the basic version or drift due to the potential difference between the inputs of the amplifier N1 and N2, on which is recycled via the negative feedback network a portion of the output voltage and the output node RL1 or RL2 regulated ideally to zero, or at least small enough so that the error component of the quiescent current control has no effect.

[0057]  As already mentioned, an essential advantage of the proposed by the invention push-pull amplifier principle consists in the largely free choice of the active components. For example, as power devices, all known types of devices used. In the embodiments of Figures 5 and 6 were M1 as a power amp, M2 showed exemplary n-channel MOSFETs. Without structural changes in these circuits can also use other power devices are used including those which are not self-conducting suchAs IGBTs and bipolar transistors. Be self-conducting components that require a negative bias, for example, used tubes and some JFETs, can be secured in the same way by a slight modification of the circuit structure of the invention works. One belonging to this block diagram shows a corresponding embodiment of FIG 7th

[0058]  7 shows an illustrative embodiment of the push-pull amplifier, power amplifier as the M1, M2 Xtr3 each tube, which Xtr4 that directly drive the load RL. The amplifier blocks N1B, N2B included in contrast to the preceding embodiments according to Figures 5 and 6 is not a combination amplifier, but consist solely of the amplifiers N1, N2. These are implemented so that they Xtr3 at rest, a negative DC output voltage of several volts to several tens of volts depending on the connected tubes Xtr4 have. The adjustable current source I1 is connected to this, a positive potential + Ub, so that its owned half current flows

through the resistors R2, R3, the level shifter 4, 5 and the internal output stage of N1, N2 to the negative supply voltage-Ub. This current flow caused across the resistors R4, R5 have the required negative grid bias for the tubes or Xtr3 Xtr4. It is thus again Xtr3 with only one control variable, the quiescent current of both tubes, Xtr4 in the floating solid mass without reference powered amplifiers M1, M2 and defines synchronous and symmetrical set. As previously described, the adjustment of 11 and thus Xtr3 the quiescent current of the tubes, Xtr4 manually in many ways, are temperature-controlled or carried out by means of an automatic bias current control loop. In the exemplary illustration of Figure 7 triodes were used. For Xtr3, Xtr4 can all known types of tubes suchTetrodes and pentodes as well be used.

[0059]  Does the load RL in a special needs case by means of a transformer on the equipped with tube output stage M1, M2 are coupled, this can be the same structure as in FIG done 7, except that then instead of the direct load coupling a transmitter in a known manner the internal resistance of the tube amplifier to the low-load stepped down. The transformer can be run cost-saving as an autotransformer, because due to the invention quiescent current setting has no DC bias. Appropriately, the autotransformer winding symmetrical, for example, in a two-chamber assembly, so that the winding resistances are equal for both halves of the winding. Due to the lack of DC bias is a possible realization of a toroidal transformer. The winding is divided in this case to integer equal parts, which are then wound in pairs bifilar. A deviation from the symmetry principle, for example, by simply winding succession of the respective windings, still allows a stable operation of the inventive push-pull amplifier, exhausted its possibilities, but not fully.

[0060]  A further embodiment of the inventive push-pull amplifier is shown in FIG 8th This circuit is almost identical to the circuit of Figure 6, except that the manipulated variable for the quiescent current adjustment is not carried out as in Figure 6 by the adjustable current source I1, but by an adjustable voltage source Vbias, which ultimately through the resistors R2, R3 but also shows a generates electricity by means of these resistors R2, R3 is split equally between and through the resistors R4, R5 produced the desired operating point determined potential offset for the respective gate-source junction of the two n-channel MOSFETs in the output amplifier M1, M2.

[0061]  According to the invention, the absolute value of the potentials at the load terminals RL1, RL2 defined and adjusted to a value approximately equal. Since the resistors R2 and R3 are equal, the currents flowing through them are I (R2) and I (R3) equal to each other. These currents flow almost entirely through the resistors R4 and R5 and the cathode to the anode of the triode or XTR1 XTR2 to the positive supply voltage + Ub. By the resulting potential offset to the resistors R4 and R5 is a defined positive potential difference between the gate-source path of the MOSFETs M1 and M2, and it flows logically, a defined quiescent current through the output stage transistors. The setting operation of the voltage source Vbias can again take place in many ways, manually or automatically by means of a temperature-controlled quiescent current control loop.

[0062]  According to the embodiment of FIG 9 is an inventive push-pull amplifier by the fact realized that the input-side amplifier N1, N2 in the Vorverstärkerblöcken N1B, N2B are each connected to its inverting amplifier input to the signal inputs-in connected or + in the push-pull amplifier. In the entrance area of the amplifier N1, N2 are made only minor circuit changes. The resistors R7 and R9 is the feedback networks are now not connected to the ground potential such as in the comparable FIG 5, but to the negative and positive signal input-in or + in the push-pull amplifier. The resistors R10 and R11 to the noninverting

inputs of the amplifier N1 or N2 are at the ends facing away from the amplifier to the ground potential GND. Further, the recycled lines usually interface bias 1 and bias 2 is no longer as in Fig.5 to the inverting input of the amplifier N1 or N2 recycled, but is non-inverting at their inputs. The recycle of the control loops B2 and B3 normally flows through the resistors R10 and generate R11 the necessary control voltages for the amplifier N1, N2.

[0063]  The embodiment of Figure 10 shows another variation of the bias current adjustment. The basic circuit of this push-pull amplifier is roughly equivalent to the circuit of FIG 5th The potential difference generator 4 and 5 contain, in addition to the resistors R4 and R5 are each an optoelectronic component or OK1 OK2 or are coupled with such devices. I1 via a common current source, the light emission is controlled respectively of an LED element, of the intensity of which the volume resistance of a light-sensitive photo-transistor dependent. This photo-transistor is parallel to the already known resistance R4 of Figure 5 or R5, which is also known across the resistor R2 and R3 is energized, the low end but in contrast to FIG 5 on the negative potential Ub is hard. The control of the potential offset in the resistance R4 and R5 and thus the quiescent current in the amplifier or M1 M2 (here in each case an n-channel MOSFET) now follows by the parallel connection of the controlled photo-transistor distance over which a more or less large proportion of the resistance R2 and R3 impressed current flows.

[0064]  In the embodiment of FIG 10, the potential difference producer 4 and 5 are thus formed by an assembly that the resistors R4 and R5 each with an opto-coupler or OK1 Trecke OK2 combined in himself. A synchronous setting the quiescent current of both branches of the floating voltage amplifier is supplied by the application of both optical coupler with the same control or regulatory power. The power of equality is the embodiment of FIG 10 in force the current looped the adjustable current source I1 from the negative supply terminal Ub by both optical coupler OK1, OK2 to the positive supply terminal + Ub is.

[0065]  The embodiment shown in Figure 11 finally shows an advantageous arrangement of the push-pull amplifier according to the invention, where, in principle rather than balanced input signals only ended input signals to be processed. For comparison, the circuit is shown in FIG 5th Thus, both branches of the same signal, ie, at each half cycle to work may be, the non-inverting input amplifier N1 and N2 of the amplifier is controlled at the inverting input. For this purpose, the non-inverting input of the amplifier to the signal input N1 + into and through resistor R10 to ground potential GND. The inverting input of the amplifier is N1 in Figure 4 at the voltage tap of the voltage divider R6, R7, but not the distal end of resistor R7 in Figure 4 is connected to ground, but to the inverting input of the amplifier N2 is connected in the second signal branch is. The inverting input of the amplifier N2 is missing from the 4 known resistor R9, R8 together with the resistance of the local voltage divider for the feedback from the output RL2. 11 shows this function assumes the aforementioned resistor R7 by both inverting inputs of both amplifiers N1 and N2 are coupled to each other. The non-inverting input of the amplifier N2 is in contrast to FIG 4, finally connected through resistor R11 to ground GND. Thus in both branches of the signal gain is equal to one another, the resistors must follow the following relationship:

[0066]  This provision equation results from the fact that the upper branch as a whole and non-inverting amplifier and the lower branch operates as an inverting amplifier. The above

resistances are therefore not strictly identical, especially for relatively small gains. For larger gains, the difference in the resistance ratio of pure R6 / R7 or R8 / R7 low.

Claims

1. Amplifier realised as a symmetrical push-pull version having a first amplifying device and a similar second amplifying device whereat the first amplifying device (N1B, M1) is connected in a first signal path between a first input (+in) and a first output (RL1) and a second amplifying device (N2B, M2) is connected in a second signal path between a second input (-in) and a second output (RL2), furthermore a signal which should be amplified could be connected to the first input and/or to the second input (+in, -in) and an external connectable load (RL) could be connected to the first and second output (RL1, RL2),

- the first input (+in) is coupled to the signal input of a first preamplifying block (N1B), which is feeded by a voltage supply device (V3, V4), and whose output is coupled to the signal input of a first final amplifier (M1) and the second input (-in) is coupled to the signal input of a second preamplifying block (N2B), which is feeded by a voltage supply device (V3, V4), and whose output is coupled to the signal input of a second final amplifier (M2);- the first final amplifier (M1) is supplied by a first floating voltage supply (V1) and the second final amplifier (M2) is supplied by a second floating voltage supply (V2);- the first final amplifier (M1) is connected via its first supply terminal to one terminal of the first voltage supply (V1) and via its second supply terminal, which is used as an output of the final amplifier too, to the first output (RL1) und via the load (RL) to the other terminal of the first voltage supply (V1), and- the second final amplifier (M2) is connected via its first supply terminal to one terminal of the second voltage supply (V2) and via its second supply terminal, which is used as an output of the final amplifier too, to the second output (RL2) und via the load (RL) to the other terminal of the second voltage supply (V2),

characterized in that- a first generator of potential difference (4) between the output of the first preamplifier block (N1 B) and the first output (RL1) produces, dependent on the quiescent current control device (Ix), a first difference voltage between the output of the first preamplifier block (N 1 B) and the first output (RL1) which determines the working point and the quiescent current of the first final amplifier (M1) and- a second generator of potential difference (5) between the output of the second preamplifier block (N2B) and the second output (RL2) produces, dependent on the quiescent current control device (Ix), a second difference voltage between the output of the second preamplifier block (N2B) and the second output (RL2) which sets the working point and the quiescent current of the second final amplifier (M2).

 2. Amplifier in accordance with claim 1, characterized in that the first and second preamplifier block (N1 B, N2B) comprise electron tubes and/or semiconductors and on the output side the voltage alignment to the first and second final amplifier (M1, M2) is done by

means of the first and second generator of potential differences (4, 5) respectively in conjunction with the quiescent current control device (Ix). 3. Amplifier in accordance with claim 2, characterized in that additional to the first and second preamplifier block (N1 B, N2B) a regulating device (B1) dependent on the offset voltages at the first and second output (RL1, RL2) applies correction signals (bias2, bias1) to the first and/or second preamplifier block (N1B, N2B) and therewith said regulating device (B1) eliminates or at least reduces offset voltages at the first and second output (RL1, RL2). 4. Amplifier in accordance with claim 3, characterized in that the regulating device (B1) comprises a first regulating entity (B2) to which as actual value a signal (avg), dependent on the direct current voltage of the working points of first and second output (RL1, RL2), and as desired value a predetermined direct current voltage level (V5) are applied. 5. Amplifier in accordance with claim 3 or 4, characterized in that the regulating device (B1) comprises a second regulating entity (B3) to which as actual value the voltage-difference (diff) between the first and second output (RL1, RL2) is applied whereat the zero value of the voltage-difference acts as predetermined value. 6. Amplifier in accordance with claim 5, characterized in that the regulating device (B1) comprises a third regulating entity (B4), so that the predetermined direct current voltage level is composed of the average potential value of the first and second input (+in, -in) by means of a filter device. 7. Amplifier in accordance with claim 5, characterized in that the predetermined direct current voltage level of the first regulating entity (B2) is composed, by means of a filter device (R26, C3), of the average potential value (avgin) of those inputs of the first and second preamplifier blocks (N1B, N2B) which are each connected via a feedback network with the first and second output (RL1, RL2) respectively. 8. Amplifier in accordance with at least one of the claims 1 to 7,characterized in that the quiescent current control device (Ix) sets the quiescent current of the first and second final amplifier (M1, M2) by means of a temperature dependent control value in which the temperature is especially provided in the area of the final amplifier. 9. Amplifier in accordance with at least one of the claims 1 to 7,characterized in that the quiescent current control device (Ix) comprises a regulator which measures the actual value of the quiescent current of at least one final amplifier (M1, M2) and readjusts the quiescent current to a desired value. 10. Amplifier in accordance with at least one of the claims 1 to 8,characterized in that local feedback and/or compensation methods in both signal chains are applied to enhance the linear transmission behavior.