C M Â « Â » m ; I

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OUTPUT M A K R t f t

© Copr. 1949-1998 Hewlett-Packard Co.

Modular i ty Means Maximum Effect iveness in Medium-Cost Universa l Counter A s ing le ma in f rame and a w ide cho ice o f op t iona l t imer , counter , and DVM modu les prov ide bet te r per fo rmance a t lower cos t , meet each user ' s needs p rec ise ly , and leave room for fu ture growth.

by James F . Horner and Bruce S . Corya

HOW DO YOU DESIGN AN INSTRUMENT for the broad middle range of electronic counter appli

cations? How do you meet each user's needs pre cisely, yet offer room for future growth? How do you decrease cost yet increase performance?

The answers to these and many similar questions, we believe, are found in the new Model 5328A Universal Counter, Fig. 1. This single instrument is the successor to six counters, Models 5326A/B/C1 and 5327A/B/C. It offers improved performance, lower cost, and sever al new features. Its modular design makes it possible for a user to choose only the features he needs, yet leaves the door open for expansion at a later date.

What I t Is The 5328A is basical ly a power supply and

counter mainframe that supports user-selected op tions. In its standard configuration it offers a com plete portfolio of universal-counter functions: • Frequency — 100-MHz direct count • Period — 100-ns resolution > Period Average — 10-MHz clock • Time Interval — 100-ns single-shot resolution • Time Interval Average • Totalize— 100 MHz • Ratio— 100 MHz/10 MHz • Check. The inputs have matched (±4 ns) 100-MHz ampli fiers with ac or dc coupling, ±2. 5V trigger-level range, three-position attenuators (xi, xio, xlOO), 1-MQ input impedance, slope controls, trigger lights that act like logic probes, and high-speed output markers.

Capabilities can be greatly expanded by selecting options. Two DVM options provide dc voltage mea surements. One is an economy version that offers millivolt sensitivity, 10-MQ single-ended inputs, 125V range, 0.5% accuracy, and the "read trigger

level" function. The other is a high-performance unit that has 10-//.V sensitivity, automatic or manual range control to 1000V, 10-MÃ1 floating inputs, switchable filter, 0.03% accuracy, read trigger level function, variable integration time, and high-speed acquisition (up to 300 readings per second with two-digit resolu tion). For applications requiring greater frequency range, a 5-to-512-MHz direct-count option is avail-

â € ¢ â € ¢ â € ¢ â € ¢ â € ¢ A ^ ^ M C o v e r : M o d e l 5 3 2 8 A U n i v e r s a l C o u n t e r a n d i t s o p t i o n a l i n p u t m o d u l e s a r e s u p e r i m p o s e d o n a b a c k g round photograph showing the counter 's inter ior and the b u s - o r i e n t e d a r c h i t e c t u r e t h a t m a k e s i t s h i g h d e g r e e of modular i ty possib le. Input

m o d u l e s a v a i l a b l e i n c l u d e t w o D V M s , t w o u n i ve rsa l t ime r / coun te r modu les , and a 512 -MHz f requency counter module .

In this Issue: M o d u l a r i t y M e a n s M a x i m u m E f fect iveness in Medium-Cost Universal C o u n t e r , b y J a m e s F . H o r n e r a n d B r u c e S . C o r y a p a g e 2 Us ing a Modu la r Un i ve r sa l Coun te r , b y A l f r e d L a n g g u t h a n d W i l l i a m D . J a c k s o n p a g e 9 Synthes ized S igna l Genera to r Opera t ion to 2 .6 GHz wi th Wideband Phase M o d u l a t i o n , b y J a m e s A . H a l l a n d Y o u n g D a e K i m p a g e 1 5 A p p l i c a t i o n s o f a P h a s e - M o d u l a t e d Signal Generator , by James A. Ha l l . page 21

Printed in U.S.A. C Hewle t t -Packard Company. 1975

© Copr. 1949-1998 Hewlett-Packard Co.

Option 010 Oven

Oscillator Option 011 HP-IB

Interface

5328A Standard Conf igurat ion

• I I • I I I I I

, - Ã ³ - 0 ,

\ i l l- > ' 3

"ò O p t i o n 0 2 0 O p t i o n 0 2 1

D V M D V M Option 030 Channel C

Option 040 Universal Module

F i g . 1 . M o d e l 5 3 2 8 A U n i v e r s a l C o u n t e r a n d o p t i o n s . S t a n d a r d con f igura t ion , cons is t ing o f ma in f rame and un ive rsa l modu le , p ro v i d e s a l l s t a n d a r d u n i v e r s a l c o u n t e r f u n c t i o n s , h a s t r i g g e r l i gh ts and ou tpu t markers . D i rec t c o u n t i n g r a n g e i s 1 0 0 M H z . O p t i o n a l u n i v e r s a l m o d u l e h a s o n e more d ig i t o f t ime in te rva l reso lu t i on and a j i t t e red c lock fo r p rob lem- f ree t ime in te rva l averag ing . Channe l C opt ion has a 512-MHz d i r e c t c o u n t i n g r a n g e . D i g i t a l vol tmeter opt ions— economy and h i g h - p e r f o r m a n c e â € ” c a n / e a c i channe l A and B t r igger leve ls as w e l l a s e x t e r n a l d c v o l t a g e s . O t h e r o p t i o n s a r e h i g h - s t a b i l i t y t ime base osc i l la to r and HP in te r face bus modu le .

able. This option offers 15-mV sensitivity across the band, 50Ã1 fuse-protected input, and a ninth digit added to the eight-digit mainframe. For applications requiring a precision time base, a high-performance oven-stabilized time base option is available.

For greater precision in the basic counter functions, there is a high-performance universal module. It has a 100-MHz clock for increased resolution in time interval and period measurements. Its functions are: « Frequency — 100-MHz direct count < Period — 10-ns resolution 3 Period Average — 100-MHz clock • Time Interval — 10-ns single-shot resolution • Time Interval Average — 100-MHz phase-jittered

clock • Totalize— 100 MHz • Ratio— 100 MHz/10 MHz • Check— 100 MHz

The inputs have matched (±2 ns) amplifiers, ac/dc coupling, ±2. 5V trigger level range, three-position attenuators (xi, X2, X20), 1-MO or 50Ü switchable input impedance, slope control, logic-probe trigger lights, channel-A high-speed marker, time-interval A-to-B high-speed marker, and a variable-delay fea ture.

For systems applications an HP interface bus (HP-IB) option is available.

Organization The new counter is organized into four main oper

ating regions (see Fig. 2): • The main counter area • The input options area • The power supply area • The HP-IB area.

Each region operates relatively independently and communicates to the others through an internal bus system (see article, page 9).

The power supply provides regulated dc voltage for the other operating areas of the instrument. Its ca pacity is sufficient to accommodate any combination of options. The main on-off switch of the instrument operates only the central power supply regulator; the main ac power line is never broken. Unregulated dc is constantly fed to the oven oscillator (if installed), eliminating the need for time-base warm-up. The fan gets its power from the ac power line by way of a triac, which is switched off by an optical isolator when the instrument is turned off.

© Copr. 1949-1998 Hewlett-Packard Co.

Main Counter Area The main counter area contains all of the function

al subunits of a standard counter with the exception of input signal conditioning and special logic, which are contained in the input options section. The de cade counting assembly contains eight decades of BCD counting logic, latches, and output multiplex ing logic. The time base assembly contains eight counting decades, output multiplexing logic, and synchronizers to generate precise timing signals for the main gate. The oscillator section contains the standard room-temperature 10-MHz oscillator cir cuit and the input/output logic to accept an external signal from the rear panel or an internal signal from the optional crystal oven oscillator.

The sample rate section controls the instrument display cycle. Inhibit, reset, main gate, transfer and sample rate signals are generated in this section, as is the BCD digit address code for the strobed display. Generation of decimal point and annunciators and BCD data decoding are accomplished by the display control circuits. Data out of the decade counting as sembly or the input option modules is decoded and displayed on the eight-digit LED display.

The function selector serves as the main signal switch of the instrument. It routes input signals through multiplexers to the decade counting assem bly and/or the time base. At the same time, it interacts with the display control circuits to determine the be ginning and end of the display cycle. The precision ECL main gate signal is created on the function selec tor through interaction with the time base assembly. The function selector also has extensive interaction with the input option modules. It is the main receiver of the high-speed data from the modules and the orig inator and receiver of module arming pulses.

The flexibility of the 5328A comes from the ability of all these operating subsections to accept diverse data from various combinations of input option modules. This is accomplished through the use of a 4K read-only memory (ROM) as the master con trol of the instrument. Located in the display con trol section of the instrument, the 4K ROM accepts the four-bit function code and the three-bit time base code from the front-panel switches or the HP-IB remote programming board, and generates 32 bits of output data which are transmitted through out the instrument to set up each subsection for the particular measurement situation. Various com binat ions of input opt ion modules are accom modated without circuit change as different ROMs are plugged into the instrument. This provides inherent obsolescence protection for the user. As each new input module is engineered, the main frame needs only a new ROM (supplied in an up date kit) to accept it.

Input Opt ions Area The input modules are the main interface between

the instrument and the outside electronic environ ment. They accept input signals and convert them in to the proper form to be handled by the main counter circuits.

In the universal module reside the main input am plifiers and Schmitt triggers. High-speed synchro nizers for complex timing measurements also reside in the universal module. One of the key performance options of the 5328A is the selection of one of the two universal modules. The prime difference between the two is the basic clock rate. In the standard unit the basic 10-MHz clock provides 100 ns as the funda mental timing unit. In the high-performance unit, a phase-lock multiplier extends this rate to 100 MHz and a basic timing unit of 10 ns. In the time interval averaging function, the multiplier unit, upon com mand from the ROM, phase modulates the 100-MHz clock with band-limited noise to prevent the syn chronous lockup problems associated with this mea surement.2 Replacement of the standard universal mod ule with the high-performance module replaces TTL technology with ECL technology without bur dening the instrument mainframe with unnecessary cost.

The middle section of the option module area pro vides the instrument with extended frequency capa bility. A SOU fuse-protected 512-MHz amplifier and Schmitt trigger feed the 512-MHz decade. Latches in this option strobe the ninth (least significant) digit from the module onto the data bus and into the dis play. In functions not requiring an input from this module, ROM lines deactivate the output strobing circuitry and the ninth digit on the display goes blank. The ninth LED digit is loaded into the main frame display board only when the channel C module is installed.

The third region of the option section contains the inputs for the optional digital voltmeters. Using a voltage-to-frequency conversion technique,1-3 these modules provide an output suited to the fre quency measuring capabilities of the mainframe. The low-cost unit provides the 5328A with inexpen sive access to the important capability of trigger lev el measurement as well as an excellent general- purpose single-ended voltmeter. In this unit and the high-performance unit , t r igger level measure ment is selected by means of switches located on the DVM panel. When the user selects either READ LEVEL A or READ LEVEL B, the DVM module discon nects itself from the external banana input jacks, attaches itself to the selected trigger level voltage, disengages the function and time base front-panel switches, places the code DVM on the function code bus and places 0.1-s gate time (1-mV sensitiv-

© Copr. 1949-1998 Hewlett-Packard Co.

Main Counter Area

R e s e t R O M 10 MHz

Oscil lator

Function S e l e c t o r M a i n G a t e

(TTL)

Fig. input the main operat ing areas are the main counter area, the input opt ions area, the power supp ly Future and the HP- IB area. These communicate v ia the master ins t rument bus. Future

op t ions can be accommodated eas i l y by a change o f read-on ly memory (ROM) .

© Copr. 1949-1998 Hewlett-Packard Co.

Mechanical Design of an Option-Configurable Counter

The mechan ica l des ign o f the 5328A Universa l Counter was a key fac to r in ach iev ing the des i red goa ls o f modu la r i t y , se r v iceabi l i ty , and low-cost producib i l i ty . The quest ion that had to be answered was, "How do we structure a package to a l low the customer to conf igure his own instrument eff ic ient ly from a wide range of opt ions?"

Adoption of the new HP System II instrument enclosure, with its structural front casting to support front panels, was the f irst step down the road to a modular universal counter with mult iple front- p a n e l h a r d T h e n e x t s t e p w a s t o d e s i g n o p t i o n a l h a r d ware pr in ted c i rcu i t cards to p lug d i rec t ly in to a common data bus on the motherboard, thus a l lowing any desi red opt ion mix. The space remain ing on the motherboard was dedicated to the counter c i rcui ts common to any set of opt ions. Placing these cir cui ts on the motherboard made space avai lable to p lug the HP- IB op t ion in to the motherboard and ou t the rea r pane l . P lace ment o f the power supp ly in the rear corner o f the package a l l o w e d f o r b o t h h e a t s i n k i n g a n d t h e m e c h a n i c a l i n t e g r i t y r e qu i red fo r many o f the heav ie r componen ts . Thus , the 5328A was d i v ided i n to fou r ma jo r sec t i ons †” coun te r ma in f rame, power supp ly , inpu t op t ions , and HP- IB in te r face card .

By tak ing advantage o f the mechan ica l s t rength inherent in the System I I corporate package and the r ig id i ty o f the mother board in the hor izon ta l p lane , the 5328A was des igned w i th a s ing le c ross -member b racke t ins tead o f a la rge chass is . Th is f e a t u r e a l l o w s a c c e s s t o t h e e n t i r e m o t h e r b o a r d f r o m b o t h sides (Fig. 1 ). The HP-IB option can be removed from the instru-

F ig . 1 . S i ng l e - c ross -member des i gn a l l ows access t o bo th s ides o f motherboard .

F ig . 2 . D iagnos t i c tes t ca rds use the d isp lay to ou tpu t tes t information, often pinpoint ing fai lures to the component level.

ment The tes t wh i le s t i l l e lec t r ica l l y a t tached by a cab le . The modular power supply can be complete ly tes ted before i t is in s ta l led . The common motherboard c i rcu i ts can be e lec t r i ca l l y iso la ted in to f i ve separa te c i rcu i ts . Ava i lab le in a k i t a re d iag n o s t i c t e s t c a r d s ( F i g . 2 ) t h a t r e p l a c e t h e f u n c t i o n s e l e c t o r c i r cu i t s and use the i ns t rumen t ' s d i sp lay to ou tpu t tes t i n fo r mat ion. This al lows rapid fai lure detect ion, of ten at the compon ent leve l . Thus the modular i ty o f the 5328A has been used to achieve the desi red level of serv iceabi l i ty .

Efforts to minimize factory cost In the 5328A were made in fab r i c a t e d p a r t s m a n u f a c t u r e a n d i n s t r u m e n t a s s e m b l y . T h e p lacement o f t he c i r cu i t s on the mothe rboard e l im ina ted the need fo r a t leas t f i ve p r in ted c i rcu i t boards and the i r connec t o r s . B y f a b r i c a t i n g t h e m o t h e r b o a r d a n d t h e p o w e r s u p p l y p r in ted c i rcu i t board on the same b lank and separa t ing them la te r , a sav ings o f near ly 50% was made. The modu la r f ron t - pane l concep t was a ided by bu i ld ing two ex t rus ions , one fo r the display and control sect ion and one for the opt ion modules. Ded ica ted too l i ng to b lank these ex t rus ions p rov ided s ign i f i cant ly super ior and lower-cost f ront panels than had previously been avai lab le because a subpanel and a dress panel were re p laced in each module by a s ingle ext rus ion panel . By loading c o m p o n e n t s o n t h e m o t h e r b o a r d a n d u s i n g p r i n t e d - c i r c u i t - m o u n t e d s w i t c h e s e x t e n s i v e l y , t h e t o t a l w i r i n g t i m e f o r t h e 5328A was reduced to less than 20% o f tha t requ i red fo r any 5326 o r 5327 un i t . Assemb ly t ime was fu r the r reduced by de c reas ing the fabr ica ted par ts count by the des ign o f mul t ip le - funct ion mechanica l par ts .

6

© Copr. 1949-1998 Hewlett-Packard Co.

ity) on the time base bus. Upon release of the READ LEVEL switch, the instrument returns to its pre vious state. Thus the user can check his trigger lev els without having to change and reset his function and time base settings.

The high-performance DVM option provides the user with the measurement capability of a manual or autoranging floating DVM that has a range of 10 piV to 1000V and a basic accuracy of 0.03%. For particu larly noisy environments, a switchable filter may be engaged to increase normal mode rejection to 50 dB at 50 or 60 Hz. Isolation for this option is accom plished through special high-speed transformers, op tical isolators, and an on-board switching dc-to-dc power supply. Although no remote programming of the front-panel controls is possible, remote con trolled voltage measurement is quite easy. Through the use of special range controls in the V-to-F con verter, a conversion factor of 10 kHz/volt is main tained regardless of the DVM's range. The volt meter may be placed in autorange and the user simp ly programs the DVM function from the HP-IB and any voltage from 10 i¿V to 1000V is measurable.

This technique results in a small problem. If, for ex ample, the user puts 900 volts on the input terminals, the output frequency is 900 x 10 kHz = 9 MHz. In a measurement time of one second, this would provide a resolution of 1 part in 9 x 106, far beyond the resolu tion limit of the V-to-F converter. To prevent the user from misinterpreting his results, the module blanks the meaningless data, thus providing the user with a display that contains only accurate data.

In the measurement of trigger levels, the high-per formance DVM performs much like the low-cost ver sion, with an important exception. Measurement of trigger levels normally requires the user to compen sate mentally for the attenuation factor used in the universal module input attenuator. For example, if one volt is the trigger level voltage, xi attenuation yields an effective trigger level of one volt, x 10 atten uation yields ten volts, and xlOO attenuation yields 100 volts. The high-performance DVM, in combina tion with the high-performance universal module, eliminates the need for mental multiplication, auto matically reading out the effective trigger level in the three possible ranges of ±2.5 volts, ±5 volts and ±50 volts.

HP-IB Area The fourth area of the instrument, the HP-IB board,

provides for control of the counter by the HP inter face bus. Plugging into the main instrument bus through a ribbon cable, the internally mounted HP-

IB board controls function, time base, cycle rate, arm ing, and virtually all other controls in the instrument with the exception of the DVM and universal module front-panel controls. A special module program ming system in the HP-IB board allows any future module to be programmed through the present HP-IB system. For a more detailed description of the capa bility of the HP-IB option, see the article beginning on page 9.

Acknowledgments The 5328A project was a team effort from begin

ning to end. The key members of the team were Karl Blankenship, Ian Brown, Mike Wilson, Al Langguth, and Bill Jackson. In addition to the design team,

James F . Horne t J im Homer , p ro jec t manager fo r

r the 5328A Universal Counter, is ^ 0 ^ t T ' 9 - a 9 r a c i u a t e o f S t a n f o r d U n i v e r -

^ L ' s i t y â € ” h e r e c e i v e d h i s B S E E T Â » * 5 Â » J J f d e g r e e i n 1 9 6 6 a n d h i s M S E E i n

^ ^ 1 9 6 8 . W i t h H P s i n c e 1 9 7 0 , h e ' s &£ cont r iDuted to the des ign o f (he

\ a I 5 3 2 7 U n i v e r s a l C o u n t e r s a n d served as pro ject leader for the 5306A Mu l t ime te r /Coun te r . Be fore jo in ing HP he served for two years as an off icer in the U.S. Army. Born in Paso Robles, Cal i f ornia, Jim now lives in Santa Clara wi th h is wi fe and two daughters .

The Homers are proud of the i r min i -ecosystem that inc ludes a garden and seven egg- lay ing ch ickens . J im en joys a l l spor ts and coaches a vol leybal l team made up of HP engineers. He's a lso a hang-g l ider p i lo t and a nov ice as t ronomer .

Bruce S. Corya B r u c e C o r y a w a s p r o d u c t d e s igner for the 5328A Universal Counter , and before that was pro duc t des igner fo r the 5381A/82A Counters , the 5300B Measur ing System, and severa l 5300-Senes snap-on modules. He joined HP in 1969, just af ter graduat ing f rom Purdue Univers i ty wi th a BSME degree. In 1971 he rece ived h is MSME deg ree f rom S tan fo rd Un i versity. Bruce was born in Greens- burg, Ind iana. He and h is wi fe , who live in Santa Clara, California, both work at HP's Santa Clara

Div is ion and are t ravel enthusiasts. Bruce enjoys spor ts , par t icu lar ly swimming and par t ic ipat ing in the Santa Clara Divis ion softbal l league.

© Copr. 1949-1998 Hewlett-Packard Co.

I n p u t C h a r a c t e r i s t i c s C H A N N E L A A N D B ( s t a n d a r d a n d o p t i o n 0 4 0 )

S E N S I T I V I T Y

2 5 m V r m s 0 â € ” 4 0 M H z I d c c o u p l e d )

2 0 H z â € ” 4 0 M H z { a c c o u p l e d )

2 0 0 k H i â € ” 4 0 M H i l a c c o u p l e d a n d S O U w i t h O p t 0 4 0 )

5 0 m v r m s 4 0 M H z - 1 0 0 M H z

M m p u l s e w i d t h 5 n s . 1 4 0 m V p - p

C O U P L I N G a c o r d c . s w i i c t i s e t e c t a b l *

I M P E D A N C E 1 M i l â € ¢ 4 0 p F ( s w i t c h s e l e c t a b l e 1 M i l o r 5 0 f t n o m i n a l w i t h

O p l 0 4 0 )

T R I G G E R L E V E L V a r i a b l e o v e r : 2 5 v o l t s l i m e s a t t e n u a t o r s e t t i n g w i t h

0 v o l t p r e s e t p o s i B o n

T R I G G E R S L O P E i n d e p e n d e n t s e l e c t o r o f - w  « o p e

A T T E N U A T O R S - ' â € ¢ 1 0 . - 1 0 0 ( â € ¢ 1 . â € ¢ 2 . - 2 0 w i t h O p t 0 4 0 )

D Y N A M I C R A N G E 2 5 m V t o 1 V r m s â € ¢ a t t e n u a t o r s e t t i n g t o r 0 â € ” 4 0 M H i

5 0 m v t o 5 0 0 m v r m s â € ¢ a t t e n u a t o r s e t t i n g l o r 4 0 â € ” 1 0 0 M H z M A X I M U M I N P U T I d C c o u p l e d )

X I 2 5 0 V r m s . d C - 5 0 k H z

1 2 5 â € ¢ 1 0 ' V r m f f r e q 5 0 K H i - 2 5 M H z

5 V r m s . 2 5 â € ” 1 0 0 M H z

X i O X 1 0 0 2 5 0 V r m s d C - 5 M H z

i 2 5 â € ¢ 1 0 * V r m a / t r a q . 5 - 1 0 0 M H z

X 2 X 2 0 2 5 0 V r m s . d c â € ” 5 0 0 k H z

( O p l 0 4 0 ) 1 2 5 â € ¢ 1 0 Â » V r m * / l r e q 0 5 â € ” 2 5 M H i

5 V r m s . 2 5 â € ” 1 0 0 M H z

A c c o u p l e d V m a x * 2 0 0 V ( p e a k - o c i f o r d c â € ” 2 0 H z . s a m e a s d c c o u

p l e d t o r f r e q u e n c y g r e a t e r m a n 2 0 H z

O p t 0 4 0 5 0 1 1 p o s i t i o n 5 V r m s . C f c - 1 0 0 M H i

C H A N N E L I N P U T : C o m m o n A o r s e p a r a t e s w i t c h s e l e c t a b l e I n C O M A p o s i t i o n

s e n s i t i v i t y r e m a i n s i n e s a m e I m p e d a n c e b e c o m e s < M l ) - 6 5 p f l o r t h e

s t a n d a r d a n d 5 0 0 K i t â € ¢ 6 5 p F l o r t h e O p t i o n 0 4 0 n i g h i m p e d a n c e p o s i t i o n

5 0 1 > p o s i t i o n r e m a i n s n o m i n a l 5 0 1 1 C H A N N E L C ( o p t i o n 0 3 0 )

S E N S I T I V I T Y 1 5 m V r m s 5 M H z â € ” 5 1 2 M H z

T R I G G E R L E V E L 0 V f i x e d

I M P E D A N C E 5 0 1 1 n o m i n a l M A X I M U M I N P U T 5 V r m  »

I N P U T P R O T E C T I O N F u s e d

C O U P L I N G d c

F r e q u e n c y M e a s u r e m e n t s

F R E Q U E N C Y A ( s t a n d a r d a n d o p b o n 0 4 0 )

R A N G E 0 â € ” 1 0 0 M H z o V e c t c o u n t R E S O L U T I O N i M H z t o 0 1 H z m d e c a d e s t e p *

A C C U R A C Y - _ i c o u n r â € ¢ t i m e b a s e e r r o r

D I S P L A Y K H z M H z

F R E Q U E N C Y C l o p b o n 0 3 0 )

R A N G E 5 â € ” 5 1 2 M H z O V e c t c o u n t

R E S O L U T I O N 1 M H z t o O i H z m d e c a d e s t e p s

A C C U R A C Y â € ¢ 1 c o u n t â € ¢ t i m à ¡ b a s e e r r o r

D I S P L A Y K H z M H z

P e r i o d M e a s u r e m e n t s P E R I O D A ( s t a n d a r d a n d o p t i o n 0 4 0 )

R A N G E 0 â € ” 1 0 M H z

R E S O L U T I O N t o o n s t o i s m d e c a d e s t e p s

( 1 0 n s l o O t s w i t h o p i 0 4 0 ) A C C U R A C Y r l c o u n t * l i m à ¡ b a s e e r r o r - t r i g g e r e r r o r '

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m a y b e m e a s u r e d b y p l a c i n g t h e c o u n t e r i n T i A â € ” e a n d t i e U n r v e r a e l M o d u l e m c h e c k ( C H K )

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M E A N I N G F U L F U N C T I O N S : F H E Q A P E R A P E R A V G A . T l

A _ B , T l A V G A â € ” B . R A T I O C A S T A R T A E V E N T S C A â € ” B

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p r o g r a m c o n t r o l ( " k s t e n e r )

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m a n u a l m o d e s A r m i n g D i s p l a y m o d e s M e a s u r e m e n t m o d e s O u t p u t m o d e s

H P - I B C O M M A N D S : r e s p o n d s l o t h e I r j l o w i n g b u s c o m m a n d s ( s e e H P - I B U s e r s

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E n a b l e S e à ± a l P o l D i s a b l e G o t o L o c a l S e l e c t e d D e v i c e C l e a r a n d G r o u p

E i e c u t r v e T n g g e r

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w h i c h h o l d s d i s p l a y i n d e f i n t e l y

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T 1 M E B A S E O U T P U T : H e a r p a n e l o u t p u t T T L l e v e l s C H E C K M H z W i t h f u n c t i o n s w i t c h , n C H E C K , c o u n t  » s h o u l d d i s p l a y 1 0 M H z

â € ¢ c o u n t W i t h o p t 0 4 0 . p l a c e f u n c t i o n s w i t c h m F r e q A a n d u n i v e r s a l m o d u l e

m C H E C K ( C H K ) â € ” c o u n t e r s h o u l d d i s p l a y 1 0 0 M H z â € ¢ 1 c o u n t

S T A N D A R D C R Y S T A L

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i M i M a x i m u m i n p u t 5 V p - p C o r r e c t r e a d i n g o b t a i n e d o n l y w i t h 1 0 M H z

i n p u t O t h e r i n p u t s g i v e s c a l e d r e a d i n g s F o r o p t 0 4 0 o n l y t h e f o l l o w i n g c o n

s t r a i n t s a p p l y o x t f r e q S t d m u s t b e 1 0 M H z f o r P e r i o d A v g . T l A v g . P e n o d

( N = 1 ) . a n d T l ( N - i |

T R I G G E R L I G H T S : L > g N i s O N w h e n i n p u t i s a b o v e t n g g e r l e v e l . O F F w h e n

i n p u t i s b e l o w t r i g g e r l e v e l B L I N K I N G w h e n c h a n n e l i s t r i g g e r i n g O p e r a t i v e

o v e r I r e q u e n c y r a n g e 0 â € ” 1 0 0 M H z

M A R K E R O U T P U T S : I n v e r t e d c h a n n e l A a n d c h a n n e l B S c h m t l t r i g g e r o u t p u t s

a v a i l a b l e o n f r o m p a n e l 0 t o - t O O m V l e v e l s i n t o S O f l â € ¢ 2 0 n s d e l a y W i t h

O p t 0 4 0 . c h a n n e l A S c h m i d t t r i g g e r a n d T l A â € ” B m a r k e r o u t p u t s ( 0 t o

- 5 0 m V ) a v a i l a b l e o n f r o n t p a n e l T l A â € ” B i s r  » g h d u n n g t h e  « m e i n t e r v a l m e a s u r e d b y t h e c o u n t e r O u t p u t s p r o t e c t e d f r o m i n a d v e r t e n t l y a p p l i e d v o l t a g e

t à ³ - 5 V d c

A R M : r e a r p a n e l s w i t c h k j m s a r m n g O N o r O F F W t h a r m i n g O N t n * m e a s u r e

m e n t i s T h e b y a n i n p u t O t h e r t h a n m e m p u t i n v o l v e d m m e m e a s u r e m e n t T h e

f o l l o w i n g a r e a r m e d b y o n e v e n t a t B F R E Q A . P E R I O D A P E R I O D A V G A

F R E O C D V M R A T I O C A t h e f c * o w i n g a r e a r m e d b y a n e v e n t a t C T l

A â € ” B T l A V G A â € ” 6 E V E N T S C A â € ” B R A T I O B A

O P E R A T I N G T E M P E R A T U R E : O * I O S O X

P O W E R R E Q U I R E M E N T S : 1 0 0 1 2 0 2 2 0 2 4 0 V r m s . 5 % . - 1 0 % ( s w i t c h

s e l e c t a b l e ) 4 8 â € ” 6 6 H z 1 5 0 V A m a  »

W E I G H T N e t 8 5 n g i i 8 C I 2 o z ) P R I C E S I N U S A

5 3 2 B A U n i v e r s a l C o u n t e r S i . 3 0 0

O p t 0 1 0 H i g h S t a b i l i t y T i m e B a s e $ 5 2 5

O p t O H H P - I B I n t e r l a c e $ 3 5 0

O p t 0 2 0 D V M . $ 2 0 0

O p t 0 2 1 H i g h P e r f o r m a n c e O V M Â « 5 0 0

O p t 0 3 0 C h a n n e l C . S 4 0 0

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O p t i o n a l A c c e s s o r i e s h a n d W s r a c k m o u n t b r a c k e t s M A N U F A C T U R I N G D I V I S I O N : S A N T A C L A R A D I V I S I O N

5 9 0 1 S t e v e n s C r e e k B o u l e v a r d

S a n t a C l a r a C a M o r m a 9 5 0 5 0

many others made significant contributions to the project. Dexter Hartke should be credited with first proposing the 5328A modular concept and Ken Jo- chim should be credited with bringing the 5328A through the investigation stage. Ian Band provided management support at critical stages in the project and along with Ken MacLeod was instrumental in product definition. Bill Kampe and Martin Neal pro vided invaluable marketing support. Credit for suc cessfully transferring the 5328A into production goes to Bob LaFollette, Rich Endo, Roy Criswell, Mike Freeman, Steve Balog, Jim Carlson, and Terry

Bales. Credit for service support and manuals goes to Richard Buchanan and the fastest pen in the West, Irv Simmons. S

References 1. K.J. Jochim and R. Schmidhauser, "Timer/Counter/ DVM: A Synergistic Prodigy?," Hewlett-Packard Journal, April 1970. 2. D.C. Chu, "Time Interval Averaging: Theory, Problems, and Solutions," Hewlett-Packard Journal. June 1974. 3. J.F. Horner, L.W. Masters, and P.T. Mingle, "DMM and DAC Modules Expand Low-Cost Measuring System," Hewlett-Packard Journal, June 1973.

© Copr. 1949-1998 Hewlett-Packard Co.

Using a Modular Universal Counter Here 's what the var ious features of the new Model 5328 A Universa l Counter can do for the user .

by Al f red Langguth and Wi l l iam D. Jackson

MODEL 5328A UNIVERSAL COUNTER provides a great deal of flexibililty and measurement

power in one instrument. New functions, new out puts , better performance at lower cost, system compat ibility, and modularity are among its contributions, it has been engineered to give the user a new confi dence in his ability to make complex measurements of time, voltage, and frequency.

Instrument-User Feedback Setting up the measurement has always been the

most difficult aspect of using a universal counter, especially adjusting trigger levels and input signal conditioning so the input amplifiers trigger at the proper points on the input waveform. Even someone very familiar with the typical instrument had to check carefully to be sure that the interval measured by the instrument was the one he hoped to measure. The 5328A makes measurement setup much more straightforward and reliable by increasing the instru ment-user feedback and making more information available. Three-state trigger lights, marker outputs, and digital readout of trigger levels all serve to in crease awareness of what measurement is actually be ing made.

The trigger lights on the 5328A have three stable states. They blink when the input amplifiers are trig gering, remain off when the input signal is below the adjusted trigger level setting, and remain on when the signal is above it. This is very similar to a logic probe function, with the important exception that the threshold voltage is adjustable over a range of ±2.5 volts times the attenuator setting.

This trigger light scheme allows the user to deter mine the status of the input amplifiers at a glance. If the LED is off, he must lower the trigger level to find the trigger point. If the LED is on, he must raise it. If changing the level causes the LED to change state

without blinking, then not enough signal is present, and either reducing the attenuator setting or increas ing the signal amplitude is required. If the LED will not change state, then the dc value of the signal is outside the trigger level range, and the user must either increase the attenuation or switch to ac coupling.

The polarity of low-repetition-rate pulses can be determined by observing the trigger lights. Negative pulses cause it to blink off, while positive pulses cause it to blink on.

Another useful technique is to attach a 1-Mfi 10:1 divider probe to the input. Setting the trigger level to + 0.14 volts makes the trigger light and probe a TTL logic probe, and a setting of -0.13 volts makes them an ECL probe. But there is an instrument behind this kind of probe. Unlike logic probes, these can be used to measure in-circuit propagation delays and rise times.

The A and B markers are digital outputs generated by internal Schmitt triggers. Whenever the input signal crosses the preselected threshold voltage, these outputs change state. They operate over the full 100-MHz frequency range of the input amplifiers and change state within twenty nanoseconds of a triggerable event. By displaying these markers and the input signal on an oscilloscope, the trigger points typically can be monitored to within a few nanoseconds and a few millivolts of their actual values. Problems involving false triggering now be come simple to solve, as shown in Fig. 1.

The high-performance universal module replaces the B marker with one labeled TI A— B OUTPUT. This is the measured time interval. It can also be superim posed upon the input signal on an oscilloscope.

Another advantage of these markers is their useful ness as a trigger source. Since the input amplifiers have better than 25-mV sensitivity and generate fast- rise-time outputs, the marker outputs can be used to

9

© Copr. 1949-1998 Hewlett-Packard Co.

Trigger Level

T r i g g e r , T ime

False Triggering

F ig . 1 . Marke r ou tpu ts show when the inpu t amp l i f i e rs a re t r i g g e r i n g . T r i g g e r l e v e l s c a n b e m o n i t o r e d w i t h a n o s c i l loscope by d isp lay ing the input s igna l and the marker . Mark e rs can a lso show fa l se t r i gger ing , and can be used to syn chronize other instruments.

synchronize oscilloscopes or other equipment when the input signal itself is not suitable.

Now suppose that an integrating DVM is installed in the 5328A. Besides measuring externally applied voltages, these modules can also read and display the internal trigger levels for the A and B input chan nels. This direct readout ability makes trigger level selection and adjustment very easy.

READ LEVEL in combination with the three-state trigger light can also be used to measure the peak-to- peak voltage of signals within the common mode range of the input. This is done by placing the counter in READ LEVEL, measuring both points where the trig ger light stops blinking, and taking the difference between these two points.

Bus St ructure Adds Power The modular structure of the 5328A would not be

so useful without a flexible and complete means for the individual modules to communicate with the in strument mainframe. For this purpose, a parallel two- way information bus was established between all modules and the mainframe (see Fig. 2, page 5). Capable of providing everything from power supplies to impor tant gating, display, and status signals, this ninety- line bus allows complete two-way communication not only between module and mainframe, but among modules as well. Any module, or the HP-IB interface, can use the bus to take independent control of dis play, function, or time base.

This convenient means of information exchange and control, combined with a mainframe capable of accepting signals from several modules simul taneously, makes the 5328A capable of making new measurements that were omitted from previous instruments because the cost of the extra circuits to implement them would not have been justified. This extra performance includes new functions, complete arming capability, and a very powerful yet simple to use HP-IB interface.

New Funct ions Two new ratio measurements, C/A and B/A, are

available. They offer the full bandwidth, sensitivity, and frequency capabilities of the A, B, and C input amplifiers.

It is also possible to totalize the channel C input during the synchronized time interval defined by TI A — B. This same time interval can be used as the inte gration time for the DVM, a feature that is useful for

Three-State Trigger Lights

The three-state trigger l ights use an ECL-to-TTL translator in a simple sig interesting feedback configuration. When the input sig na l is h igher than the t r igger vo l tage. À is an ECL low leve l ; when the input is be low the t r igger vo l tage, A is an ECL h igh. For quiescent s ignals, V, is at VBB, the ECL threshold vol tage^ and Vout is a TTL s ignal of polar i ty opposi te that of A. When A changes s ta te , caus ing Vou l to change, C, fo rces V, to fo l low i t and then decay exponent ia l ly toward VBB wi th t ime constant T = t /RjC- i . For example, le t À go f rom h igh to low. Voul then goes to a TTL high, turning on the tr igger l ight and forcing V, to some vol tage between VBB and Vout. Unt i l V-, decays to below the ECL high level, À wi l l have no further effect and the tr igger l ight wi l l remain on. I f À stays low, V, wi l l reach VBB, but wi l l stil l be higher than À and the LED will stay on. If À has returned high, a when V, crosses the ECL high level, Vout wi l l switch to a TTL low, turning of f the LED and forc ing V, downwards. Simi lar but opposite waveforms occur for À changing from low to high. Hence, a pulse of e i ther polar i ty as short as two or three nano s e c o n d s i s i n v e r t e d a n d s t r e t c h e d t o g r e a t e r t h a n 2 0 m i l l i seconds , l ong enough fo r the human eye to see . Fo r s igna ls whose f requency i s h ighe r than the b l i nk ing ra te , t he c i r cu i t reaches a s teady-s ta te cond i t i on where the pu lse du ty cyc le is 50% and the LED bl inks at a v is ib le rate.

Input E C L T T L

Voot

L E D O n O f f O n O f f O n O f f O n O H

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© Copr. 1949-1998 Hewlett-Packard Co.

applying a scale factor to voltage measurements. Of course, all of the standard universal counter

functions are offered, including Frequency A, Period A , Period Average A, TI A— B , TI Average A— B , Total ize A, Totalize Clock, Check, Frequency C, and DVM.

Arming In the armed mode, an input channel that is nor

mally not used in a particular measurement can be used as a trigger source to give independent control of the time a measurement is to begin. This mode is activated by setting the rear-panel NORMAL/ARM switch to the ARM position.

As can be seen in Fig. 2, only two measurements are armed in the normal operating mode: Frequency A and Frequency C. All other measurements are free running ("armed by the sample rate"). In the armed mode, the measurement is generally armed by an am plifier not used in the measurement. Channel B was chosen whenever possible.

Another means of arming the counter is the LINH (Low Inhibit) input on the rear panel. Grounding this input will inhibit arming. Fur arming any mpasure- ment that is free running, the LINH input need be opened only momentarily (about 1 /AS) for arming to occur. For measurements armed from amplifier chan nels A, B, or C, LINH must be high when that channel is triggered for arming to occur.

F ig . 2 . Norma l l y measurements a re i n i t i a ted when the f i r s t des i red s igna l en te rs the counte r , o r by the sample ra te c i r c u i t s . I n t h e a r m e d m o d e , t h e m e a s u r e m e n t b e g i n s o n l y a f ter an event occurs in the arming channel .

Ramp Generator

Function: F R E Q U E N C Y A

Gate T ime <- ' - Per iod of Ramp

Armed Mode

Frequency Read Out

S Â « m . , , - H 3 V

d o 3

= ~ -

Input Voltage

Gate Time

\ Ã̄

í f m i n , - 3 V

Channel B Trigger Level

Time

Fig. 3 . In the armed mode the counter can be used to check the ou tpu t f requency o f a vo l tage-con t ro l led osc i l l a to r as a funct ion of i ts input vol tage.

The counter can also be armed, and the NORMAL/ ARM switch programmed, via the HP-IB interface.

Many interesting applications can be devised for the arming system. One is shown in Fig. 3. Here a ramp voltage is applied to a voltage-controlled os cillator and to channel B of the 5328A. The VCO fre quency output is applied to channel A. The counter is then put into FREQ A, ARM, with a gate time that is a small fraction of the ramp period. The trigger level voltage of Channel B can then be varied, and the counter will display the VCO output frequency as a function of the input voltage given by trigger level B.

For doing armed TI measurements without a C channel, the LINH input is available. Placing the ARM/ NORMAL switch in NORMAL, and pulsing the LINH in put high will cause the counter to arm. With channel C, an arming signal may be applied to channel C with the switch in the ARM position.

An example of a more complex armed measure ment is shown in Fig. 4. Here a signal composed of bursts of two different frequencies is connected to channel A. An arming signal is put into channel B to trigger the counter whenever a burst of the desired frequency comes along, thereby causing the counter to measure only that frequency. The LINH input is also used to control the triggering, so that only selected arming signals in channel B will be effective. This can be useful for measuring frequency-shift-keyed signals.

Universal Modules T h e s t a n d a r d u n i v e r s a l m o d u l e i s c a p a b l e o f

11

© Copr. 1949-1998 Hewlett-Packard Co.

B Arm

RF Input

RF Signal: f , . f2

B Arm: Detected R F to B Channel (Negat ive Slope Used)

Low Inhibit (LINH)

Main Gate

F i g . 4 . T h e c o u n t e r c a n b e a r m e d b y t w o i n p u t s s i m u l t a n e o u s l y t o m a k e m o r e c o m p l e x m e a s u r e m e n t s . H e r e t h e a r m i n g c h a n n e l i s c h a n n e l B , b u t t h e L I N H i n p u t m u s t b e h igh when channe l B i s t r iggered fo r a rming to occur .

100-ns resolution for single-shot time intervals while the high-performance version multiplies the 10-MHz mainframe oscillator up to 100 MHz and can therefore resolve 10 ns in single time intervals. In TI average, the high-performance unit introduces a ran dom phase jitter into its clock to prevent lockup on in tervals occurring at rates synchronous with harmon ics of the instrument clock.1 Since resolution in this mode increases as VÑà where N is the number of time intervals averaged,1 the ten-nanosecond module will converge to a given resolution 100 times faster than the standard 100-ns module.

The input amplifiers of both universal modules have greater than 1 50-MHz bandwidth and 25-mV ba sic sensitivity over a ±2.5-volt common mode range. Because ECL logic levels are nominally from -800 mV to -1600 mV, the sensitivity and trigger level range in x l attenuation are adequate to measure prop agation delays and rise times of these devices. Cau tion is advisable when making such measurements, however. Even though both modules are fully capable of making repeatable average measurements with resolution much less than a nanosecond, the basic ab solute accuracy is only specified to be within ±4 ns for the standard module and ±2 ns for the high-per formance version. This means that each instrument must be individually calibrated against some stan dard and its absolute measurement error subtracted from the displayed time interval if absolute accuracy better than the instrument specification is required. The specified accuracy is adequate for measuring

such logic families as TTL and MOS directly. The x 2 attenuator setting of the high-performance module creates a ±5-volt trigger level range with 50-mV sen sitivity and optimizes the amplifiers to measure 5-volt logic families.

In the READ LEVEL function, the high-performance module multiplies the actual trigger level by 1 or 2, depending on the attenuator setting, before it goes to the DVM. Thus the DVM reads the actual trigger point including the effects of input attenuation. A 50ÃÃ/1-MQ switch is provided in the high-perfor mance module to optimize impedance matching at the amplifier inputs: 50Ã1 for fast signals in a 50ÃÃ en vironment, 1 Mil for reduced circuit loading or use with oscilloscope probes. Hysteresis compensation for time interval measurements allows more accurate determination of trigger points.2 This causes the amplifier to trigger typically within 5 mV of the se lected trigger level.

Delay Measurements with delay are also possible with

the high-performance universal module. This fea ture causes the counter to ignore unwanted signals that would normally trigger the input amplifiers and make a particular measurement impossible. When a non-zero delay is selected, once channel A triggers, it is prevented from triggering again until after the selected delay. Channel B is continuously disabled until after the delay. The delay is adjustable from 20 /us to 20 ms by means of a front-panel control. It can be measured by placing the counter in TI A— B and the universal module in check.

Input

Delay Time

Counted Normally

Counted With Delay

Input

Delay Time

Counted Normally

Counted With Delay

uiiiM 1 ui/uyw 1 UMW I I m V F I J m V F I / m V F I

F ig . 5 . H igh -pe r fo rmance un i ve rsa l modu le has a va r i ab le de lay fea ture . In t ime in te rva l measurements , de lay causes the counte r to ignore s top even ts un t i l the de lay t imes ou t . Delay can also be used in frequency, period, ratio, and total ize measurements to inh ib i t t r igger ing on unwanted t r iggerab le events.

12

© Copr. 1949-1998 Hewlett-Packard Co.

HP-IB Option Fits Simple or Sophisticated Systems Applications

Mounted inside the 5328A mainframe, the optional HP-IB mod ule makes the counter compatible with the HP Interface Bus.1 2 The counter 's d isp lay read ing can be read out onto the HP- IB and most of the f ront-panel and rear-panel mainf rame contro ls can be programmed by commands on the HP- IB.

Programmable controls include al l posit ions of the FUNCTION and TIME BASE switches, the sample rate, and the NORMALARM swi tch. The counter can be commanded to make a s ing le mea surement or self-tr igger continuously.

P r o g r a m s c o n s i s t o f s t r i n g s o f a l p h a n u m e r i c c h a r a c t e r s coded accord ing to the Amer i can S tandard Code fo r In fo rma t ion In terchange (ASCI I ) . The counters reper to i re inc ludes 43 commands , each cons i s t i ng o f one o r two cha rac te rs . The P command se ts up s tandard i n i t i a l cond i t i ons and mus t come f i rst in the ASCII str ing. R, reset, or T, reset and tr igger, should come last . Other commands can be in any order .

Data output is very f lexible. The counter can be programmed to wai t unt i l i ts current measurement has been read out before sel f - t r igger ing for another measurement, or to sel f - t r igger wi th out wa i t ing . Normal ly , data output occurs a t the end o f a mea surement , but the counter can be programmed to read out the con ten ts o f i t s d isp lay any t ime i t i s addressed , regard less o f whether a measurement ¡s in progrés5; (readout on the f ly) . A n o t h e r c o m m a n d d i s a b l e s d i s p l a y s t o r a g e ; c o m b i n e d w i t h readout on the f ly, this makes i t possible to read out the current i n fo rmat ion in the decade coun t ing assemb ly a t any t ime , re gardless of the state of the measurement.

Sti l l another command disables the reset signal to the decade count ing assembly . Th is causes the resul ts o f a ser ies of mea surements to be added together, a useful feature for averaging.

1 0 C M D " ? U 9 " , " P F 4 G 5 T " , " ? Y 5 "

2 0 I N P U T ( 1 3 , * ) D

3 0 E N D

Fig. 1 . A 9830 A Calcu la tor program to measure a f requency in channe l A o f the 5328 /4 Counte r . Coun te r and ca lcu la to r are in terconnected by the HP In ter face Bus.

Simple or Sophisticated In most appl icat ions the counter wi l l be to ld s imply to take a

r ead ing and ou tpu t t he r esu l t s . Ve ry l i t t l e know ledge o f t he counter 's reper to i re i s requ i red to wr i te th is k ind o f p rogram. Fig. 1 shows an example of a 9830A Calculator program to mea su re a f requency in channe l A . In the command sequence , P sets arm standard ini t ial condit ions (maximum sample rate, arm i n g o f f , s t o r a g e o n , a n d s o o n ) . F 4 s e t s t h e f u n c t i o n t o F r e quency A. G5 sets the gate t ime mult ip l ier at 10s (10-Hz resolu t ion) , and T in i t ia tes a s ingle reading. Not ice that the user can s imp ly ignore p rogramming fea tu res such as s to rage o f f , d i s able decade reset, arming on, readout on the f ly, and so on, be cause the P command turns them al l of f .

The 5328A lends i t se l f to ra ther soph is t i ca ted app l i ca t ions

HP-IB 5328A Counter 9820A 30A

Calculator

Channel A Input (Arm on A)

uu • Counter

Gate Open

Reset OCA N = 1

Output 1 = Counter

Display Reading

Calculator Display

Fig . 2 . Measurement averag ing us ing a 5328A Counter and a 9830A or 9820A Calcu la tor .

when necessary . As an example, cons ider the prob lem in mea surement averaging shown in Fig. 2. A ser ies of measurements may be taken w i th t he decade coun t i ng assemb ly rese t d i sa bled, causing al l of the measurement values to be accumulated in the decade count ing assembly. When the desi red number of readings have been taken, the resul t can be read out and div id ed by the number o f read ings to obta in the average. A l though each ind i v idua l measuremen t cou ld be read ou t and accumu lated external ly, the method shown here, using internal accumu lat ion, can be nearly an order of magnitude faster. References 1 D.W RICCI and G E Nelson. "Standard ins t rument in ter face s impl i f ies system de s ign. Electronics. November 14, 1974. 2. Hewlett-Packard Journal , January 1975

Many measurements that would not otherwise be possible can be made using this feature. Time inter vals from a start event to one of many stop events can

be made by adjusting the delay to ignore intervening stop events until the correct one occurs (see Fig. 5). Frequency measurements of signals containing

13

© Copr. 1949-1998 Hewlett-Packard Co.

noise that would normally cause extra trigger events can be made by setting the delay to disable the input during that part of the waveform where the noise nor mally occurs. Contact bounce in electromechanical re lays and other unwanted inputs can thus be pre vented from introducing error into a measurement.

Frequency C Module 512-MHz direct counting capability is provided

by the channel C module even though the mainframe circuitry can only count 100 MHz. In the Frequency C function, the gate on the mainframe counting as sembly is held open and a second gate in the C mod ule is enabled by the main gate bus line. A high speed decade in the module is inserted into the de cade counting chain without prescaling. After a mea surement is completed, circuits in the module use display information from the bus to strobe the least significant digit onto the display along with the mainframe display (see Fig. 2, page 5).

DVM The integrating DVM options interface to the main

frame by converting the input voltage to a frequency at 10 kHz/volt. In the READ LEVEL function, these mod ules also use the bus lines to disable the mainframe function and time base and force the mainframe to make a DVM measurement.

In the armed mode, DVM measurements are ini tiated only when channel B triggers. This ability to control a measurement externally is especially use ful for measuring dc levels that vary with time, such as in digital-to-analog converters.

The high-performance DVM combined with the HP-IB interface makes remote controlled voltage measurements extremely simple. It has a floating in put with internal optical isolators to translate its out put to chassis ground so it outputs directly onto the module bus like any other module. The front panel for the DVM is not programmable, but an autoranging scheme has been designed with systems applications in mind. It automatically selects the proper attenua tor range for any input signal, yet always maintains the same output format. As higher attenuation levels are chosen, the least significant digits that become invalid are blanked from the display. Hence, any voltage from 10/¿V to 1000V is measurable. READ LEVEL is not programmable.

HP- IB Module The HP-IB module makes the 5328A compatible with the Hewlett-Packard Interface Bus (see box, page 13). The counter can serve both as a talker and a lis tener. The simplest configuration consists of the 5328A connected to an HP-IB printer, such as the HP 5150A. Only one cable is required. All measure ments made by the counter are printed for a perma

nent record. The full power of the HP-IB interface is realized by

including the counter in a system that has a con troller, such as an HP 9830A Calculator. In such a sys tem, the counter can operate in an interactive mode as an universal counter, timer, and DVM. Acknowledgments

The 5328A would never have existed without the creative ideas and hard work of the rest of the design team: Mike Wilson, Ian Brown, and Karl Blankenship. Thanks to Dee Norman who put the 2000 holes in our motherboard, and to Ken MacLeod, Gary Sasaki, Ken Jochim, Bill Kampe, Ian Band and many others for their inputs. Finally, a special word of appreciation to Jim Homer. As project leader, he kept us working closely together toward precise goals, but in an at mosphere encouraging personal creativity.^ References 1. D.C. Chu, "Time In terva l Averaging: Theory , Prob lems, and Solutions," Hewlett-Packard Journal, June 1974. 2. K.J. Jochim and R. Schmidhauser, "Timer/Counter/DVM: A Synergistic Prodigy?," Hewlett-Packard Journal, April 1970.

Wil l iam D. Jackson Bi l l Jackson des igned the HP interface bus option for the 5328A Universal Counter. He's been with HP since 1 970 and had previously des igned a remote p rog ramming in te r face fo r the 5326/27 Coun ters. B i l l was born in Tuscon, Ar izona, graduat ing in 1970 wi th a BSEE degree. He's s ingle, l ikes to play tennis and l isten to music, and is an of f ic ia l o f the HP Santa Clara Div is ion basketbal l league. He l ives in San Jose.

Alfred Langguth Al Langguth des igned the two un ive rsa l modu les and the chan ne l C modu le and cont r ibu ted to the overa l l system des ign of the 5328A Universa l Counter . Wi th HP s ince 1971, he 's a lso done design work on the 5327 Counter ser ies. Original ly f rom Blythe, Cal i forn ia, Al received his BSEE degree f rom Massachuset ts Inst i tu te of Technology in 1970 and h is MSEE f rom Stan fo rd Un i vers i ty in 1971 Las t year , he and h is w i fe moved to Los A l tos , Cal i fo rn ia , where A l now spends

much of h is t ime adding a new garage and woodworking shop to the i r home He's a lso in terested in photography and en joys outdoor act iv i t ies l ike bik ing, camping, and exercis ing his dog.

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© Copr. 1949-1998 Hewlett-Packard Co.

Synthesized Signal Generator Operation to 2 .6 GHz with Wideband Phase Modulat ion New p lug- ins g ive the Mode l 8660-ser ies Synthes ized S i g n a l G e n e r a t o r s p r o g r a m m a b l e o p e r a t i o n t o 2 . 6 G H z and a capabi l i ty fo r phase modula t ion.

by James A . Ha l l and Young Dae K im

INCREASING USE OF THE 2-GHz bands for long- haul communications and for satellite tracking

and telemetry, coupled with the growth of navigation al systems, radio astronomy and meteorology in the 1 to 2 GHz band, has intensified the need for precision test instrumentation for this range of frequencies. At the same time there is increasing use of phase modulation ((/>M) in communications, telemetry, and navigation systems, so phase-modulated signal generators are needed for testing these systems.

These needs have led to the development of new plug-in sections for the Hewlett-Packard 8660-series Synthesized Signal Generators: a l-MHz-to-2.6-GHz

RF section, and modulation sections with the ca pability for wideband phase modulation. (Fig. 1).

First introduced in mid-1971, the 8660-series Synthesized Signal Generators combine the fre quency resolution, stability, and programmability of a synthesizer with the output level calibration and AM/FM capability of a signal generator. l2 The new Model 86603A RF Section plug-in increases the fre quency range, providing synthesized frequencies from 1 MHz to 2.6 GHz in 1-Hz steps up to 1.3 GHz, and 2-Hz steps up to 2.6 GHz. The output level is cali brated and adjustable over a range of more than 140 dB, from at least +7 dBm down to - 136 dBm (0.5 V to

F i g . 1 . T h e 8 6 6 0 - s e r i e s S y n t h e s ized Signal Generator fami ly has t h r e e n e w p l u g - i n s a n d a n e w m a i n f r a m e . T h e n e w R F S e c t i o n p lug- in g ives f requency coverage t o 2 . 6 G H z a n d t h e o t h e r t w o p lug - ins a re modu la t i on sec t i ons t h a t a d d p h a s e m o d u l a t i o n c a pab i l i t y . The new ma in f rame and p l u g - i n s a r e c o m p a t i b l e w i t h p r e v i o u s l y a n n o u n c e d p l u g - i n s and main f rames (background) .

15

© Copr. 1949-1998 Hewlett-Packard Co.

0.03 fj.V rms). Phase modulation capability (<¿>M) is provided by

two new modulation section plug-ins, the Models 86634A and 86635A, operating in conjunction with the RF section. Model 86635A provides both FM and 0M capability with maximum modulation rates of 1 MHz in either mode. Model 86634A is a phase- modulation-only plug-in that provides wideband modulation rates with good linearity from dc to 10 MHz. Peak 4>M deviation of either plug-in is me- tered and adjustable up to ±100° for center frequen cies up to 1.3 GHz, and ±200° for center frequencies between 1.3 and 2.6 GHz. This capability, coupled with the >140 dB calibrated output range, makes the synthesized signal generators useful for testing a wide variety of phase-modulated systems, and for lab-bench applications such as examining the be havior of phase-lock loops.

The previously announced modulation sections in the 8660 series may also be used with the new 2.6-GHz RF section to give AM, FM, and pulse modu lation, in addition to the new phase modulation ca pability. Programmability of frequency, output level and modulation via either BCD or the HP Interface Bus also makes the new 2.6-GHz capability available for computer or calculator controlled test systems.

Operat ion to 2 .6 GHz The 2.6-GHz frequency range is obtained in the

new Model 86603 A RF Section with the help of a fre quency doubler. A block diagram is shown in Fig. 2. Basic operation for output frequencies below 1300

MHz is identical to the earlier Model 86602 A 1 - 1 300 MHz RF Section:3 two microwave signals, one that is varied in 100-MHz steps and one that is varied in 1-Hz steps, are supplied by the synthesizer main frame; these are heterodyned to derive a difference frequency that is routed through an ALC detector and an attenuator to become the output signal. The micro wave signal that varies in 1-Hz steps has a range of 3.95 to 4.05 GHz, and the 100-MHz-step signal ranges from 2.75 to 3.95 GHz, giving a maximum difference frequency of 1.3 GHz.

To obtain frequencies above 1.3 GHz, the differ ence frequency is routed via a relay to the frequency doubler. The doubler consists of a transmission-line balun that produces two signals 180° out of phase, bal anced with respect to ground. The signals drive a full- wave Schottky-diode rectifier resulting in a rectified sine wave that has a strong component at twice the input frequency. The input fundamental and odd harmonics are suppressed by the circuit's symmetry.

The double-frequency signal is processed through a bandpass filter that further suppresses the input fundamental and the third and higher harmonics. The filter, however, has a wide enough passband (500 MHz) to allow wideband modulation sidebands to pass with minimum distortion. The filter is tuned to track the output frequency by a varactor under control of digital signals originating in the mainframe digital control unit (DCU).

Following the filter, the signal is amplified by two identical amplifiers in cascade to bring it up to the de sired level. It was possible to use inexpensive

Doubler Assembly D C U

Phase Modulator (Option 002)

3 .95-4 .05 GHz 1-Hz Steps

From Mainf rame

2 .75-3 .95 GHz 100-MHz Step;

4-GHz Bandpass

Filter

Isolator

DCU A M P u l s e I n M o d u l a t i o n

In

-2600 MHz Out

Fig. Microwave supplied diagram of the Model 86603A 2.6-GHz RF Section. Microwave signals supplied by the main f rame g ive d i f fe rence f requenc ies over a range o f 1-1300 MHz in 1-Hz s teps. The f re

quency doub le r ex tends the ou tpu t range to 2600 MHz.

16

© Copr. 1949-1998 Hewlett-Packard Co.

5 0 0 1 0 0 0 1 5 0 0 2 0 0 0 C e n t e r F r e q u e n c y ( M H z )

2600

Fig . 3 . Outpu t leve l as a func t ion o f f requency fo r a t yp ica l Mode l 86603A RF Sec t ion a t an ou tpu t leve l o f +3 dBm.

printed-circuit micros trip techniques for these ampli fiers because they need to work only over an octave bandwidth and the discrete transistors used have more than enough gain in this frequency range. Each amplifier uses two HP type 35821E transistors in a common-emitter configuration and has appiuAÃ- mately 12-dB gain over the 1.3-to-2.6-GHz range.

The amplified signal goes to the output attenuator. It is also detected, amplified, and routed back to a PIN-diode modulator in the signal path of one of the input microwave signals to give closed-loop control of the output amplitude, thus compensating for the frequency response of the system. The output flatness achieved is shown in Fig. 3.

Because the output frequency is double that de rived from the microwave signals provided by the mainframe, a new mainframe (Model 8660C) has been designed using a modified digital control unit to give direct selection and display of the output fre quency throughout the full 1-2600-MHz range. This mainframe operates identically to the Model 8660B at output frequencies below 1300 MHz, but when used with the Model 86603A 2.6-GHz RF Section, it automatically switches in the RF section's doubler when a frequency above 1300 MHz is selected and computes the two microwave frequencies needed to supply one-half the output frequency to the doubler. As far as the operator is concerned, the Model 8660C operates the same whether the output is below 1300 MHz, above 1300 MHz, or when stepping, sweeping, or tuning through 1300 MHz. Frequency resolution above 1300 MHz, however, is 2 Hz rather than 1 Hz because of the frequency doubling.

For use in the earlier 8660A and 8660B main frames, an optional version of the2.6-GHz RF section (opt 003) is available. This version has a front-panel switch for inserting the doubler in the signal path

(the switch may also be activated by remote control). With the doubler in, the output frequency is twice that indicated by the mainframe.

Phase Modulator Operat ion Equipping an 8660-series generator for phase

modulation requires one of the two $M modulation sections and an RF section equipped with the phase- modulator factory-installed option (opt 002). Either Model 86602B (1-1300 MHz) or Model 86603A (1-2600 MHz) may be supplied with this option. The actual phase modulation takes place in the RF sec tion while the modulation section supplies the internally-generated modulating signals, establishes the modulation level for either internal or external signals, and meters the peak deviation.

As was indicated in Fig. 2, the phase modulator is in the signal path of the 3.95-4.05 GHz microwave in put signal in the RF section. The modulator is a four-port circulator with varactor-diode phase shifters at two of the ports (Fig. 4). Each diode is biased at an operating point that causes the diode capacitance to

" O l d e r t h a t w i t h s e r i a l n u m b e r p r e f i x 1 3 4 9 A a n d l o w e r n e e d a f i e l d u p d a t e k i t t h a t i n s u r e s p r o p e r d a m a g e c o n d i t i o n s a n d p o w e r s u p p l y r e g u l a t i o n t o a v o i d p o s s i b l e d a m a g e t o I t i e n e w R F s e c t i o n . D e t a i l s a r e a v a i l a b l e a t H P f i e l d o f f i c e s

-w^- P h a s e - M o d u l a t e d

3 . 9 5 - 4 . 0 5 G H z

U n m o d u l a t e d 3.95-4.05 GHz

f rom Mainframe

Modulating Signal from Modulating

Section

A r c t s n g e n t C o m p e n s a t i o n

Fig. 4 . Phase modulator is inser ted in the path of one of the m i c r o w a v e s i g n a l s i n e i t h e r t h e 8 6 6 0 2 A 1 . 3 - G H z o r t h e 86603A 2.6-GHz RF Sect ions.

17

© Copr. 1949-1998 Hewlett-Packard Co.

Phase Modulation

What is Phase Modula t ion? P h a s e m o d u l a t i o n ( & M ) i s a f o r m o f a n g u l a r m o d u l a t i o n

in which the phase of a carrier wave is varied in proportion to the instantaneous ampl i tude of the modulat ing s ignal . To character ize the amount of <¿>M on a carrier wave, a modulation index, m, is def ined:

m = A * p e a k where A<2>Deak = peak phase deviation of the carrier in radians. The re i s a d i rec t re la t i onsh ip be tween modu la t i on i ndex and the resul tant carr ier and s ideband leve ls , as shown for the car r ier and f i rst through third sidebands in Fig. A. Note that with no phase dev ia t ion (m = 0) the car r ie r leve l i s un i ty and the s ide band levels are zero. As the deviat ion is increased, the carr ier l e ve l dec reases and t he s i deband l eve l s i nc rease . The s i de bands are spaced a t a f requency in terva l equa l to the modula t ion f requency.

-Carr ier First Sideband

Second Sideband

1 2 3 Modulat ion Index (Radians)

Fig. A

How is Phase Modulat ion L ike Frequency Modulat ion? In frequency modulat ion (FM), a more famil iar type of angular

modulat ion, the f requency of a carr ier wave is var ied in propor t ion to the ins tan taneous ampl i tude o f the modu la t ing s igna l . Because f requency is the t ime der iva t ive o f phase, < t>M and FM are direct ly related. The equivalent modulat ion index, m, for FM is:

where Af = peak f requency deviat ion in Hz and fm is the modu lat ion frequency. With sinusoidal modulat ion, <t>M and FM with the same modulat ion index have exactly the same RF spectrum. Th is is i l lus t ra ted by a compar ison of F igs. B1 and B3.

H o w D o e s P h a s e M o d u l a t i o n D i f f e r f r o m F r e q u e n c y Modulat ion?

In <¡>M the modulation index depends only on the amplitude of t he modu la t i ng s i gna l and i s i ndependen t o f t he modu la t i on f requency . There fo re , as the modu la t ion f requency i s var ied , the s ideband leve ls in <2>M remain constant and on ly the f re quency spac ing var ies . Th is is i l lus t ra ted by F igs . B1 and B2. Cont ras ted w i th th is , the modu la t ion index in FM depends on both the f requency deviat ion, determined by modulat ing s ignal a m p l i t u d e , a n d t h e m o d u l a t i n g f r e q u e n c y . F o r c o n s t a n t f r e quency dev ia t ion , the modu la t ion index var ies inverse ly w i th modulat ion frequency. This is i l lustrated by comparing Figs. B3 and B4.

Look ing a t th is another way, the amount o f f requency dev ia t i on genera ted by phase modu la t i ng a ca r r i e r va r ies d i rec t l y wi th the modulat ing f requency. For th is reason large f requency deviat ions can be generated using high-rate <i>M.

The re la t ionsh ip between FM and <¿>M is impor tant in un ders tand ing the e r ro rs tha t can resu l t i n measurements invo l ving a mixture of FM and d>M equipment. For example, when us ing an FM signal generator to test a <i>M receiver, the receiver demodu la ted ou tpu t d rops 6 dB per oc tave as the modu la t ing frequency increases. This must be accounted for when measur-

be series-resonant with its own lead inductance at 4 GHz.

The capacitance C of the abrupt-junction diodes varies approximately as l/Vv7 where V is the bias voltage:

C = K/VV~ where K is a constant. Neglecting the loss in the varac tor circuit, the reactance X of the series-tuned cir cuit as seen by the circulator is:

v ! ^ X = w 0 L â € ” = c o 0 L -

Therefore, for a given RF frequency u>0, the reac tance X of the series-tuned circuit varies approxi mately as the square root of the voltage that biases the diode.

Again neglecting the loss in the varactor circuit, a voltage wave incident on the varactor circuit is re flected with a phase shift <b:

(¿> = -2 arctan (X/ZJ,

where X is the varactor circuit reactance and Z0 is the characteristic impedance of the transmission- line circuit. From this it can be seen that as X varies from -se to + 3c, c¿> varies ±TT radians (±180°).

The non-reciprocal characteristic of the ferrite cir culator separates the reflected wave from the wave in cident on the varactor circuit. The signal applied to port 1 is coupled to port 2. The part reflected from port 2, which is phase-shifted according to the varac tor reactance, is coupled to port 3. Using two circu lators in series with their varactor circuits driven in parallel gives a theoretical phase-modulation range of ±360°. However, by restricting the range to ±100°, good modulation linearity is assured.

The phase-modulator driver circuit uses the square-law response of a FET to compensate for the square-root relationship between the reactance X and the varactor voltage V. A diode-resistor circuit gives a non-linear attenuation that compensates for the arctangent relationship between the reactance X

18

© Copr. 1949-1998 Hewlett-Packard Co.

i ng modu la t i on f r equency response o r when cons ide r i ng t he FM genera tor ' s con t r ibu t ion to overa l l harmon ic d is to r t ion . A l so, care must be exerc ised to ensure that the maximum al lowa ble phase deviat ion of the r f>M receiver is not exceeded at low modulat ion f requencies.

W h e n m o d u l a t i n g w i t h n o n - s i n u s o i d a l w a v e f o r m s ( s q u a r e wave, triangular wave, etc.), the task of simulating <¿>M with an FM generator is more di f f icul t . This is because these more com p l e x w a v e f o r m s c o n t a i n h a r m o n i c f r e q u e n c i e s t h a t m u s t p r o d u c e t h e p r o p e r a m o u n t o f d e v i a t i o n w i t h r e s p e c t t o t h e fundamental .

These d i f f i cu l t ies can be overcome to some ex ten t by pass ing the modulat ing signal through a di f ferent iator to give a 6 dB pe r oc tave r i se i n amp l i t ude as t he modu la t i on f r equency i n c reases (p re -emphas is ) , thus ma in ta in ing a cons tan t modu la t i o n i n T h e l i m i t a t i o n s o f t h i s a p p r o a c h a r e d i s c u s s e d i n the main text.

A Second Di f ference With FM, the instantaneous carr ier f requency var ies d i rect ly

w i t h t he modu la t i ng s i gna l amp l i t ude and t he max imum and minimum carr ier f requencies occur at the posi t ive and negat ive peaks o f a s inuso ida l modu la t ing wavefo rm. In <¿>M the in

stantaneous frequency varies as the t ime derivat ive of phase so t he max imum and m in imum ca r r i e r f r equenc ies occu r a t t he z e r o c r o s s i n g s o f t h e s i n u s o i d w h e r e t h e w a v e f o r m r a t e o f change is h ighest .

This 90° phase difference is only important when considering secondary e f fec ts such as how inc identa l AM generated in an FM or ¿M modulator affects the output RF spectrum. It has l itt le p r a c t i c a l s i g n i f i c a n c e w h e n u s i n g < t > M t o s i m u l a t e F M o r vice versa.

Why <t>M With the many similarities that exist between <i>M and FM, why

use phase modu la t i on? One impo r tan t r eason i s t ha t phase modulat ion can be appl ied di rect ly to a carr ier s ignal and does not require modulation of the carrier osci l lator as in FM. For this reason, <t>M is often used in systems where a highly stable car r ie r f requency is requ i red , as in nar rowband communica t ions and t e l eme t r y sys tems . I n d i g i t a l commun i ca t i ons sys tems , p h a s e s h i f t k e y i n g ( p h a s e m o d u l a t i o n w i t h d i s c r e t e p h a s e s t a t e s ) i s o f t e n u s e d b e c a u s e a s t a b l e c a r r i e r f r e q u e n c y a t t h e s o u r c e e a s e s t h e p r o b l e m o f c a r r i e r r e c o v e r y a t t h e r e c e i v e r , a n d i n m a n y a p p l i c a t i o n s i t o f f e r s b e t t e r s p e c t r a l e f f i c i ency (more b i t s pe r Hz o f RF bandw id th ) than f requency - sh i f t key ing (FM wi th d iscrete f requency states) .

Rate = 20 kHz Index = 1.45

1 0 0 k H z 1 . 4 5

<',M: Deviation = 83 (Peak)

Rate = 20 kHz Index = 1.45

1 0 0 k H z 0.29

FM; Deviation = 29 kHz (Peak)

and phase shift 4>. Although the tuned varactor circuit would theo

retically allow optimum modulation linearity to be achieved at only one microwave frequency, good phase linearity for all the RF section's output fre quencies is obtained because the input microwave frequency varies only ±1.25% (3.95 - 4.05 GHz). Phase modulation distortion is typically less than 2% at modulation rates up to 1 MHz and typically is less than 5% at 10 MHz. The system can actually be driven through a range of ±115° with very little loss of lin earity. This range is doubled to ±230° in the Model 86603A RF section at carrier frequencies above 1.3 GHz where the frequency doubler is used.

The sensitivity of the modulator varies by about 5% as the microwave frequency varies from 3.95 to 4.05 GHz. This is compensated for by a variable atten uator that is controlled digitally by the mainframe.

Modulator incidental AM is typically less than 10%. This relatively low AM and the low distortion are

preserved by the use of wideband circuits following the modulator.

An alternative approach to phase modulation would have been to use an FM modulator with the fre quency response of the modulator shaped so its out put would simulate phase modulation (see box). A problem arises, however, because the frequency deviation then varies as a function of the modu lation frequency. At high modulation frequencies, very wide deviations are required, placing a practi cal limit on the upper modulation frequency. At low modulation frequencies, very low deviations result and a lower limit is imposed by residual FM noise. At best, these two extremes encompass a frequency range of three or four octaves.

No such limits exist with the phase-modulation scheme used in the 8660 system. A modulating fre quency range of dc to 10 MHz is thus possible. Acknowledgments

Our acknowledgment and thanks to Hamilton

19

© Copr. 1949-1998 Hewlett-Packard Co.

Chisholm for the design of the digital control unit in the 8660C mainframe, to Skip Crilly for the design of the 86634A Modulation Section, to Charlie Sallberg for the design of the 8663 5A Modulation Section and also for environmental testing of the new mainframe and modulation and RF section plug-ins, to Charles Cook for the plug-in product design, to John Hasen for his guidance of the modulation section design and his investigation of phase modulation character ization, and especially to Section Manager Brian Unter for his guidance and counsel throughout the

project and for his many contributions to product definition.^

References 1. J.C. Shanahan, "Uniting Signal Generation and Signal Synthesis," Hewlett-Packard journal, December 1971. 2. J.E. Stinehelfer, "The Well-Modulated Synthesizer," Hewlett-Packard Journal, January 1972. 3. R. Hassun, M. Humphreys, D. Scherer, Y.D. Kim, B. Stribling, C. Cook, "Synthesized Signal Generation to 1.3 GHz," Hewlett-Packard Journal, March 1973.

James A. Hal l With 13 years experience in micro wave communica t ions , J im Ha l l joined HP ¡n 1972 to work on a prototype satel l i te TV receiver (HPJ Aug. 74) . When tha t was completed, he was named pro ject manager fo r the 8660 sys tem Born in Hal i fax, Virg in ia, J im has a BSEE from North Carol ina State Univers i ty and an MS in Phys ics f rom Lynchburg Co l lege (Va. ) . Wi th a daughter , 8 , and a son, 5 , J im's major le isure- t ime act iv i t ies have become fami ly camping and fishing. \

Young Dae K im A graduate o f Seoul Nat iona l University (BSEE), Young Dae Kim worked as a m ic rowave eng ineer in Korea before obta in ing a scholarsh ip that led to an MSEE degree at South Dakota State Univers i ty . He jo ined HP in 1969 and is doing further graduate work at Stanford in the HP Honors Co-op p rogram. K im worked on the 11 661 A Frequency Extension Modu le be fo re becoming p ro jec t eng ineer fo r the 86603A RF Sec t ion. Kim's married, and he enjoys table tennis and oriental checkers.

8 6 6 0 C S y n t h e s i z e d S i g n a l G e n e r a t o r M a i n f r a m e F R E Q U E N C Y S E L E C T I O N : k e y b o a r d c o n u o t p a n e l a l l o w s s e i e c l t o n o à ­ C W ( o r

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R E F E R E N C E O S C I L L A T O R :

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h o u r s E X T E R N A L R e a r - p a n o l s w i t c h a M o w s o p e r a t i o n f r o m 5 - o r 1 0 - M H z s i g n a l

R E F E R E N C E O U T P U T : R e a r - p a n e l B M C c o n n e c t o r p r o v i d e s 0 5 t o 1 V r m s

( I N T ) i n t o 1 7 0 o h m s o r v o l t a g e n o m i n a l l y e q u a l t o e x t e r n a l i n p u t ( E X T )

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D I G I T A L S W E E P :

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M a n u a l S w e e p S W E E P W I D T H c o n t i n u o u s l y a d j u s t a b l e o v e - f a n g e o l R F s e c t i o n i n s t a l l e d S W E E P E N D P O I N T A C C U R A C Y s a m e a s r e f e r e n c e o s c i l l a t o r a c c u r a c y

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s i n g S T E P o r S T E P , k e y i n c r e m e n t s f r e q u e n c y u p o r d o w n b y d e s i r e d s t e p s i z e

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f r e q u e n c y b y p r e n o u V y s M e c l e u s t e p s i z e O U T P U T L E V E L : p r o g r a m m a b l e m t - d B S t e p s o v e r o u O u t r a n g e o l R F s e c t i o n

P R O G B A M t a - i l G I N P U T

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( a t 0 - 3 0 * . A M a n d C F * 1 0 M H z ) D I S T O R T I O N l i t 3 0 % A M )

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W E I G H T : 5 k g ( 1 1 C s i H 6 6 1 B F r e q u e n c y E x l e n s . o n M o d u l e ( r e q u i r e d ) a d d s

i 8 k g < 4 K M

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- M R A T E : d c 1 0 M H z

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I N P U T L E V E L R E Q U I R E D O F E X T E R N A L S O U R C E 2 V r m s m a x

I N P U T I M P E D A N C E 5 0 1 1

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M o d e l 8 6 6 3 5 A M o d u l a t i o n S e c t i o n ' R e q u i r e s R F S e c t i o n w i t h o p t i o n 0 0 2 )

F M A N D Â « M R A T E : d c 1 M H z

M O D U L A T I N G S I G N A L : ( S a m e a s t o r M o d e l 8 6 6 3 4 * a b o v e e x c e p t i n p u t i m

p e d a n c e 6 0 0 1 1 )

M E T E R " . - : 1 0 0 1 0 0 0 k H z F M 0 - 1 0 0 a * l f o > C F - - 1 3 G H z 0 - 2 0 2 0 0 . 2 0 0 0 K H z

F M 0 - 2 0 0 Â ° Â « M

F M C E N T E R F R E Q U E N C Y L O N G - T E R M S T A B I L I T Y : T y p i c a H y l e s s t h a n

2 0 0 H i h r D r i f t c a n b e r e m o v e d b y p r e s s i n g F M C F C A L o u t r a n w h i c h p h a s e

' O C k s F M o s o l a t o r m o m e n t a r i l y t o m a i n f r a m e r e l e r e n c e

R E M O T E P R O G R A M M I N G

FUNCTIONS im Ext «M or FM AISO FM CF CAL M O D U L A T I O N S E T T I N G R E S O L U T I O N 1 5 0 o f r a n g e s e l e c t e d

W E I G H T : 2 6 k g ( 6 t  » )

P R I C E S I N U S Â »

M n m u m k e y b o a r d - c o n t r o H e d M t r a m u m C W Â « M S y s t e m

2 6 - G H Z S y s t e m

8 6 6 0 C M a w t f r a m i $ 7 9 0 0 0 6 6 0 A M * n f r a m e  « 2 0 0

8 6 6 3 1 8 A u u h a r y B 6 6 3 4 A * M M o d u l a t i o n S e c f t o n $ 1 . 4 0 0

S e c t o r $ 3 0 0 1 1 6 6 1 B F r e q u e n c y 1 1 8 6 1 B F r e q u e n c y E x t e n s i o n M o d u l e S 3 0 0 0

E x t e n s * * M o d u l e S 3 0 0 0 9 6 6 0 3 A 2 6 - G M z R F B 6 6 0 2 B t 3 G H z R F S e c t i o n & 4 . 1 0 0

S e c t i o n S 6 0 0 0 O p t i o n 0 0 2 ( Â « M c a p a b M y ) S ' 5 0 0

T o t a l 1 1 7 2 0 0 $ 1 6 2 0 0

M o d e 8 6 6 3 5 *  « M F M M o d u l a d o r S A C K * $ 2 2 0 0 M o O W 8 6 6 0 3 A ( o p t 0 0 3 ) 2 6 G H i R F S e c  » o n t o r u s e w K h 9 6 6 0 A 8 6 6 0 6

m a r t r a m e S 6 2 5 0 M A N U F A C T U R I N G D I V t S I O N S T A N F O R D P A R K D I V I S I O N

1 5 0 1 P a g e M i R o a d

P a l o A M O C a M o m a 9 4 3 0 4

2 0

© Copr. 1949-1998 Hewlett-Packard Co.

Applications of a Phase-Modulated Signal Generator Once a capabi l i ty is made avai lab le, appl icat ions emerge. Here are a few that have been found for the phase-modulated synthes ized s ignal generator in HP's own laborator ies.

by James A. Hal l

AS THE NUMBER OF APPLICATIONS for phase- lock loops (PLLs) grows, more and more en

gineers are faced with the need to evaluate the perfor mance of these devices. FM signal generators can be used for this purpose but there are several advan tages to using a synthesized signal generator with phase modulation (<£M) capability. For example, a synthesizer's output is locked to a crystal reference that is accessible to external circuits so all the oscilla tors in a complicated measurement set-up can be locked to a common reference.

A phase-modulated synthesizer permits a mea surement of frequency response to be swept while maintaining constant phase deviation. This avoids the excessive phase deviation at low modulation fre quencies and the loss of signal-to-noise ratio at high frequencies that would occur with FM. For measure ments of PLL transient response, phase modulation is a necessity.

Fig. 1 shows a basic PLL measurement system. By measuring the PLL error voltage, the closed-loop error as a function of input phase deviation and as a function of slew rate can be determined. From these measurements, loop bandwidth and dynamic range can be inferred.

Closed-loop gain can be measured by using a spec trum analyzer to examine the output of the PLL's voltage-controlled oscillator. Closed-loop gain is easily determined by taking advantage of the fact that with <¿>M the modulation index remains constant as the modulation frequency varies. At a modulation index of less than 0.5 (30° peak deviation), the first- order sidebands of a phase-modulated wave can be considered proportional to the modulation index and the higher sidebands are of insignificant pro portions. By noting the VCO sideband level as a func tion of modulation frequency, one has a measure of the PLL closed-loop response. Loop bandwidth and peaking can be measured readily with this technique.

" T h e e q u a l s o f t h e f i r s t s i d e b a n d a m p l i t u d e t o c a r r i e r a m p l i t u d e e q u a l s a p p r o x i m a t e l y o n e - h a l f t h e m o d u l a t i o n i n d e x .

Phase-Modulated Signal Generator

PLL Under Test

PLL Error Voltage

M o d u l a t i o n S o u r c e

VCO Low-Pass Filter

F i g . 1 . S y s t e m f o r m e a s u r i n g a p h a s e l o c k l o o p ' s c l o s e d - loop ga in and er ror .

These measurements give results in terms of mag nitude only. By using a network analyzer to measure phase response as well as magnitude, one obtains the complete closed-loop transfer function. For this test, the output of the PLL VCO must be demodulated in a phase detector.

One set-up for doing this is diagrammed in Fig. 2. Synthesized signal generator #2 serves as a stable lo cal oscillator for the phase detector. By using a vari able dc source at the phase modulator input, the out put phase of the signal generator can be offset to operate the phase detector in the center of its linear range. Gain-phase measurements are then made in the usual way.

At high modulation rates, the excess phase contri bution of synthesized signal generator #1 must be considered. Typical excess phase shif t of the 86602A/86603A RF Section operated with the 86634A or 86635A Modulation Section is shown as a function of frequency in Fig. 3. Note that the excess phase shift is a linear function of frequency and may therefore be considered as a constant time delay, equivalent to a length of lossless transmission line.

The method of generating a local-oscillator signal shown in Fig. 2 is only one of several possible. With it, however, measurements are possible at modula-

21

© Copr. 1949-1998 Hewlett-Packard Co.

PLL Under Test

vco

F i g . 2 . M e a s u r i n g t h e c l o s e d - loop t rans fe r func t ion o f a phase lock loop.

lion rates down to dc, unlike systems that use a phase- lock loop to recover the carrier for use as an LO signal. The set-up of Fig. 2 also allows measurements in sys tems where the input RF frequency differs from the VCO frequency, for example where frequency divid ers are included in the loop.

The transient response of the PLL can be measured by observing the phase demodulator output with an oscilloscope while driving the modulation input of signal generator #1 with a square wave or pulse train.

Phase Detector Character izat ion Phase detector circuits can be difficult to evaluate

for linearity and sensitivity and for changes in these characteristics with changes in frequency or input level. This is particularly true of some of the new all- digital phase-frequency detectors.

One quick method of evaluating a digital phase detector is shown in Fig. 4. The "n-N" block is one means of getting a reference frequency within the range of the phase detector under test. Another syn

thesized signal generator could be used in place of a frequency divider as was done in Fig. 2.

The response of a widely-used digital phase- frequency detector obtained by the set-up of Fig. 4 at two different frequencies is shown in Fig. 5.

Digita l Phase Modulat ion The fact that the phase-modulation technique

used in the 8660 system is an analog method does not preclude its use as a digital phase modulator when evaluat ing various digi ta l phase-modulat ion schemes or formats.

Biphase modulation is obtained by driving the syn thesized signal generator's modulation section with a square wave. If the square wave is symmetrical with respect to ground (no dc offset) and the modula tion meter reads 90° peak deviation, the modulation will be ± 90°. This results in a 1 80° difference between the two phase states.

Adding the output of two digital generators to ob tain a four-level modulation signal results in a quad- riphase PCM signal. If phase states of 0°, 90°, 180°,

240

£ 200

W 160=

| 120

£ 80

40

D

86635A

8 6 6 3 5 A D C M o d e

AC Mode Td - 160 ns

Ta - 230 ns

1 2 3 4 5 6 7 8 9 Modulat ion Frequency (MHz)

60 ns

10

F i g . 3 . E x c e s s p h a s e s h i f t o f t h e 8 6 6 0 - s e r i e s m o d u l a t i o n sys tem measured be tween the modu la t ion sec t ion inpu t and the RF sect ion output .

10 MHz Reference Out

Synthesized Signal

Generator /

Frequency • 10 MHz N

.f.M Signal

Function Generator

Detector Under Test

Osci l loscope

Fig. 4 . Set-up for measur ing phase detector character is t ics .

22

© Copr. 1949-1998 Hewlett-Packard Co.

Measuring Modulation Linearity

T h e d i s t o r t i o n i n t h e o u t p u t o f a p h a s e m o d u l a t o r c a n b e e x p r e s s e d i n a n u m b e r o f w a y s . F o r t h e 8 6 6 0 s y s t e m , i t i s spec i f i ed i n t e rms o f t o ta l ha rmon i c d i s to r t i on i n t he demod u la ted output s igna l . Th is is measured wi th a very l inear w ide band phase demodu la to r .

For some appl icat ions, l inear i ty in terms of dev iat ion f rom a bes t by l i ne i s mo re mean ing fu l . Th i s can be de te rm ined by measuring the der ivat ive ( incremental s lope) of the modulator 's p h a s e - v s - v o l t a g e c h a r a c t e r i s t i c , a t e c h n i q u e t h a t h a s b e e n used fo r many years in tes t ing FM communica t ions sys tems. " The derivative of the l inear part of the characteristic is a dc term, second-order curvature resul ts in a s lope, and so on. By numer i c a l l y i n t e g r a t i n g t h e d e r i v a t i v e , a n a c c u r a t e p l o t o f p h a s e change versus vo l tage is obta ined. Th is method has an advan tage in that a per fect demodulator is not required for the meas urement.

A system for measuring the incremental s lope is shown in the diagram. Two modulat ing s ignals are appl ied s imul taneously to the phase modu la to r under tes t . One i s a ve ry low f requency signal (~0.1 Hz) wi th suff ic ient ampl i tude to give ful l -scale mod u la t ion , in th is case ±100°. The o ther i s a low- leve l , h igh- f requency tone (>50 kHz) that g ives about 2° dev iat ion. Th is s ignal var ies the input vo l tage by a smal l amount AV whi le the low-frequency s ignal moves the AV var iat ion over the complete input range o f the modula tor , and hence over the modula tor 's complete character is t ic .

The modulated output of the unit under test is demodu lated by a nar row-bandwid th phase- lock loop . The VCO o f the phase-

Phase-Modulated Signal Source

100 kHz Bandpass

Fi l ter

Envelope Detector

(-100°) Modulat ing Voltage

(+100=)

l o c k l o o p t r a c k s t h e l o w - f r e q u e n c y p h a s e v a r i a t i o n s q u i t e closely, maintaining a very low error output. This means that the phase de tec to r opera tes a t essen t ia l l y the same po in t on i t s cha rac te r i s t i c a t a l l t imes and t hus does no t con t r i bu te any nonl ineanty to the measurement.

The high-f requency tone, being outs ide the bandwidth of the phase- lock loop, appears super imposed on the er ror .vo l tage wi th an ampl i tude propor t iona l to the s lope of the modula tor 's phase-versus-vo l tage charac te r is t i c . I t i s ampl i tude de tec ted and then appl ied to the Y axis of an X-Y recorder. The X axis is d r i ven by the low- f requency modu la t ing s igna l . The resu l t ing plot is a graph of modulat ion sensit iv i ty versus input voltage or, pu t t ing i t another way, the der iva t ive o f phase change versus input vol tage.

By i n t eg ra t i ng t h i s cu r ve numer i ca l l y , an accu ra te p l o t o f phase change versus vol tage input is der ived. I f the dc compo nen t o f t he modu la t i on sens i t i v i t y cu rve i s i gno red , t hen the constant-s lope term in the modulat ion character ist ic drops out. The resulting curve is a plot of deviation from best straight l ine. A p l o t t y p i c a l o f a M o d e l 8 6 6 0 3 A R F s e c t i o n ' s m o d u l a t o r a t minimum and maximum modulator carr ier f requencies is shown above.

•R. Urquhart, A New Microwave Link Analyzer with High-Frequency Test Tones. Hewlett- Packard Journal, September 1972.

and 270° are desired, the modulator is driven ±135°. This is well within the capabilities of the 8660 sys tem at RF frequencies above 1.3 GHz and can typical ly be obtained at lower frequencies by overdriving the system to ±135°.

The capabilities of the 8660 system make it un necessary to build VHP, UHF, or microwave digital phase modulators from scratch when evaluating digi tal phase modulation schemes. The problem is re duced to building the necessary baseband waveform generators. Waveform Synthesis

In designing circuits and systems, it is often neces

sary to evaluate the effects of harmonic distortion on performance. For this, a test signal with known and controllable distortion is useful. This can be done with the set-up shown in Fig. 6.

The two synthesized signal generators are locked to a common reference frequency and their outputs are summed in a resistive adder. By using a control lable dc voltage to vary the phase of the 2f0 generator, and by varying its output with the level control, a sig nal is obtained that has a second harmonic variable in amplitude and phase. An analysis of the effect of second harmonic distortion on the circuit under test can then be made. For example, the effect of second

2 3

© Copr. 1949-1998 Hewlett-Packard Co.

-100° + 100C I n p u t F r e q u e n c y = 1 M H z

s, i Ã̄ o

-100° O c + 1 0 0  ° I n p u t F r e q u e n c y = 1 0 M H z

Fig. 5 . Ef fect o f f requency on the p e r f o r m a n c e o f a d i g i t a l p h a s e - f r e q u e n c y d e t e c t o r , u s i n g t h e se t - up o f F i g . 4 f o r t he measu re ment.

harmonic distortion on the RF spectrum of an FM sig nal is readily made with this method.

Microwave Frequencies There are many times when it would be desirable to

use the frequency setability and modulation capabili ties of the 8660 system for conducting tests at fre quencies well above 2.6 GHz. These can be obtained by upconverting the 8660 output using a mixer with

a microwave signal generator serving as the local os cillator.

Because the 8660 system makes available a wide range of high frequencies, it is possible to select mixer input frequencies that space undesired mixing products far away from the desired output fre quency. The filtering problem is therefore greatly simplified and may even be non-existent because of the passband of the following circuits. S

Reference Out

S y n t h e s i z e d Signal

Generator F M M o d u l a t o r

S p e c t r u m Ana lyze r

. / , A d j u s t F i g . 6 . U s i n g t w o p h a s e - l o c k e d s ignal generators to synthes ize a s i gna l w i t h con t r o l l ab l e second - harmonic distort ion.

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© Copr. 1949-1998 Hewlett-Packard Co.